DESIGN FEATURES Tiny, Resistor-Programmable, µPower 0.4V to 18V Voltage Reference by Dan Serbanescu and Jon Munson Introduction The LT6650 is a 0.4V to 18V adjustable voltage reference that runs from low voltage and consumes only a few microamps. It features a low-dropout (LDO) characteristic, can source or sink current, can be configured in either series or shunt mode and saves space in the tiny 5-lead ThinSOT-23 package. Figure 1 shows a block diagram of the reference. Its 400mV internal voltage reference is connected to the non-inverting input of an operational amplifier. The inverting input is brought to a pin, thus making a series-mode reference adjustable to any output voltage from 400mV up to (VSUPPLY – 0.35V) by using two external resistors. It can also be configured as IN 4 to produce any precision “zener” voltage within the wide supply range (1.4V to 18V) by selection of the two external resistors. LT6650 VR = 400mV REFERENCE + 5 OUT – DNC 3 Specifications 1 FB 2 GND Figure 1. Block diagram of 1% accurate micropower 0.4V to 18V adjustable reference. a fixed 400mV reference by simply connecting the output to the op amp inverting input. While the LT6650 is designed as a series reference, it can be used as a shunt-mode reference simply by shorting the positive rail to the output pin—it can be programmed Table 1 summarizes the performance of the LT6650. The supply current is only 5.6µA and the supply voltage may range from 1.4V to 18V, which permits battery-powered equipment to be plugged into an unregulated wall adapter without the need for peripheral circuitry to limit the voltage input to the reference. The 400mV internal reference is ±1% accurate over the –40°C to 85°C temperature range and is also fully specified from –40°C to 125°C for extended temperature range Table 1. LT6650 Performance (Ta = 25°C, VIN = 5V, VOUT = 400mV, CL = 1µF, unless otherwise noted) Parameter Conditions Min Input Voltage Range –40ºC ≤ TA ≤ 125°C 1.4 Output Voltage –40ºC ≤ TA ≤ 85°C Line Regulation 1.4V ≤ VIN ≤ 18V Load Regulation Max Units 18 V 404 mV 1 % 1 6 mV 0 to –200µA (Sourcing) –0.04 –0.2 mV 0 to 200µA (Sinking) 0.24 1 mV Output Voltage Temperature Coefficient 396 Typ 400 –1 12 µV/°C VOUT = 1.4V Dropout Voltage IOUT = 0µA 75 IOUT = 200µA sourcing mV 250 mV Supply Current 1.4V ≤ VIN ≤ 18V 5.6 12 µA FB Pin Input Current VFB shorted to VOUT 1.2 10 nA Turn-On Time 16 100 0.5 ms Output Voltage Noise 0.1Hz to 10Hz 20 µVP–P Thermal Hysteresis –40°C to 85°C 100 µV Linear Technology Magazine • November 2004 DESIGN FEATURES 4 IN applications. The rail-to-rail output delivers 200µA in both sourcing and sinking modes of operation. Q3 Q4 Q5 Q6 Q7 Q17 R6 R1 R2 IN R3 R4 I1 Q8 Q10 RF = 2.5 • (VOUT – 0.4) • RG The worst-case FB pin bias current (IBIAS) can be neglected with an RG of 100kΩ or less. In ultra-low-power applications where current in the voltage programming resistors might OUT CN 1nF 1 FB RF 150k VOUT 1V CL 1µF RG 100k Figure 3. Battery powered pocket voltage reference runs for years on a coin cell. VOUT = 0.4V • (1 + RF/RG) VIN VOUT 5 OUT LT6650 GND 2 FB 1 CN 1nF RF CL 1µF RG Figure 4. Simple input network for improved supply rejection Linear Technology Magazine • November 2004 Q11 with impedance over about 50Ω. The output is adjustable from 0.4V up to the battery supply by selecting two feedback resistors (or setting a trimmer potentiometer position) to configure the non-inverting gain of the internal operational amplifier. A feedback resistor RF is connected between the OUT pin and the FB pin and a gain resistor RG is connected from the FB pin to GND. The resistor values are related to the output voltage by the following relationship: 5 IN Q15 Q16 IN D1 D2 Figure 2. LT6650 simplified schematic showing detail of low-dropout topology 2 CIN 1µF R8 1 FB VOUT = 0.4V • (1 + RF/RG) 4 Q21 Q14 2 GND GND RIN 1k Q9 I2 D3 VIN VS R5 5 OUT Q12 Battery Powered Pocket Reference The unique pocket reference shown in Figure 3 can operate for years on a pair of AAA alkaline cells or a single Lithium coin-cell, as the circuit draws just 10µA supply current. An input capacitor of 1µF as shown should be used when the LT6650 is operated from small batteries or other sources LT6650 Q20 IN Q13 Applications IN R7 Q18 Figure 2 shows the simplified schematic of the reference. Transistors Q1–Q7 form the band-gap-derived 400mV reference that is fed to the non-inverting input of the error amplifier formed by Q8–Q12. The resistors R1–R3 set the correct current flow into the internal reference, while R4 provides for post-package trimming capability. Transistors Q20 and Q21 form the rail-to-rail output stage and are driven by Q13–Q19. Resistors R5–R8 and the I2 current generator establish the gain and quiescent operating current of the output stage. In conjunction with the minimum recommended output capacitance of 1µF, stabilization is assured through Miller compensation inside error amplifier Q8–Q12. Pins are ESD protected by diodes D1–D3. 4 Q19 Q1 How it Works Inside CIN 1F Q2 be reduced to where the 1.2nA typical IBIAS becomes relatively significant loading, the relationship between the resistors then becomes: RF = RG • VOUT – 0.4 ( 0.4 – IBIAS • RG ) The minimum allowable gain resistor value is 2kΩ established by the 400mV FB pin voltage divided by the maximum guaranteed 200µA output current sourcing capability. In applications that scale the reference voltage, intrinsic noise is amplified along with the DC level. To minimize noise amplification, a 1nF feedback capacitor (CN) as shown in Figure 3 is recommended. Any net load capacitance of 1µF or higher assures amplifier stability. Automotive Reference In the presence of high supply noise, such as in automotive applications or DC-DC converters, an RC filter can be used on the VIN input as shown in Figure 4. Due to the exceptionally low supply current of the LT6650, the input resistor (RIN) of this filter can be 1kΩ or higher, depending on the difference in VIN and VOUT. Figure 5 shows supply rejection better than 30dB over a wide frequency spectrum, for a minimum sourcing output current of 40µA and an input filter comprising RIN = 1kΩ and CIN = 1µF. If even higher rejection is necessary, the input filter structure presented in Figure 6 effectively eliminates any supply transients continued on page 24 17 DESIGN IDEAS and lower their total solution cost. Smaller output capacitor values also speed up the changing of the output voltage when the CPU generates a different VID code. Other Features The LTC3738 has a differential amplifier for remote sensing of both the high and low sides of the output voltage. There is no reverse current during start-up, which allows the LTC3738 to power up into a pre-biased output without sinking current from the output. The LTC3738 also has a defeatable short-circuit shutdown timer. Three operation modes—PWM, pulse skip and Stage Shedding™—allow power supply designers to optimize for efficiency and noise. LT6650, continued from page 17 OUTPUT IMPEDANCE (Ω) 1000 IOUT = –40µA 100 CL = 10µF CL = 1µF NOISY POWER BUS IOUT = –40µA 10 CIN = 1µF RIN = 1k 0 33k –20 CL = 10µF –40 10 100 1k 10k FREQUENCY (Hz) 100k Figure 7. Output impedance is reduced while sourcing moderate current (40µA). 24 100k Figure 5. Improved supply noise rejection of Figure 4 reference circuit hundreds of ohms to the tens of ohms shown in Figure 7. Shunt-Mode Reference When the output voltage is tied to the input voltage, the high side of the rail-to-rail buffer amplifier is effectively disabled and only the low side remains active. In this mode of operation the LT6650 operates as a shunt reference as shown in Figure 8. Any shunt reference voltage from ±1.4V up to ±18V can be established by the feedback resistor selection. The noise and load capacitors have the same functions as in the series mode of operation. A 10µF minimum load capacitance is recommended for best stability and transient response. In shunt mode, an external biasing resistor RB is connected from 1nF 5 OUT GND 1k 10k FREQUENCY (Hz) Figure 6. High noise-immunity input network allows 50V transients on automotive power bus. –70 LT6650 100 22µF CL = 47µF –60 –80 VIN 1N751 5.1V CL = 1µF –50 4 10 1µF –30 IN 1 4.7k –10 CL = 47µF 10 LTC3738 is specifically designed to simplify power supply designs for Intel VRM9/VRM10 applications. It is a complete power supply solution with essenntial thermal management features, accurate load line control, precise output voltage sensing, and comprehensive fault protection. 20 POWER SUPPLY REJECTION RATIO (dB) from affecting the output by the inclusion of a pre-regulating Zener diode. With this extra input decoupling and the LT6650 circuitry operating from a 12V bus, 50V transients induce less than 0.5% VOUT perturbation. To obtain the micropower performance of the LT6650, quiescent currents of the internal circuitry are minute, which by nature, results in a higher output impedance than traditional references. Since output impedance is inversely related to the output stage operating current, a modest additional load current can easily reduce the output impedance by an order of magnitude from the unloaded case. Thus in applications where the output impedance and noise must be minimized, a light DC loading of the output provides enhanced performance. This loading can exist naturally in the application, or the feedback resistors can be designed to provide it. For example, setting the gain resistor value to 10kΩ establishes a moderate IOUT = –40µA and decreases the output peak resistance value from Conclusion 2 FB the power supply to the output, and delivers all the current required for supplying the LT6650 and the load current. RB is selected to ensure the operating current of the reference (IZ in the Figure 8 zener-diode analogy) is in the range of 30µA to 220µA under all loading conditions. Conclusion The LT6650 voltage reference incorporates a unique blend of low voltage, micropower operation and functional versatility. With the additional features of series and shunt mode configurability, source and sink output current, wide output voltage range, adjustability, and a tiny ThinSOT-23 package, the LT6650 provides an excellent solution to the many design challenges in both portable and industrial voltage control. CATHODE RF 1 10µF RG ANODE CATHODE = RB VS 1.4V VZ 18V 30µA IZ 220µA VZ = 0.4V • (1 + RF/RG) ANODE RB –VS Figure 8. Create you own adjustable micropower “zener” 2-terminal reference. Linear Technology Magazine • November 2004