LINEAR TECHNOLOGY NOVEMBER 2004 IN THIS ISSUE… COVER ARTICLE Versatile Op Amps Need No Resistors ...........................1 VOLUME XIV NUMBER 4 Versatile Op Amps Need No Resistors by Glen Brisebois and Jon Munson Glen Brisebois and Jon Munson Issue Highlights ............................ 2 LTC in the News… .......................... 2 DESIGN FEATURES 30V, Dual Output Regulator Controller is Efficient, Rich in Features, and Saves Space ............ 5 Teo Yang Long and Theo Phillips Dual Switcher with Spread Spectrum Reduces EMI ................. 9 Jason Leonard Superfast Fixed-Gain Triple Amplifiers Simplify Hi-Res Video Designs ................... 12 Jon Munson Power Supply Tracking for Linear Regulators ........................ 14 Dan Eddleman Tiny, Resistor-Programmable, µPower 0.4V to 18V Voltage Reference..................................... 16 Dan Serbanescu and Jon Munson Hot Swap for High Availability Systems ........... 18 David Soo DESIGN IDEAS ............................................... 22–36 (complete list on page 22) New Device Cameos...................... 37 Design Tools ................................ 39 Sales Offices................................ 40 Introduction What Do You Need: The LT1990, LT1991 and LT1995 High Precision, High Input are ready-to-use op amps with their Voltage or High Speed? own resistors and internal compensation capacitors. Many difference or instrumentation amps offer precisely matched internal components, but such devices are usually designed to solve a specific application problem, and thus have limited versatility. Not the LT1990, LT1991, and LT1995. These are flexible parts that can be configured into inverting, non-inverting, difference amplifiers, and even buffered attenuators, just by strapping their pins (Figure 1). The internal precisely matched resistors and capacitors make it possible to configure these op amps into hundreds of different application circuits without external components. Simply hook them up for type and gain and move on. By reducing the external components in your design, you simplify inventory, reduce pick and place costs, and make for easy probing. VIN– VIN+ VS+ 8 M9 9 M3 10 M1 1 P1 2 P3 3 P9 7 VCC LT1991 VEE OUT REF 5 6 VOUT 4 VS– DIFFERENCE GAIN = 11 Figure 1. What could be easier? A precision difference amplifier; no resistors in sight. This is only one of hundreds of possible configurations. See Figure 3 for a few more. Figure 2 shows simplified schematics of the three new amplifiers and in general, their comparative performance. The LT1990 is optimized for supporting high input common mode voltages of up to ±250V. The LT1991 is The LT1990, LT1991 and LT1995 are ready-to-use op amps with their own resistors and internal compensation capacitors. Just wire them up. optimized for gain flexibility and overall precision, and supports common mode ranges up to ±60V. The LT1995 is designed for high speed applications up to 30MHz. The LT1991 for the Greatest Flexibility and Precision The LT1991 is the most flexible and most precise of the three new devices. Its internal resistors guarantee 0.04% ratio-matching and 3ppm/°C MAX matching temperature coefficient. The op amp offers 15µV typical input offset voltage and 50pA of input offset current. The LT1991 operates on supplies from 2.7V to 36V with rail to rail outputs, and remains stable while driving capacitive loads up to 500pF. Gain bandwidth product is continued on page 3 , LTC, LT, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology Corporation. Adaptive Power, C-Load, DirectSense, FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products. DESIGN FEATURES LT1990, LT1991 and LT1995, continued from page 1 LT1990 ALL THIS IN AN SO-8 PACKAGE VCC 1pF LT1991 900k –IN +IN 1M GAIN1 1M + 40k M3 150k M1 450k 10k GAIN2 P1 450k P3 150k P9 50k 100k 1pF LT1995 450k 4pF – VOUT 900k 40k 50k 10k 100k – M9 ALL THIS IN AN MSOP-10 PACKAGE VCC M4 1k M2 2k M1 4k 4k 0.3pF – VOUT + 450k 4pF REF P1 4k P2 2k P4 1k VOUT + 4k 0.3pF REF VEE VEE VEE ALL THIS IN AN MSOP-10 PACKAGE VCC REF LT1990—HIGHEST VOLTAGE LT1991—MOST PRECISE, MOST FLEXIBLE LT1995—HIGH SPEED MORE VOLTAGE MORE SPEED Figure 2. The LT1990, LT1991, and LT1995 are ready-to-use op amps with their own resistors and internal compensation capacitors. Just wire them up. 560kHz, while drawing only 100µA supply current. The resistors are nominally 50k, 150k, and 450k. One end of each resistor is connected to an op amp input, and the other is brought out to a pin. The pins are named “M” or “P” depending on whether its resistor goes to the “minus” or “plus” input, and numbered “M1” M3” or “P9” etcetera VS + 8 M9 9 M3 10 M1 1 P1 2 P3 3 P9 according to the relative admittance of the resistor. So the “P9” pin has 9 times the admittance (or force) of the “P1” pin. The 450k resistors connected to the M1 and P1 inputs are not diode clamped, and can be taken well outside the supply rails, ±60V maximum. To use the LT1991, simply drive, ground or float the P, M, and REF inputs to set the configuration and gain. LT1991 VEE OUT REF 5 6 VOUT 1 P1 2 P3 3 P9 4 VS – VS VEE OUT REF 5 6 VOUT 1 P1 2 P3 3 P9 4 VIN– 7 VCC LT1991 OUT REF 5 6 VOUT = VS/2 4 VIN+ 1 P1 2 P3 3 P9 7 VCC LT1991 VEE OUT REF 5 VEE OUT REF 5 6 VOUT 4 6 VOUT DIFFERENCE GAIN = 11 5V 8 M9 9 M3 10 M1 VIN – 1 P1 2 P3 3 P9 VIN + 4 VS – MID-SUPPLY BUFFER LT1991 INVERTING GAIN = –3 VS+ 8 M9 9 M3 10 M1 7 VCC V S– GAIN = 14 VS VEE LT1991 VIN VIN GAIN = 5 1 P1 2 P3 3 P9 7 VCC 8 M9 9 M3 10 M1 VS – VIN 8 M9 9 M3 10 M1 VS+ VS+ 8 M9 9 M3 10 M1 7 VCC There is a whole series of high input common mode voltage circuits that can be created simply by just strapping the pins. Figure 3 demonstrates the flexibility of the LT1991 with just a few examples of different configurations and gains. In fact, there are over 300 unique achievable gains in the noninverting configuration alone. Gains –5V 7 VCC LT1991 VEE OUT REF 5 6 VOUT 4 –5V DIFFERENCE GAIN = 1 VCM = 1V TO 60V Figure 3. Non-inverting, inverting, difference amplifiers, and buffered attenuators achieved simply by connecting pins. This illustration shows only a small sample of the op amp configuration circuits possible with the LT1991, all without requiring external resistors. Linear Technology Magazine • November 2004 3 DESIGN FEATURES of up to 14 and buffered attenuations down to 0.07 are possible. 16 VCC The LT1990 for High Input Voltages 12 14 15 11 4-CELL BATTERY CELL VOLTAGE = 0.75V TO 1.7V The LT1990 has internal components with similar precision to the LT1991, but it is configured for high input voltages, up to ±250V. The high input voltage capability is achieved by the 1MΩ:40kΩ attenuation at the inputs, and by careful internal layout and shielding. The LT1990 has two effective gain settings, 1 and 10. A gain of 1 is set by floating the Gain1 and Gain2 pins, and a gain of 10 is set by shorting the Gain1 and Gain2 pins. Bandwidth is 100kHz in a gain of 1, and 6.5kHz in a gain of 10. The op amp operates on supplies from 2.7V to 36V with rail to rail outputs, on 105µA supply current. Like the LT1991, it remains stable while driving capacitive loads up to 500pF. 74HCD4052 1 5 2 4 6 EN EN 10 LSB LSB 8 M9 9 M3 10 M1 13 7 VCC LT1991 1 P1 2 P3 3 P9 3 8 GND 7 VEE MSB VEE OUT REF 5 6 VOUT 4 TRUTH TABLE MSB LSB VOUT 9 L L H H MSB L H L H 3 • V4 3 • V3 3 • V2 3 • V1 Figure 4. LT1991 applied as an individual battery cell monitor for a 4-cell battery LT1995 takes its pin names from the relative admittances of their resistors and the amplifier input polarity: hence “M1”, “M2”, “P4”, etcetera. For a difference gain of 6, short the M2 and M4 pins, and short the P2 and P4 pins (2 + 4 = 6). In this example, the difference amplifier is formed by the minus input of the shorted M2 and M4 inputs, and the plus input of the shorted P2 and P4 inputs. The LT1995 for High Speed The LT1995 offers high speed with 30MHz bandwidth, 24MHz full power bandwidth and 1000V/µs slew rate. It works on supplies from ±2.5V to ±15V drawing 7mA supply current. Accuracy is unusually good for a high speed amplifier, with input offset typically 600µV and guaranteed better than 4mV. It is pinned out identically to the LT1991, but with different resistor ratios and values. The resistors are lower impedance (1k, 2k, and 4k) than those in the LT1991 and LT1990 to support this device’s higher speed. They are as high a quality as you should ever need in a high speed application, guaranteeing 0.25% matching, worst case over temperature. As with the LT1991, the Applications Battery Monitor Many batteries are composed of individual cells with working voltages of about 1.2V each, as for example NiMH and NiCd. Higher total voltages are achieved by placing these in series. The reliability of the entire battery pack is limited by the weakest cell, so battery designers often like to maintain data on individual cell charge characteristics and histories. Figure 4 shows the LT1991 configured as a difference amplifier in a gain of 3, applied across the individual cells of a battery through a dual 4:1 mux. Because of the high valued 150kΩ resistors on its M3 and P3 inputs, the error introduced by the multiplexer switch ON-resistance is negligible. As the mux is stepped through its addresses, the LT1991 takes each cell voltage, multiplies it by 3, and references it to ground for easiest measurement. Note that worst case combinations of very different cell voltages can cause the LT1991 output to clip. Connecting the MSB line to the M1 and P1 inputs helps reduce the effect of the wide input common mode fluctuations from cell to cell. The low supply current of the LT1991 makes it particularly suited to battery powered applications. Its 110µA maximum supply current specification is about the same as that of the CMOS mux! Single-Supply Video Driver Most op amps operating from a single supply voltage require several highquality external resistors to generate a local bias voltage—to optimize the DC continued on page 35 0 5V –3 –6 + 47µF + VIN 1 P1 2 P2 3 P4 7 LT1995 5 4 75Ω 6 + 47µF 220µF 10k GAIN (dB) –9 8 M4 9 M2 10 M1 VOUT f–3dB = 27MHz RL = 75Ω –12 –15 –18 –21 –24 –27 1Hz 10Hz 100Hz 1kHz 10kHz 100kHz FREQUENCY 1MHz 10MHz 100MHz Figure 5. This single-supply composite video output-port driver requires no DC-biasing or gain-setting resistors 4 Linear Technology Magazine • November 2004 DESIGN IDEAS Optimizing for Efficiency While the LT3461A (boost) and LT3462A (inverting) are optimized for small size, the LT3461 (boost) and LT3462 (inverting) are intended for applications requiring higher efficiencies or high conversion ratios. The lower switching frequencies translate to higher efficiencies because of a reduction in switching losses. The LT3461 (boost) is guaranteed to a maximum switch duty cycle of 92% in continuous conduction mode, and the LT3462 (inverting) is guaranteed to a maximum switch duty cycle of 90%, which enables high conversion ratios at relatively high output currents. LTC2923, continued from page 15 rent produces an unacceptable output voltage error. Drivers for External, High Current Pass Devices Table 3 summarizes the characteristics of the LT1575 and LT3150 low dropout regulators. These devices drive external N-channel MOSFET pass devices for high current/high power applications. The LTC3150 Although high conversion ratios can also be obtained using discontinuous conduction mode (DCM)—where current in the inductor is allowed to go to zero each cycle—the DCM technique requires higher switch currents and larger inductors/rectifiers than a system operating in continuous conduction mode at the same load current. Because the LT3461 can switch at 1.3MHz in continuous conduction mode with up to 92% switch duty cycle, and the LT3462 at 1.2Mhz, 90% duty, they are the most compact solutions available for outputs 5 to 10 times the supply voltage. For example, the LCD bias circuit of Figure 7 provides additionally includes a boost regulator that generates gate drive for the external FET. The LTC2923 tracks the outputs of the LT1575 and LT3150 without any special modifications. Because these linear regulators only pull the FET’s gate down to about 2.6V, low-threshold FETs may not allow the output to fall below a few hundred millivolts. This is acceptable for most applications. 18mA at 25V from a 3.3V supply and occupies as little as 50mm2 of board space. Figure 8 shows that the efficiency of the 25V converter is quite good, peaking at 79% for a 4.2V supply. Figure 9 shows a 3.3V to –25V, 14mA inverter with efficiency above 70% (Figure 10). Conclusion The LT3461, LT3461A, LT3462 and LT3462A provide very compact boost and inverter solutions for a wide input voltage range of 2.5V to 16V, and outputs to ±38V, making these devices a good fit in a variety of applications. VIN IN OUT VOUT LT1963-1.5 SENSE GND VIN 1.5V R LTC2923 FBx R2 + 0.75V LT1006 – R1 R Table 3. Drivers for external, high current pass devices Regulator IOUT(MAX) (V) VIN(MIN) (V) VIN(MAX) (V) VDROPOUT (V) LT3150 10A* 1.4 10 0.13 LT1575 * N/A 22 * *Depends on selection of external MOSFET LT1990/91/95, continued from page 4 operating-point—and resistors to set gain. High quality resistors consume precious printed circuit board real estate, and test time. In contrast, the LT1995 provides on-chip resistors for voltage division and gain setting in a highly integrated video-speed op amp. Figure 5 shows a simple way to drive AC-coupled composite video signals over 75Ω coaxial cable using minimum component count. In this circuit, the input resistors form a supply splitter Linear Technology Magazine • November 2004 for biasing and a net attenuation of 0.75. The feedback configuration provides an AC-coupled gain of 2.66, so that the overall gain of the stage is 2.0. The output is AC-coupled and series back-terminated with 75Ω to provide a match into terminated video cable and an overall unity gain from signal input to the destination load. An output shunt resistor (10kΩ in this example) is always good practice in AC-coupled circuits to assure nominal biasing of the output coupling capacitor. Figure 6. Using an op amp with the LT1963-1.5 allows lower output voltages and removes error due to the SENSE pin current. Authors can be contacted at (408) 432-1900 Full Bridge Load Current Monitor Many new motor-drive circuits employ an H-bridge transistor configuration to provide bidirectional control from a single-voltage supply. The difficulty with this topology is that both motor leads “fly,” so current sensing becomes problematic. The LT1990 offers a simple solution to the problem by providing an integrated difference amp structure with an unusually high common-mode voltage rating, up to ±250VDC. 35