AND8124/D 90 W, Universal Input, Single Stage, PFC Converter http://onsemi.com General Description The NCP1651 demo board uses a quad operational amplifier on the secondary to perform multiple functions. One section of the amplifier is used as the error amplifier. A voltage divider comprised of R23, R24, R25 and R33 senses the output voltage and divides it down to 2.5 V. This signal is applied to the negative input of the error amplifier; the 2.5 V reference is applied to the non−inverting input of the error amplifier. The output of the error amplifier provides a current sink that drives the LED of the optocoupler. The primary side optocoupler circuit sinks current from pin 8. This varies the voltage into the Voltage−to−Current converter that feeds the reference multiplier. The loop operation is as follows: If the output voltage is less than its nominal value, the voltage at the output of the voltage divider (inverting input to the error amplifier) will be less than the reference signal at the non−inverting error amplifier input. This will cause the output of the error amplifier to increase. The increase in the output of the error amplifier will cause the optocoupler LED to conduct less current, which in turn will reduce the current in the optocoupler photo−transistor. This will increase the voltage at pin 8 of the chip, and in turn increase the output of the reference multiplier, causing an increase in the NCP1651 duty cycle. The current shaping network is comprised of the ac error amplifier, buffer and current sense amplifier. This network will force the average input current to maintain a scaled replica of the current reference on pin 10. The increase of the reference voltage will cause the current shaping network to draw more input current, which translates into an increase in output current as it passes through the transformer. The increase in current will increase the output power and therefore, the output voltage. To calculate the loop stability, it is recommended that the On Semiconductor spread sheet be used. This is an easy and convenient way to check the gain and phase of the control loop. This application note describes the implementation of a 90 W, universal input Flyback Power−Factor−Correction (PFC) converter using On Semiconductor’s NCP1651 controller. The NCP1651 enables a low cost single−stage (with a low voltage isolated output) PFC converter as demonstrated in this application circuit, which is designed for 48 Vdc, at 1.9 A of output current. The NCP1651 is designed to operate in the fixed frequency, continuous mode (CCM), or discontinuous (DCM) mode of operation, in a Flyback converter topology. The converter described in this application note has the following valuable features: Features • Wide Input Voltage Range (85 − 265 Vac) • Galvanic Isolation • Primary Side Cycle−by−Cycle and Average Current Limit • Secondary Side Power Limiting • High Voltage Start−up Circuit Detailed Circuit Description Operational description and design equations are contained in the NCP1651 Data Sheet. This application note addresses specific design issues related to this converter design. Please refer to Figure 2 for component reference designators. Voltage Regulation Loop With a Flyback topology, the output is isolated from the input by the power transformer. Output voltage regulation can be accomplished in two ways. The first, and the simplest method is by sensing the primary side voltage of the auxiliary winding. This eliminates the feedback isolation circuitry, at the expense of accuracy of voltage regulation and current sensing. The second method is to sense the secondary side voltage which is more complex, but provides better voltage regulation and transient response. Semiconductor Components Industries, LLC, 2003 December, 2003 − Rev. 4 1 Publication Order Number: AND8124/D J1 2 Input C26 1.2 F R3 180 k 2 R11 1.2 k NCP1651 16 Start−up 11 AC cmp 5 1st littr 8 Lavg 7 Ct 3 Ramp 4 GND 2 R35 4.7 k R8 680 http://onsemi.com C3 1 Out C16 2.2 F C6 VCC 10 ACref 12 13 Vref .68 F 9 ACin 8 FB/SD U1 0 R1 10 1.5kE68CA D16 1.5kE25CA D12 R34B 56 k 180 k R2 D4 1N4006 D3 1N4006 D7 1N4006 Q1 C25 1 nF R34A 56 k C11 .001 F 02 2N2222A 100 H L3 100 H 6 3 4 T3 o R5 .12 2 D6 MUR160 C13 .1 F C12 .1 F 470 pF R7 8.66 k 470 pF R4 35 k C10 1 nF C8 .022 F 1 R9 3.6 k C9 .022 F Figure 1. Applications Circuit Schematic .07 R25 2k 4 U3A 1 + − 13 12 11 + − 4 MC3303 U3D 14 R29 2.0 k C18 .047 F R22 C17 392 22 F 4 10 + U3C 8 9 − MC3303 R26 11 3.3 k 11 MC3303 − MC3303 11 C5 470 pF 4 U5B 5 7 + 6 2 3 C23 1500 F BAS19LT1 D10 C20 1 F C19 1 F 0 3 BAS19LT1 2 C4 1000 pF 4 1 D9 R21 2k R27 7.5 k R32 R30 300 C22 1500 F 0 D13 AZ23C18 U4 R20 2k Output R31 .07 C24 .01 F R24 174 D11 BAS19LT1 R23 210 R33 40.2 k D5 14 MUR1620CT 15 11 5 10 D9 BAS19LT1 7 C21 220 F TP2 GND R36 12 k 2 D2 1N4006 o o L2 J2 TP1 Shutdown 1 F1 C27 o D1 1N4006 AND8124/D C28 1 F 1 2 TL431 U2 R28 3.3 k AND8124/D Overshoot/Undershoot Circuit Two sections of the quad amplifier are used as comparators. One of these monitors the output for overvoltage condition and the other for undervoltage condition. The voltage divider requires four resistors (R33, R23, R24, and R25) in order to make the various ratios available for the two comparators as well as the error amplifier. The undervoltage comparator provides the drive for the opto−coupler. Its output is normally in the saturated high state, which allows the flow of current into the opto−coupler to be determined by the error amplifier or overvoltage comparator. If an undervoltage condition occurs, the output of the UV comparator goes low, which reduces the drive current to the opto−coupler LED. This causes the NCP1651 to go into a high duty cycle state, and will increase the flow of current into the output until the output voltage is above the UV limit. The over−voltage comparator’s output is OR’ed with the output of the error amplifier. During an overvoltage event (e.g. a transient load dump), the output of this comparator will go to ground, and cause the maximum current to flow in the opto−coupler LED. This will pull pin 8 low and reduce the duty cycle to zero until the output voltage is below the OV limit. It should be noted that the purpose of the 680 resistor (R8) in series with the opto−coupler photo transistor, is there to keep the voltage at pin 8 above the 0.5 V threshold during such events. This keeps the control chip operational and will allow immediate operation when the output voltage is again in its normal operating range. Without this resistor, the voltage on pin 8 would drop below 0.5 V, causing the NCP1651 to enter a low power shutdown mode of operation. in regulation the inverting input voltage is typically 2.5 V). This causes the error amplifier signal to go low, sinking more current through the LED in the opto−coupler. This in turn drives more current in opto−coupler transistor collector, pulling it low reducing the duty cycle, folding back the output voltage. Output Voltage Ripple The output voltage ripple on the secondary of the transformer has two components, the traditional high frequency ripple associated with a flyback converter, and the low frequency ripple associated with the line frequency (50 Hz or 60 Hz). In this application our goal was to have the output ripple 5% of the nominal output voltage, or 2.4 V pk−pk. The High Frequency Ripple can be Calculated by: (eq. 5) (toff 4T) (Ipk Iped )2)) (eq. 6) irms ((3.85 10 )) (((13.382 13.38 10.27 10.272) 3) 3.85 10 4) (13.38 10.27)2) 5.78 (eq. 7) To meet the capacitors ripple current requirements and lower the equivalent esr, two 1500 F capacitors were used in parallel. Vcap (5.78 3.85 3000 ) 0.00742 Where: n Ipk Iped CO esr T (eq. 1) (eq. 2) (eq. 8) =Transformer Turns Ratio (3.89) =Peak Current Secondary (13.38) =Pedestal Current Secondary (10.27) =Output Capacitance (1500 each) =Output Capacitor Equivalent Series Resistance (0.03 Each) =Switching Interval Vesr Ipksec esr The voltage to the input of the differential amplifier is: (eq. 9) Vesr 13.38 Apk 0.015 0.20 V (eq. 10) V 0.007422 0.22 0.200 (eq. 11) The Low Frequency Portion of the Ripple: The output voltage from the differential amplifier is: VO 0.33 11 3.63 V Vcap irms dt CO irms (toff T) (((Ipk2 (Ipk Iped) Iped2) 3 )) The fourth section of the amplifier is biased as a differential amplifier. This section senses the DC output current, and provides a signal that is diode OR’ed into the feedback divider. In the demo board the overload current limit was set to 125% of full load, or 2.375 A. Two resistors are used in series (to limit their maximum power dissipation) to sense the output current (R31 and R32). R29 and R30 set−the current sense amplifier gain. Where the gain of the amplifier is: 2.375 A 0.14 0.33 V (eq. 4) The RMS current at the peak of the sinewave (phase angle 90°). Current/Power Limit Circuit G (R29R30) 1 3000300 1 11 V Vcap2 Vesr2 (eq. 3) When the output load current increases, the output of the current sense amplifier will also increase. When the amplifiers output voltage, minus a diode drop (D11), increases above the 2.5 V, it pulls up the feedback signal at the inverting input of the error amplifier ( when the loop is V Ipk t CO (eq. 12) IAVG PO VO (eq. 13) Ipk IAVG 0.637 (eq. 14) Ipk PO VO 0.637 90 (48)(0.637) 2.95 (eq. 15) http://onsemi.com 3 AND8124/D If we divided the output ripple into 10° increments over one cycle (180°) the sinusoidal ripple voltage with respect to phase angle is: V (PO 0.637 VO) sin() CO 18 fline th = One Cycle of the Line 16.67ms (60Hz) Vmax = 48 V Vmin = 36 V Pout = 90 W (eq. 16) CO (2 90 16.67 ms) (482 362) 3000 F In Figure 2, the low frequency output voltage ripple are plotted with respect to phase angle. (eq. 19) It is a coincidence that the output capacitor calculated for voltage ripple and hold−up time are the same value. 1.50 MOSFET Turn−off Snubber RIPPLE (V) 1.00 The MOSFET in our design has a VDS rating of 800 V, the peak voltage across the device at turn−off (including the leakage inductance spike) is: 0.50 VpkTotal Vinmax 1.414 ((VO Vf)n) Vspike 0.00 (eq. 20) −0.50 Where: Vinmax VO n Vspike =265 Vrms =the Output Voltage (48 V) =the Transformer Turns Ratio (4) =Voltage Spike Due to Transformer Leakage Inductance To provide a safe operating voltage for the MOSFET we have selected Vspike to be 130 Vpeak, so when the MOSFET turns off, the maximum Drain to Source voltage is: −1.00 −1.50 0 45 90 DEGREES (°) 135 180 Figure 2. Calculated Output Ripple 265 1.414 48(4) 130 697 V Figure 3. Measured Output Voltage Ripple It can be seen from the calculations, and the scope waveform that as long as a capacitor with a low esr is used, that the output voltage ripple is dominated by the low frequency (120 Hz) ripple. E 1 le Ipk2 2 If the user would like to select CO for Hold−Up time versus, voltage ripple: E 1 C V2 2 (eq. 17) Where: C= Snubber Capacitor V= the Voltage Across the MOSFET Rearranging the equation: CO 2 Pout th V max 2 V min 2 (eq. 22) Where: le = Leakage Inductance (9 H Measured) Ipk = Peak Primary Current A Second Relationship is: Hold−Up time Pout 1 CO V2 f 2 (eq. 21) To minimize the effect of the leakage inductance spike, the coupling between the primary and secondary of the transformer needs to be as tight as possible. This can be accomplished, if your transformer requires a primary with multiple layers, by interleaving the primary and secondary windings. In our 48 Vdc application the transformer primary has 74 turns, and the secondary has 19 turns. The manufacture of the transformer, TDK, wound one layer of the primary with 45 turns, then the 19 turn secondary, and the remaining 29 turns of the primary. The results were a leakage inductance of approximately 9 H. If we compare this to a transformer where the entire 74 turns were wound, in two layers, then the 19 turn secondary, the leakage inductance increased to 37 H. The energy stored in the transformer leakage: (eq. 18) http://onsemi.com 4 (eq. 23) AND8124/D Combining Equations: In Figure 4, the output voltage drops to 40 Vdc, and recovers in less than 160 ms. In Figure 6 the input voltage was increased to 230 Vac, and the load was switched from 10% to 100% load. The output voltage now drops only to 44 Vdc, and recovers in approximately 50 ms. The significant improvement in transient response performance is attributed to an increase in the DC gain and loop bandwidth at high line. As the input ac line voltage increases the control loop DC gain (Refer to www.onsemi.com for a copy of the excel design spreadsheet for details) increases from 42 dB at 115 Vac to 62 dB at 230 Vac and the control loop bandwidth increases from 2 Hz to 8 Hz. The result is that at high line, there is an improvement in transient response, but because there is less attenuation of the output 120 Hz ripple, it results in an increase in the input Total Harmonic Distortion (THD). The system designers will need to trade off their overall system performance THD, Power Factor, and transient response to optimize the control loop to meet their requirements. C Ipk2 le ((VO Vf)n Vpk Vspike)2 (eq. 24) ((VO Vf)n Vpk)2 Csnubber 3.82 9 H ((192 375 130)2 (eq. 25) (192 375)2 790 pF During the MOSFET turn−off, the capacitor C25 is charge through the Diode D6. Prior to the next ton switching cycle the capacitor C25 must be fully discharged, so Rsnubber is selected to be: Rsnubber ((VO Vf)n Vinmax 1.414 Vspike) 0.63 (Vspike * Csnubber) (eq. 26) ((192 375 130)0.63(6.5 ) (130 * 790 pF) 28 k (eq. 27) The power in the snubber is: P 1 C V2 2 (0.5)790 pF(1302) 100 kHz 0.68 W (eq. 28) After installing the snubber in the NCP1651 Demo Board, and measuring the voltage spike, the snubber components where adjusted for maximum performance, C25 was increased to 1000 pF, and R34 was changed to 30 k. The difference between the measured and calculated value can be attributed to the PWB board layout, and other parasitic components. Evaluation Board Test Results The results from the NCP1651 Demo Board show that using a flyback topology for a PFC converter can provide a low input Total Harmonic Distortion (THD), a high input power factor, and excellent steady state output voltage regulation. The NCP1651 achieved a THD at 115 Vac input at full load of 3.12% with a PF of 0.998. The input THD to 6.8% THD at 230 Vac in, with a PF of 0.971. The steady state output voltage regulation from 85 Vac to 230 Vac, and no load to full load is less than 0.02%, with an output voltage ripple meeting our design goal of 2.4 Vpk−pk, measured 2.0 V pk−pk. Figure 4. Transient Response Figures 4 through 7 show the output transient response for the 90 W converter. The test conditions for each Figure are listed below: Table 1. Test Conditions Vin IO Figure 4 115 Vac 0.19 – 1.92 A Figure 5 115 Vac 1.92 – 0.19 A Figure 6 230 Vac 0.19 – 1.92 A Figure 7 230 Vac 1.92 – 0.19 A Figure 5. http://onsemi.com 5 AND8124/D Power Dissipation Estimates The NCP1651 Demo Board power dissipation (measured) at 115 Vrms, full load, is (106.27 – 47.95 •1.92) = 14.21 W. Following table provides the calculated and estimated power loss spread among different power train components. Components Pd average D1−D4 Input Rectifier 1.65 W Q1 MOSFET 4.1 W D5 Output rectifier 1.7 W T3 Flyback transformer 3.5 W (estimate) R34 Snubber resistor 0.84 W D12 Transient suppressor 2.0 W miscellaneous 0.41 W Figure 6. Total 14.20 W Demo Board Operating Instructions Connect an Ac source, 85 − 265 Vac, 47 − 64 Hz to the input terminals J1. Connect a load to the output terminals J2, the PWB is market +, for the positive output, − for the return. Turn on the ac source, and the NCP1651 will automatically start, providing 48 Vdc to the load. Shutdown Circuit The shutdown circuit will inhibit the operation of the power converter and put the NCP1651 into a low power shutdown mode. To activate this circuit, apply 5 V to the red test point, with the black jack being “ground”. Be aware that the black jack is actually hot as it is connected to the output of the input bridge rectifiers. An isolated 5 V supply should be used. If this circuit is not being used, it can be left open as there is enough resistance built in to the circuit to keep the transistor (Q2) in it’s off state. Figure 7. Table 2. Performance Data Regulation Line/Load No Load 45 W 90 W 85 Vrms 47.94 47.95 47.95 115 Vrm 47.94 47.95 47.95 230 Vrms 47.94 47.95 47.95 265 Vrms 47.94 47.94 47.95 http://onsemi.com 6 AND8124/D Table 3. Harmonics & Distortion 115 Vac 90 W 230 Vac 90 W V harmon A harm. % V harm A harm% 2nd 0.143 0.156 0.08 0.2 3rd 0.203 1.94 0.25 4.74 5th 0.13 0.6 0.12 2.88 7th 0.08 0.28 0.07 0.22 9th 0.04 0.19 0.09 0.76 11th 0.08 0.29 0.08 0.27 13th 0.16 0.32 0.06 0.33 15th 0.28 0.41 0.14 0.68 17th 0.4 0.41 0.28 0.95 19th 0.05 0.29 0.12 0.3 PF 0.998 0.971 THD(A) 3.12 6.8 Ifund 0.918 0.468 Table 4. Efficiency 85 Vrms 115 Vrms 230 Vrms 265 Vrms 1.5 1.52 1.51 1.59 Pin 109.42 106.27 105.35 105.25 Vo 47.95 47.95 47.95 47.95 Io 1.92 1.92 1.92 1.92 Efficiency 0.841 0.866 0.874 0.875 Pin @ No Load Table 5. Vendor Contact List Vendor U. S. Phone / Internet ON Semiconductor 1−800−282−9855 www.onsemi.com/ TDK 1−847−803−6100 www.component.tdk.com/ Vishay www.vishay.com/ Bussman (Cooper Ind.) 1−888−414−2645 www.cooperet.com/ Coiltronics (Cooper Ind.) 1−888−414−2645 www.cooperet.com/ Fairchild www.fairchildsemi.com/ Panasonic www.eddieray.com/panasonic/ Weidmuller www.weidmuller.com/ Keystone 1−800−221−5510 www.keyelco.com/ HH Smith 1−888−847−6484 www.hhsmith.com/ Aavid Thermalloy www.aavid.com/ http://onsemi.com 7 AND8124/D Table 6. NCP1651 Application Circuit Parts List (Specifications:, 90 W, 85 vac to 265 vac Input Range, 48 V Output) Ref Des Description Part Number Manufacturer C1 Cap, Ceramic, Chip, 1000 pF, 50 V VJ0603Y102KXAAT VISHAY C3 Cap, Ceramic, Chip, 470 pF, 50 V VJ0603Y471JXAAT VISHAY C5 Cap, Ceramic, Chip, 470 pF, 50 V VJ0603Y471JXAAT VISHAY C6 Cap, Ceramic, Chip, 470 pF, 50 V VJ0603Y471JXAAT VISHAY C8 Cap, Ceramic, Chip, .022 F, 50 V VJ0603Y223KXXAT VISHAY C9 Cap, Ceramic, Chip, 0.022 F, 50 V VJ0603Y223KXXAT VISHAY C10, C11 Cap, Ceramic, chip, 0.001 F, 50 V VJ0603Y102KXAAT VISHAY C12, C13 Cap, Ceramic, Chip, 0.1 F, 50 V VJ0606Y104KXXAT VISHAY C16 2.2 F, alum elect, 450 V (0.394dia x 0.492H) (.394dia x .492H) ECA−2WHG2R2 EKA00DC122P00 Panasonic (Digi – P5873) Vishay Sprague (20) C17 Cap, Ceramic, Chip, 22 F, 10 V C3225X5R0J226MT TDK C18 Cap, Ceramic, Chip, .047 F, 50 V VJ0603Y473KXXAT VISHAY C19 Cap, Ceramic, Chip, .01 F, 50 V VJ0603Y103KXAAT VJ0603Y103KXAAT C20 Cap, Ceramic, Chip, 1 F, 25 V C3216X7R1E105KT TDK C21 220 F, alum elect, 25 V ECA1EM331 Panasonic C22, 23 1800 F, alum elect, 63 V (2.2A rms min) 1500 F, alum elect, 63 V EEU−FC1J182 EKB00JL415J00 Panasonic (Digi – P11283) Vishay Sprague (20) C24 Cap, Ceramic, Chip, .01 F, 50 V VJ0603Y103KXAAT VISHAY C25 Cap,Ceramic, .001 F, 1 KV ECK−03A102KBP Panasonic C26 1.2 F, 275 vac, X cap F1778−512K2KCT0 VISHAY C27 Cap, polypropylene, .68 uF, 400 VDC MKP1841−468−405 Vishey − Sprague C28 Cap, Ceramic, Chip, 1 F, 25 V VJ1206V105ZXXAT VISHAY D1 – D4 Diode, Rectifier, 800 V, 1 A 1N4006 ON Semiconductor D5 Diode, Ultrafast, 200 V, 16 A MUR1620CT ON Semiconductor D6 Diode, Ultrafast, 600 V, 1 A MUR160 ON Semiconductor D7 Diode, Rectifier, 800 V, 1 A 1N4006 ON Semiconductor D8 – D11 Diode, Switching, 120 V, 200 mA, SOT−23 BAS19LT1 ON Semiconductor D12 TVS, 214 V, 5 W 1.5KE250A ON Semiconductor D13 Zener Diode, 18 V AZ23C18 VISHAY D16 Zener Diode, 68 V 1.5kE68CA ON Semiconductor F1 Fuse, 2 A, 250 Vac 1025TD2A Bussman L2 2.5 A sat, 100 H inductor, diff mode TSL1315−101K2R5 TDK L3 2.5 A sat, 100 H inductor, diff mode TSL1315−101K2R5 TDK Q1 FET, 11 a, 800 V, .45 , N−channel SPA11N80C3 Infineon Q2 Bipolar, npn, 30 V, SOT−23 MMBT2222ALT1 ON Semiconductor R1 Resistor, SMT1206, 10 CRCW1206100JRE4 Vishey R2 Resistor, Axial Lead, 180k, ¼ W CMF−55−180K00FKRE Vishey R3 Resistor, Axial Lead, 180k, ¼ W CMF−55−180K00FKRE Vishey R4 Resistor, SMT1206, 35k CRCW120635KOJNTA Vishey R5 Resistor, SMT, 0.12 , 1 W WSL2512 .12 1% Vishey Dale R7 Resistor, SMT1206, 8.66 k CRCW12068661F Vishey R8 Resistor, SMT1206, 680 CRCW12066800F Vishey http://onsemi.com 8 AND8124/D Table 6. NCP1651 Application Circuit Parts List (Specifications:, 90 W, 85 vac to 265 vac Input Range, 48 V Output) Ref Des Description Part Number Manufacturer R9 Resistor, axial lead, 3.6k, ¼ W CMF−55−3K600FKBF Vishey R11 Resistor, SMT1206, 1.2k CRC12061K20JNTA Vishey R20 Resistor, SMT1206, 2.0k CRC12062K00JNTA Vishey R21 Resistor, SMT1206, 2.0k CRC12062K00JNTA Vishey R22 Resistor, SMT1206, 392 CRC12052K10JNTA Vishey R23 Resistor, SMT1206, 210, 1% CRCW12062100F Vishey R24 Resistor, SMT1206, 174, 1% CRCW12061740F Vishey R25 Resistor, SMT1206, 2.05k, 1% CRCW12062051F Vishey R26 Resistor, SMT1206, 3.3k CRC12063K30JNTA Vishey R27 Resistor, SMT1206, 7.5k CRC12067K50JNTA Vishey R28 Resistor, SMT1206, 3.3k CRC12063K30JNTA Vishey R29 Resistor, SMT1206, 3.01k, 1% CRCW12063011F Vishey R30 Resistor, SMT1206, 301, 1% CRCW12063010F Vishey R31 1w, .07 resistor WSL251R0700FTB Vishey R32 1w, .07 resistor WSL251R0700FTB Vishey R33 Resistor, SMT1206, 40.2k, 1% CRCW120640022F Vishey R34 Resistor, axial lead, 20k, 2W R35 Resistor, SMT1206, 4.7k CRCW12064K70NTA Vishey R36 Resistor, SMT1206, 12k CRCW120612K0JNTA Vishey R37 Resistor, SMT1206, 100k CRCR1206100K0JNTA Vishey T1 Transformer, Flyback (Lp 1 mH) SRW42EC−U04H14 TDK U1 PFC Controller NCP1651 ON Semiconductor U2 2.5 V programmable ref, SOIC TL431ACD ON Semiconductor U3 Quad Op A MC3303D ON Semiconductor U4 Optocoupler, 1:1 CTR, 4 pin SFH615AA−X007 Vishay Hardware H1 Printed Circuit Board H2 Connector 171602 Weidmuller (Digi 281−1435−ND) H3 Connector 171602 Weidmuller (Digi 281−1435−ND) H4 Standoff, 4−40, alum, hex, .500 inches 8403 HH Smith (Newark 67F4111) H5 Standoff, 4−40, alum, hex, .500 inches 8403 HH Smith (Newark 67F4111) H6 Standoff, 4−40, alum, hex, .500 inches 8403 HH Smith (Newark 67F4111) H7 Standoff, 4−40, alum, hex, .500 inches 8403 HH Smith (Newark 67F4111) H8 Heatsink, TO−220 590302B03600 Aavid Thermalloy H9 Heatsink, TO−220 590302B03600 Aavid Thermalloy H10 Test point, red 5005 Keystone (Digi 5005K−ND) H11 Test point, black 5006 Keystone (Digi 5006K−ND) H12 Shoulder Washer 3049K−ND Digi−Key H13 Insulator 4672 Keystone http://onsemi.com 9 AND8124/D ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. “Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. 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