LM3406/06HV
1.5A Constant Current Buck Regulator for Driving High
Power LEDs
General Description
Features
The LM3406/06HV are monolithic switching regulators designed to deliver constant currents to high power LEDs. Ideal
for automotive, industrial, and general lighting applications,
they contain a high-side N-channel MOSFET switch with a
current limit of 2.0A (typical) for step-down (Buck) regulators.
Controlled on-time with true average current and an external
current sense resistor allow the converter output voltage to
adjust as needed to deliver a constant current to series and
series-parallel connected LED arrays of varying number and
type. LED dimming via pulse width modulation (PWM) is
achieved using a dedicated logic pin or by PWM of the power
input voltage. The product feature set is rounded out with lowpower shutdown and thermal shutdown protection.
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Integrated 2.0A MOSFET
VIN Range 6V to 42V (LM3406)
VIN Range 6V to 75V (LM3406HV)
True average output current control
1.7A Minimum Output Current Limit Over Temperature
Cycle-by-Cycle Current Limit
PWM Dimming with Dedicated Logic Input
PWM Dimming with Power Input Voltage
Simple Control Loop Compensation
Low Power Shutdown
Supports All-Ceramic Output Capacitors and Capacitorless Outputs
■ Thermal Shutdown Protection
■ eTSSOP-14 Package
Applications
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LED Driver
Constant Current Source
Automotive Lighting
General Illumination
Industrial Lighting
Typical Application
30020301
© 2009 National Semiconductor Corporation
300203
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LM3406/06HV 1.5A Constant Current Buck Regulator for Driving High Power LEDs
August 21, 2009
LM3406/LM3406HV
Connection Diagram
LM3406/06HV
30020302
14-Lead Exposed Pad Plastic TSSOP Package
NS Package Number MXA14A
Ordering Information
Order Number
Package Type
NSC Package Drawing
Supplied As
LM3406MH
95 units in anti-static rails
LM3406MHX
eTSSOP-14
LM3406HVMH
MXA14A
LM3406HVMHX
2500 units on tape and reel
95 units in anti-static rails
2500 units on tape and reel
Pin Descriptions
Pin(s)
Name
Description
Application Information
1,2
SW
Switch pin
3
BOOT
MOSFET drive bootstrap pin
4
NC
No Connect
5
VOUT
Output voltage sense pin
Connect this pin to the output node where the inductor and the first
LED's anode connect.
6
CS
Current sense feedback pin
Set the current through the LED array by connecting a resistor from
this pin to ground.
7
GND
Ground pin
8
DIM
Input for PWM dimming
9
COMP
Error amplifier output
Connect a 0.1 µF ceramic capacitor with X5R or X7R dielectric from
this pin to ground.
10
RON
On-time control pin
A resistor connected from this pin to VIN sets the regulator controlled
on-time.
11
VCC
Output of the internal 7V linear
regulator
Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor
with X5R or X7R dielectric.
12
VINS
Input voltage PWM dimming
comparator input
Connect this pin to the anode of the input diode to allow dimming by
PWM of the input voltage
13,14
VIN
Input voltage pin
Nominal operating input range for this pin is 6V to 42V (LM3406) or 6V
to 75V (LM3406HV).
DAP
DAP
Thermal Pad
Connect to ground. Place 4-6 vias from DAP to bottom layer ground
plane.
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Connect these pins to the output inductor and Schottky diode.
Connect a 22 nF ceramic capacitor from this pin to the SW pins.
No internal connection. Leave this pin unconnected.
Connect this pin to system ground.
Connect a logic-level PWM signal to this pin to enable/disable the
power MOSFET and reduce the average light output of the LED array.
Logic high = output on, logic low - output off.
2
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN to GND
-0.3V to 45V
(76V LM3406HV)
-0.3V to 45V
(76V LM3406HV)
-0.3V to 45V
(76V LM3406HV)
-0.3V to 59V
(90V LM3406HV)
-1.5V to 45V
(76V LM3406HV)
-0.3V to 45V
(76V LM3406HV)
-0.3V to 14V
-0.3V to 14V
-0.3V to 7V
VINS to GND
VOUT to GND
BOOT to GND
SW to GND
BOOT to VCC
BOOT to SW
VCC to GND
DIM to GND
LM3406/LM3406HV
COMP to GND
-0.3V to 7V
CS to GND
-0.3V to 7V
RON to GND
-0.3V to 7V
Junction Temperature
150°C
Storage Temp. Range
-65°C to 125°C
ESD Rating (Note 2)
2kV
Soldering Information
Lead Temperature (Soldering,
10sec)
260°C
Infrared/Convection Reflow (15sec)
235°C
Absolute Maximum Ratings
LM3406/LM3406HV (Note 1)
Operating Ratings
(Note 1)
VIN
Junction Temperature Range
6V to 42V
(75V LM3406HV)
−40°C to +125°C
Thermal Resistance θJA
(eTSSOP-14 Package)
(Note 4)
50°C/W
Electrical Characteristics
VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply
for TA = TJ = +25°C (Note 3). Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/
max specification limits are guaranteed by design, test, or statistical analysis.
LM3406/LM3406HV
Symbol
Parameter
Conditions
Min
Typ
187.5
200
Max
Units
210
mV
REGULATION COMPARATOR AND ERROR AMPLIFIER
VREF
CS Regulation Threshold
CS Decreasing, SW turns on
191.0
(Note 5)
210.0
(Note 5)
V0V
CS Over-voltage Threshold
CS Increasing, SW turns off
300
mV
ICS
CS Bias Current
CS = 0V
0.9
µA
IVOUT
VOUT Bias Current
VOUT = 24V
83
µA
ICOMP
COMP Pin Current
CS = 0V
25
µA
Gm-CS
Error Amplifier
Transconductance
150 mV < CS < 250 mV
145
µS
VSD-TH
Shutdown Threshold
RON Increasing
VSD-HYS
Shutdown Hysteresis
RON Decreasing
SHUTDOWN
0.3
0.7
1.05
40
V
mV
ON AND OFF TIMER
tOFF-MIN
Minimum Off-time
CS = 0V
tON
Programmed On-time
VIN = 24V, VO = 12V, RON = 200 kΩ
230
tON-MIN
Minimum On-time
800
1300
ns
1800
280
VINS COMPARATOR
VINS-TH
VINS Pin Threshold
VINS decreasing
70
%VIN
IIN-2WD
VINS Pin Input Current
VINS = 24V * 0.7
25
µA
INTERNAL REGULATOR
VCC-REG
VCC Regulated Output
0 mA < ICC < 5 mA
VIN-DO
VIN - VCC
ICC = 5 mA, 6.0V < VIN < 8.0V, Nonswitching
VCC-BP-TH
VCC Bypass Threshold
VIN Increasing
VCC-LIM
VCC Current Limit
VIN = 24V, VCC = 0V
3
6.4
7
300
4
7.4
V
mV
8.8
V
20
mA
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LM3406/LM3406HV
Symbol
Parameter
Conditions
Min
Typ
Max
Units
VCC-UV-TH
VCC Under-voltage Lock-out
Threshold
VCC Increasing
5.3
V
VCC-UV-HYS
VCC Under-voltage Lock-out
Hysteresis
VCC Decreasing
150
mV
IIN-OP
IIN Operating Current
Non-switching, CS = 0.5V
1.2
mA
IIN-SD
IIN Shutdown Current
RON = 0V
240
350
µA
2.1
2.7
A
CURRENT LIMIT
ILIM
Current Limit Threshold
1.7
DIM COMPARATOR
VIH
Logic High
DIM Increasing
VIL
Logic Low
DIM Decreasing
IDIM-PU
DIM Pull-up Current
DIM = 1.5V
V
2.2
0.8
80
V
µA
MOSFET AND DRIVER
RDS-ON
Buck Switch On Resistance
ISW = 200 mA, BOOT = 6.3V
VDR-UVLO
BOOT Under-voltage Lock-out
Threshold
BOOT–SW Increasing
VDR-HYS
BOOT Under-voltage Lock-out
Hysteresis
BOOT–SW Decreasing
1.7
0.37
0.75
Ω
2.9
4.3
V
370
mV
THERMAL SHUTDOWN
TSD
Thermal Shutdown Threshold
165
°C
TSD-HYS
Thermal Shutdown Hysteresis
25
°C
50
°C/W
THERMAL RESISTANCE
θJA
Junction to Ambient
eTSSOP-14 Package (Note 4)
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability
and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in
the Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and the device should not be
operated beyond such conditions.
Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 3: Typical values represent most likely parametric norms at the conditions specified and are not guaranteed.
Note 4: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1oz. copper on the top or bottom PCB layer.
Note 5: Specified with junction temperature from 0°C - 125°C.
Note 6: VIN = 24V, IF = 1A, TA = 25°C, and the load consists of three InGaN LEDs in series unless otherwise noted. See the Bill of Materials table at the end of
the datasheet.
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4
LM3406/LM3406HV
Typical Performance Characteristics
Efficiency Vs. Number of InGaN LEDs in Series
(Note 6)
Efficiency Vs. Output Current
(Note 6)
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VREF vs Temperature
VREF vs VIN, LM3406
30020335
30020336
Current Limit vs Temperature
VREF vs VIN, LM3406HV
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30020337
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LM3406/LM3406HV
Current Limit vs VIN, LM3406
Current Limit vs VIN, LM3406HV
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30020340
VCC vs VIN
VO-MAX vs VIN, LM3406
30020341
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30020342
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LM3406/LM3406HV
Block Diagram
30020303
CONTROLLED ON-TIME OVERVIEW
Figure 1 shows a simplified version of the feedback system
used to control the current through an array of LEDs. A differential voltage signal, VSNS, is created as the LED current
flows through the current setting resistor, RSNS. VSNS is fed
back by the CS pin, where it is integrated and compared
against an error amplifier-generated reference. The error amplifier is a transconductance (Gm) amplifier which adjusts the
voltage on COMP to maintain a 200 mV average at the CS
pin. The on-comparator turns on the power MOSFET when
VSNS falls below the reference created by the Gm amp. The
power MOSFET conducts for a controlled on-time, tON, set by
an external resistor, RON, the input voltage, VIN and the output
voltage, V O. On-time can be estimated by the following simplified equation (for the most accurate version of this expression see the Appendix):
Application Information
THEORY OF OPERATION
The LM3406 and LM3406HV are buck regulators with a wide
input voltage range, low voltage reference, and a fast output
enable/disable function. These features combine to make
them ideal for use as a constant current source for LEDs with
forward currents as high as 1.5A. The controlled on-time
(COT) architecture uses a comparator and a one-shot ontimer that varies inversely with input and output voltage instead of a fixed clock. The LM3406/06HV also employs an
integrator circuit that averages the output current. When the
converter runs in continuous conduction mode (CCM) the
controlled on-time maintains a constant switching frequency
over changes in both input and output voltage. These features
combine to give the LM3406/06HV an accurate output current, fast transient response, and constant switching frequency over a wide range of conditions.
At the conclusion of tON the power MOSFET turns off and
must remain off for a minimum of 230 ns. Once this tOFF-MIN
is complete the CS comparator compares the integrated
VSNS and reference again, waiting to begin the next cycle.
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LM3406/LM3406HV
buck regulator cannot provide an output voltage that is higher
than the minimum input voltage, and in pratice the maximum
output voltage of the LM3406/06HV is limited by the minimum
off-time as well. VO-MAX determines how many LEDs can be
driven in series. Referring to the illustration in Figure 1, output
voltage is calculated as:
VO-MAX = VIN-MIN x (1 - fSW x tOFF-MIN)
tOFF-MIN = 230 ns
Once VO-MAX has been calculated, the maximum number of
series LEDs, nMAX, can be calculated by the following espression and rounding down:
nMAX = VO-MAX / VF
VF = forward voltage of each LED
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At low switching frequency VO-MAX is higher, allowing the
LM3406/06HV to regulate output voltages that are nearly
equal to input voltage, and this can allow the system to drive
more LEDs in series. Low switching frequencies are not always desireable, however, because they require larger, more
expensive components.
FIGURE 1. Comparator and One-Shot
SWITCHING FREQUENCY
The LM3406/06HV does not contain a clock, however the ontime is modulated in proportion to both input voltage and
output voltage in order to maintain a relatively constant frequency. On-time tON, duty cycle D and switching frequency
fSW are related by the following expression:
CALCULATING OUTPUT VOLTAGE
Even though output current is the controlled parameter in LED
drivers, output voltage must still be calculated in order to design the complete circuit. Referring to the illustration in Figure
1, output voltage is calculated as:
fSW = D / tON
D = (VO + VD) / (VIN - VSW + VD)
VO = n x VF + VSNS
VD = Schottky diode (typically 0.5V)
VSNS = sense voltage of 200 mV, n = number of LEDs in series
VSW = IF x RDSON
MINIMUM ON-TIME
The minimum on-time for the LM3406/06HV is 280 ns (typical). One practical example of reaching the minimum on-time
is when dimming the LED light output with a power FET
placed in parallel to the LEDs. When the FET is on, the output
voltage drops to 200 mV. This results in a small duty cycle
and in most circuits requires an on-time that would be less
than 280 ns. In such a case the LM3406/06HV keeps the ontime at 280 ns and increases the off-time as much as needed,
which effectively reduces the switching frequency.
The LM3406/06HV regulators should be operated in continuous conduction mode (CCM), where inductor current stays
positive throughout the switching cycle. During steady-state
CCM operation, the converter maintains a constant switching
frequency that can be estimated using the following equation
(for the most accurate version, particularly for applications
that will have an input or output voltage of less than approximately 12V, see the Appendix):
HIGH VOLTAGE BIAS REGULATOR (VCC)
The LM3406/06HV contains an internal linear regulator with
a 7V output, connected between the VIN and the VCC pins.
The VCC pin should be bypassed to the GND pin with a 0.1
µF ceramic capacitor connected as close as possible to the
pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical)
and then regulates at 7V as VIN increases. The
LM3406/06HV comes out of UVLO and begins operating
when VCC crosses 5.3V. This is shown graphically in the
Typical Performance curves. Connecting an external supply
to VCC to power the gate drivers is not recommended, however it may be done if certain precautions are taken. Be sure
that the external supply will not violate any absolute maximum
condition and will at no point exceed the voltage applied to
the VIN pins.
SETTING LED CURRENT
LED current is set by the resistor RSNS, which can be determined using the following simple expression due to the output
averaging:
RSNS = 0.2 / IF
MAXIMUM NUMBER OF SERIES LEDS
LED driver designers often want to determine the highest
number of LEDs that can be driven by their circuits. The limit
on the maximum number of series LEDs is set by the highest
output voltage, VO-MAX, that the LED driver can provide. A
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INTERNAL MOSFET AND DRIVER
The LM3406/06HV features an internal power MOSFET as
well as a floating driver connected from the SW pin to the
8
Dimming frequency, fDIM, and duty cycle, DDIM, are limited by
the LED current rise time and fall time and the delay from
activation of the DIM pin to the response of the internal power
MOSFET. In general, fDIM should be at least one order of
magnitude lower than the steady state switching frequency in
order to prevent aliasing.
INPUT VOLTAGE COMPARATOR FOR PWM DIMMING
Adding an external input diode and using the internal VINS
comparator allows the LM3406/06HV to sense and respond
to dimming that is done by PWM of the input voltage. This
method is also referred to as "Two-Wire Dimming", and a typical application circuit is shown in Figure 2. If the VINS pin
voltage falls 70% below the VIN pin voltage, the
LM3406/06HV disables the internal power FET and shuts off
the current to the LED array. The support circuitry (driver,
bandgap, VCC) remains active in order to minimize the time
needed to the turn the LED back on when the VINS pin voltage rises and exceeds 70% of VIN. This minimizes the response time needed to turn the LED array back on.
FAST LOGIC PIN FOR PWM DIMMING
The DIM pin is a TTL compatible input for PWM dimming of
the LED. A logic low (below 0.8V) at DIM will disable the internal MOSFET and shut off the current flow to the LED array.
While the DIM pin is in a logic low state the support circuitry
(driver, bandgap, VCC) remains active in order to minimize
the time needed to turn the LED array back on when the DIM
pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up
current ensures that the LM3406/06HV is on when DIM pin is
open circuited, eliminating the need for a pull-up resistor.
30020304
FIGURE 2. Typical Application using Two-Wire Dimming
ming method maintains a continuous current through the
inductor, and therefore eliminates the biggest delay in turning
the LED(s) or and off. The trade-off with parallel FET dimming
is that more power is wasted while the FET is on, although in
most cases the power wasted is small compared to the power
dissipated in the LEDs. Parallel FET circuits should use no
output capacitance or a bare minimum for noise filtering in
order to minimize the slew rate of output voltage. Dimming
FET Q1 can be driven from a ground-referenced source because the source stays at 0.2V along with the CS pin.
PARALLEL MOSFET FOR HIGH-SPEED PWM DIMMING
For applications that require dimming at high frequency or
with wide dimming duty cycle range neither the VINS comparator or the DIM pin are capable of slewing the LED current
from 0 to the target level fast enough. For such applications
the LED current slew rate can by increased by shorting the
LED current with a N-MOSFET placed in parallel to the LED
or LED array, as shown in Figure 3. While the parallel FET is
on the output current flows through it, effectively reducing the
output voltage to equal the CS pin voltage of 0.2V. This dim-
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LM3406/LM3406HV
BOOT pin. Both rise time and fall time are 20 ns each (typical)
and the approximate gate charge is 9 nC. The high-side rail
for the driver circuitry uses a bootstrap circuit consisting of an
internal high-voltage diode and an external 22 nF capacitor,
CB. VCC charges CB through the internal diode while the power
MOSFET is off. When the MOSFET turns on, the internal
diode reverse biases. This creates a floating supply equal to
the VCC voltage minus the diode drop to drive the MOSFET
when its source voltage is equal to VIN.
LM3406/LM3406HV
30020327
FIGURE 3. Dimming with a Parallel FET
PEAK CURRENT LIMIT
The current limit comparator of the LM3406/06HV will engage
whenever the power MOSFET current (equal to the inductor
current while the MOSFET is on) exceeds 2.1A (typical). The
power MOSFET is disabled for a cool-down time that of approximately 100 µs. At the conclusion of this cool-down time
the system re-starts. If the current limit condition persists the
cycle of cool-down time and restarting will continue, creating
a low-power hiccup mode, minimizing thermal stress on the
LM3406/06HV and the external circuit components.
OUTPUT OPEN-CIRCUIT
The most common failure mode for power LEDs is a broken
bond wire, and the result is an output open-circuit. When this
happens the feedback path is disconnected, and the output
voltage will attempt to rise. In buck converters the output voltage can only rise as high as the input voltage, and the
minimum off-time requirement ensures that VO(MAX) is slightly
less than VIN. Figure 4 shows a method using a zener diode,
Z1, and zener limiting resistor, RZ, to limit output voltage to
the reverse breakdown voltage of Z1 plus 200 mV. The zener
diode reverse breakdown voltage, VZ, must be greater than
the maximum combined VF of all LEDs in the array. The maximum recommended value for RZ is 1 kΩ.
The output stage (SW and VOUT pins) of the LM3406/06HV
is capable of withstanding VO(MAX) indefinitely as long as the
output capacitor is rated to handle the full input voltage. When
an LED fails open-circuit and there is no output capacitor
present the surge in output voltage due to the collapsing magnetic field in the output inductor can exceed VIN and can
damage the LM3406/06HV IC. As an alternative to the zener
clamp method described previously, a diode can be connected from the output to the input of the regulator circuit that will
clamp the inductive surge to one VD above VIN.
Regardless of which protection method is used a resistance
in series with the VOUT pin, ROUT, is recommended to limit
the current in the event the VOUT pin is pulled below ground
when the LED circuit is reconnected. This can occur frequently if the lead lengths to the LEDs are long and the inductance
is significant. A resistor between 1 kΩ and 10 kΩ is recommended.
OVER-VOLTAGE/OVER-CURRENT COMPARATOR
The CS pin includes an output over-voltage/over-current
comparator that will disable the power MOSFET whenever
VSNS exceeds 300 mV. This threshold provides a hard limit
for the output current. Output current overshoot is limited to
300 mV / RSNS by this comparator during transients. The OVP/
OCP comparator limits the maximum ripple voltage at the CS
pin to 200 mVP-P.
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LM3406/LM3406HV
30020311
FIGURE 4. Two Methods of Output Open Circuit Protection
as long as the logic low voltage is below the over temperature
minimum threshold of 0.3V. Noise filter circuitry on the RON
pin can cause a few pulses with longer on-times than normal
after RON is grounded or released. In these cases the OVP/
OCP comparator will ensure that the peak inductor or LED
current does not exceed 300 mV / RSNS.
LOW POWER SHUTDOWN
The LM3406/06HV can be placed into a low power state (IINSD = 240 µA) by grounding the RON pin with a signal-level
MOSFET as shown in Figure 5 . Low power MOSFETs like
the 2N7000, 2N3904, or equivalent are recommended devices for putting the LM3406/06HV into low power shutdown.
Logic gates can also be used to shut down the LM3406/06HV
30020312
FIGURE 5. Low Power Shutdown
25°C hysteresis (both values typical). During thermal shutdown the MOSFET and driver are disabled.
THERMAL SHUTDOWN
Internal thermal shutdown circuitry is provided to protect the
IC in the event that the maximum junction temperature is exceeded. The threshold for thermal shutdown is 165°C with a
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LM3406/LM3406HV
BUCK CONVERTERS WITH OUTPUT CAPACITORS
A capacitor placed in parallel with the LED(s) can be used to
reduce the LED current ripple while keeping the same average current through both the inductor and the LED array. With
an output capacitor the output inductance can be lowered,
making the magnetics smaller and less expensive. Alternatively, the circuit could be run at lower frequency but keep the
same inductor value, improving the power efficiency. Both the
peak current limit and the OVP/OCP comparator still monitor
peak inductor current, placing a limit on how large ΔiL can be
even if ΔiF is made very small. Adding a capacitor that reduces ΔiF to well below the target provides headroom for
changes in inductance or VIN that might otherwise push the
peak LED ripple current too high.
Figure 6 shows the equivalent impedances presented to the
inductor current ripple when an output capacitor, CO, and its
equivalent series resistance (ESR) are placed in parallel with
the LED array. Note that ceramic capacitors have so little ESR
that it can be ignored. The entire inductor ripple current still
flows through RSNS to provide the required 25 mV of ripple
voltage for proper operation of the CS comparator.
Design Considerations
SWITCHING FREQUENCY
Switching frequency is selected based on the trade-offs between efficiency (better at low frequency), solution size/cost
(smaller at high frequency), and the range of output voltage
that can be regulated (wider at lower frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the LM3406/06HV can be programmed
for switching frequencies ranging from the 10’s of kHz to over
1 MHz. This on-time varies in proportion to both VIN and VO
in order to maintain first-order control over switching frequency, however in practice the switching frequency will shift in
response to large swings in VIN or VO. The maximum switching frequency is limited only by the minimum on-time and
minimum off-time requirements.
LED RIPPLE CURRENT
Selection of the ripple current, ΔiF, through the LED array is
similar to the selection of output ripple voltage in a standard
voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC output voltage, LED
manufacturers generally recommend values for ΔiF ranging
from ±5% to ±20% of IF. Higher LED ripple current allows the
use of smaller inductors, smaller output capacitors, or no output capacitors at all. Lower ripple current requires more output
inductance, higher switching frequency, or additional output
capacitance, and may be necessary for applications that are
not intended for human eyes, such as machine vision or industrial inspection.
BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS
The buck converter is unique among non-isolated topologies
because of the direct connection of the inductor to the load
during the entire switching cycle. By definition an inductor will
control the rate of change of current that flows through it, and
this control over current ripple forms the basis for component
selection in both voltage regulators and current regulators. A
current regulator such as the LED driver for which the
LM3406/06HV was designed focuses on the control of the
current through the load, not the voltage across it. A constant
current regulator is free of load current transients, and has no
need of output capacitance to supply the load and maintain
output voltage. Referring to the Typical Application circuit on
the front page of this datasheet, the inductor and LED can
form a single series chain, sharing the same current. When
no output capacitor is used, the same equations that govern
inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a controlled on-time converter such as
LM3406/06HV the ripple current is described by the following
expression:
30020314
FIGURE 6. LED and CO Ripple Current
To calculate the respective ripple currents the LED array is
represented as a dynamic resistance, rD. LED dynamic resistance is not always specified on the manufacturer’s
datasheet, but it can be calculated as the inverse slope of the
LED’s VF vs. IF curve. Note that dividing VF by IF will give an
incorrect value that is 5x to 10x too high. Total dynamic resistance for a string of n LEDs connected in series can be
calculated as the rD of one device multiplied by n. Inductor
ripple current is still calculated with the expression from Buck
Regulators without Output Capacitors. The following equations can then be used to estimate ΔiF when using a parallel
capacitor:
The triangle-wave inductor current ripple flows through RSNS
and produces a triangle-wave voltage at the CS pin. To provide good signal to noise ratio (SNR) the amplitude of CS pin
ripple voltage, ΔvCS, should be at least 25 mVP-P. ΔvCS is described by the following:
The calculation for ZC assumes that the shape of the inductor
ripple current is approximately sinusoidal.
Small values of CO that do not significantly reduce ΔiF can
also be used to control EMI generated by the switching action
of the LM3406/06HV. EMI reduction becomes more important
ΔvCS = ΔiF x RSNS
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12
INPUT CAPACITORS
Input capacitors at the VIN pin of the LM3406/06HV are selected using requirements for minimum capacitance and rms
ripple current. The input capacitors supply pulses of current
approximately equal to IF while the power MOSFET is on, and
are charged up by the input voltage while the power MOSFET
is off. All switching regulators have a negative input
impedance due to the decrease in input current as input voltage increases. This inverse proportionality of input current to
input voltage can cause oscillations (sometimes called ‘power
supply interaction’) if the magnitude of the negative input
impedance is greater the the input filter impedance. Minimum
capacitance can be selected by comparing the input
impedance to the converter’s negative resistance; however
this requires accurate calculation of the input voltage source
inductance and resistance, quantities which can be difficult to
determine. An alternative method to select the minimum input
capacitance, CIN(MIN), is to select the maximum input voltage
ripple which can be tolerated. This value, ΔvIN(MAX), is equal
to the change in voltage across CIN during the converter ontime, when CIN supplies the load current. CIN(MIN) can be
selected with the following:
RECIRCULATING DIODE
The LM3406/06HV is a non-synchronous buck regulator that
requires a recirculating diode D1 (see the Typical Application
circuit) to carrying the inductor current during the MOSFET
off-time. The most efficient choice for D1 is a Schottky diode
due to low forward drop and near-zero reverse recovery time.
D1 must be rated to handle the maximum input voltage plus
any switching node ringing when the MOSFET is on. In practice all switching converters have some ringing at the switching node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average current, ID, calculated as:
ID = (1 – D) x IF
This calculation should be done at the maximum expected
input voltage. The overall converter efficiency becomes more
dependent on the selection of D1 at low duty cycles, where
the recirculating diode carries the load current for an increasing percentage of the time. This power dissipation can be
calculating by checking the typical diode forward voltage,
VD, from the I-V curve on the product datasheet and then
multiplying it by ID. Diode datasheets will also provide a typical
junction-to-ambient thermal resistance, θJA, which can be
used to estimate the operating die temperature of the device.
Multiplying the power dissipation (PD = ID x VD) by θJA gives
the temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below the
operational maximum.
A good starting point for selection of CIN is to use an input
voltage ripple of 5% to 10% of VIN. A minimum input capacitance of 2x the CIN(MIN) value is recommended for all
LM3406/06HV circuits. To determine the rms current rating,
the following formula can be used:
Design Example 1
The first example circuit uses the LM3406 to create a flexible
LED driver capable of driving anywhere from one to five white
series-connected LEDs at a current of 1.5A ±5% from a regulated DC voltage input of 24V ±10%. In addition to the ±5%
tolerance specified for the average output current, the LED
ripple current must be controlled to 10%P-P of the DC value,
or 150 mAP-P. The typical forward voltage of each individual
LED at 1.5A is 3.9V, hence the output voltage ranges from
4.1V to 19.7V, adding in the 0.2V drop for current sensing. A
complete bill of materials can be found in Table 1 at the end
of this datasheet.
Ceramic capacitors are the best choice for the input to the
LM3406/06HV due to their high ripple current rating, low ESR,
low cost, and small size compared to other types. When selecting a ceramic capacitor, special attention must be paid to
the operating conditions of the application. Ceramic capacitors can lose one-half or more of their capacitance at their
rated DC voltage bias and also lose capacitance with extremes in temperature. A DC voltage rating equal to twice the
expected maximum input voltage is recommended. In addi-
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LM3406/LM3406HV
tion, the minimum quality dielectric which is suitable for
switching power supply inputs is X5R, while X7R or better is
preferred.
as the length of the connections between the LED and the
rest of the circuit increase.
LM3406/LM3406HV
30020318
FIGURE 7. Schematic for Design Example 1
tON(3 LEDs) = 1014 ns
tON(5 LEDs) = 1512 ns
RON and tON
A moderate switching frequency of 500 kHz will balance the
requirements of inductor size and overall power efficiency.
The LM3406 will allow some shift in switching frequency when
VO changes due to the number of LEDs in series, so the calculation for RON is done at the mid-point of three LEDs in
series, where VO = 11.8V. Note that the actual RON calculation
is done with the high accuracy expression listed in the Appendix.
OUTPUT INDUCTOR
Since an output capacitor will be used to filter some of the AC
ripple current, the inductor ripple current can be set higher
than the LED ripple current. A value of 40%P-P is typical in
many buck converters:
ΔiL = 0.4 x 1.5 = 0.6AP-P
With the target ripple current determined the inductance can
be chosen:
RON = 144 kΩ
The closest 1% tolerance resistor is 143 kΩ. The switching
frequency and on-time of the circuit should be checked for
one, three and five LEDs using the equations relating RON and
tON to fSW. As with the RON calculation, the actual fSW and
tON values have been calculated using the high accuracy expressions listed in the Appendix.
LMIN = [(24 – 11.8) x 1.01 x 10-6] / (0.6) = 20.5 µH
The closest standard inductor value is 22 µH. The average
current rating should be greater than 1.5A to prevent overheating in the inductor. Inductor current ripple should be
calculated for one, three and five LEDs:
ΔiL(1 LED) = [(24 - 4.1) x 5.28 x 10-7] / 22 x 10-6
= 478 mAP-P
fSW(1 LED) = 362 kHz
fSW(3 LEDs) = 504 kHz
ΔiL(3 LEDs) = [(24 - 11.8) x 1.01 x 10-6] / 22 x 10-6
= 560 mAP-P
fSW(5 LEDs) = 555 kHz
ΔiL(5 LEDs) = [(24 - 19.7) x 1.51 x 10-6] / 22 x 10-6
= 295 mAP-P
tON(1 LED) = 528 ns
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14
ZC = [0.15 / (0.478 - 0.15] x 0.35 = 0.114Ω
A ceramic capacitor will be used and the required capacitance
is selected based on the impedance at 362 kHz:
IL(PEAK) = IL + 0.5 x ΔiL(MAX)
CO = 1/(2 x π x 0.16 x 3.62 x 105) = 3.9 µF
IL(PEAK) = 1.5 + 0.5 x 0.56 = 1.78A
This calculation assumes that CO will be a ceramic capacitor,
and therefore impedance due to the equivalent series resistance (ESR) and equivalent series inductance (ESL) of of the
device is negligible. The closest 10% tolerance capacitor value is 4.7 µF. The capacitor used should be rated to 25V or
more and have an X7R dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 1206
case size. A typical value for ESR of 3 mΩ can be read from
the curve of impedance vs. frequency in the product
datasheet.
In order to prevent inductor saturation the inductor’s peak
current rating must be above 1.8A. A 22 µH off-the shelf inductor rated to 2.1A (peak) and 1.9A (average) with a DCR of
59 mΩ will be used.
USING AN OUTPUT CAPACITOR
This application does not require high frequency PWM dimming, allowing the use of an output capacitor to reduce the
size and cost of the output inductor while still meeting the 10%
P-P target for LED ripple current. To select the proper output
capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following:
RSNS
Using the expression for RSNS:
RSNS = 0.2 / IF
RSNS = 0.2 / 1.5 = 0.133Ω
Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.13Ω device is the closest value, and a 0.33W, 1210
size device will handle the power dissipation of 290 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% tolerance requirement. From the expression for average LED
current:
The dynamic resistance, rD,of one LED can be calculated by
taking the tangent line to the VF vs. IF curve in the LED
datasheet. Figure 8 shows an example rD calculation.
IF = 0.2 / 0.13 = 1.54A, 3% above the target current
INPUT CAPACITOR
Following the calculations from the Input Capacitor section,
ΔvIN(MAX) will be 24V x 2%P-P = 480 mV. The minimum required capacitance is calculated for the largest tON, corresponding to five LEDs:
CIN(MIN) = (1.5 x 1.5 x 10-6) / 0.48 = 4.7 µF
As with the output capacitor, this required value is low enough
to use a ceramic capacitor, and again the effective capacitance will be lower than the rated value with 24V across CIN.
Reviewing plots of %C vs. DC Bias for several capacitors reveals that a 4.7 µF, 1812-size capacitor in X7R rated to 50V
loses about 40% of its rated capacitance at 24V, hence two
such caps are needed.
Input rms current is high in buck regulators, and the worstcase is when the duty cycle is 50%. Duty cycle in a buck
regulator can be estimated as D = VO / VIN, and when this
converter drives three LEDs the duty cycle will be nearly 50%.
30020324
FIGURE 8. Calculating rD from the VF vs. IF Curve
Extending the tangent line to the ends of the plot yields values
for ΔVF and ΔIF of 0.7V and 2000 mA, respectively. Dynamic
resistance is then:
IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA
rD = ΔVF / ΔIF = 0.5V / 2A = 0.25Ω
Ripple current ratings for 1812 size ceramic capacitors are
typically higher than 2A, so two of them in parallel can tolerate
more than enough for this design.
The most filtering (and therefore the highest output capacitance) is needed when rD is lowest, which is when there is
only one LED. Inductor ripple current with one LED is 478
mAP-P. The required impedance of CO is calculated:
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LM3406/LM3406HV
The peak LED/inductor current is then estimated. This calculation uses the worst-case ripple current which occurs with
three LEDs.
LM3406/LM3406HV
RECIRCULATING DIODE
The input voltage of 24V ±5% requires Schottky diodes with
a reverse voltage rating greater than 30V. The next highest
standard voltage rating is 40V. Selecting a 40V rated diode
provides a large safety margin for the ringing of the switch
node and also makes cross-referencing of diodes from different vendors easier.
The next parameters to be determined are the forward current
rating and case size. The lower the duty cycle the more thermal stress is placed on the recirculating diode. When driving
one LED the duty cycle can be estimated as:
Gate charging and VCC loss, PG, in the gate drive and linear
regulator:
PG = (IIN-OP + fSW x QG) x VIN
PG = (600 x 10-6 + 5 x 105 x 9 x 10-9) x 24 = 122 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW
PS = 0.5 x 24 x 1.54 x 40 x 10-9 x 5 x 105 = 370 mW
D = 4.1 / 24 = 0.17
AC rms current loss, PCIN, in the input capacitor:
The estimated average diode current is then:
PCIN = IIN(rms)2 x ESR = 0.752 0.003 = 2 mW (negligible)
ID = (1 - 0.17) x 1.54 = 1.28A
DCR loss, PL, in the inductor
A 2A-rated diode will be used. To determine the proper case
size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD
for a case size such as SMB in a 40V, 2A Schottky diode at
1.5A is approximately 0.4V and the θJA is 75°C/W. Power dissipation and temperature rise can be calculated as:
PL = IF2 x DCR = 1.542 x 0.06 = 142 mW
Recirculating diode loss, PD = (1 - 0.5) x 1.54 x 0.4 = 300 mW
Current Sense Resistor Loss, PSNS = 293 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) =
18.2 / (18.2 + 2.1) = 89%
Temperature Rise in the LM3406 IC is calculated as:
PD = 1.28 x 0.4 = 512 mW
TRISE = 0.51 x 75 = 38°C
CB, CC AND CF
The bootstrap capacitor CB should always be a 22 nF ceramic
capacitors with X7R dielectric. A 25V rating is appropriate for
all application circuits. The COMP pin capacitor CC and the
linear regulator filter capacitor CF should always be 100 nF
ceramic capacitors, also with X7R dielectric and a 25V ratings.
TLM3406 = (PC + PG + PS) x θJA = (0.89 + 0.122 + 0.37) x 50 =
69°C
Design Example 2
The second example circuit uses the LM3406 to drive a single
white LED at 1.5A ±10% with a ripple current of 20%P-P in a
typical 12V automotive electrical system. The two-wire dimming function will be employed in order to take advantage of
the legacy 'theater dimming' method which dims and brightens the interior lights of automobiles by chopping the input
voltage with a 200Hz PWM signal. As with the previous example, the typical VF of a white LED is 3.9V, and with the
current sense voltage of 0.2V the total output voltage will be
4.1V. The LED driver must operate to specifications over an
input range of 9V to 16V as well as operating without suffering
damage at 28V for two minutes (the 'double battery jumpstart' test) and for 300 ms at 40V (the 'load-dump' test). The
LED driver must also be able to operate without suffering
damage at inputs as low as 6V to satisfy the 'cold crank' tests.
A complete bill of materials can be found in Table 2 at the end
of this datasheet.
EFFICIENCY
To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. Electrical efficiency, η, should not be confused
with the optical efficacy of the circuit, which depends upon the
LEDs themselves. One calculation will be detailed for three
LEDs in series, where VO = 11.8V, and these calculations can
be repeated for other numbers of LEDs.
Total output power, PO, is calculated as:
PO = IF x VO = 1.54 x 11.8 = 18.2W
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (1.542 x 0.75) x 0.5 = 890 mW
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16
LM3406/LM3406HV
30020325
FIGURE 9. Schematic for Design Example 2
tON(VMIN) = 1090 ns
RON and tON
A switching frequency of 450 kHz helps balance the requirements of inductor size and overall power efficiency, but more
importantly keeps the switching frequency below 530 kHz,
where the AM radio band begins. This design will concentrate
on meeting the switching frequency and LED current requirements over the nominal input range of 9V to 16V, and will then
check to ensure that the transient conditions do not cause the
LM3406 to overheat. The LM3406 will allow a small shift in
switching frequency when VIN changes, so the calculation for
RON is done at the typical expected condition where VIN =
13.8V and VO = 4.1V. The actual RON calculation uses the
high accuracy equation listed in the Appendix.
tON(VMAX) = 650 ns
OUTPUT INDUCTOR
Since an output capacitor will be used to filter some of the
LED ripple current, the inductor ripple current can be set higher than the LED ripple current. A value of 40%P-P is typical in
many buck converters:
ΔiL = 0.4 x 1.5 = 0.6AP-P
The minimum inductance required to ensure a ripple current
of 600 mAP-P or less is calculated at VIN-MAX:
RON = 124 kΩ
LMIN = [(16 – 4.1) x 6.5 x 10-7] / (0.6) = 12.9 µH
The closest 1% tolerance resistor is 124 kΩ. The switching
frequency and on-time of the circuit should be checked at
VIN-MIN and VIN-MAX which are 9V and 16V, respectively. The
actual fSW and tON values have been calculated with the high
accuracy equations in the Appendix.
The closest standard inductor value is 15 µH. The average
current rating should be greater than 1.5A to prevent overheating in the inductor. Inductor current ripple should be
calculated for VIN-MIN and VIN-MAX:
ΔiL(VMIN) = [(9 - 4.1) x 6.5 x 10-7] / 15 x 10-6
= 357 mAP-P
ΔiL(VMAX) = [(16 - 4.1) x 1.09 x 10-6] / 15 x 10-6
= 516 mAP-P
fSW(VMIN) = 463 kHz
fSW(VMAX) = 440 kHz
The peak LED/inductor current is then estimated. This calculation uses the worst-case ripple current which occurs at VINMAX.
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LM3406/LM3406HV
A ceramic capacitor will be used and the required capacitance
is selected based on the impedance at 440 kHz:
IL(PEAK) = IL + 0.5 x ΔiL(MAX)
CO = 1/(2 x π x 0.49 x 4.4 x 105) = 1.03 µF
IL(PEAK) = 1.5 + 0.5 x 0.516 = 1.76A
This calculation assumes that CO will be a ceramic capacitor,
and therefore impedance due to the equivalent series resistance (ESR) and equivalent series inductance (ESL) of of the
device is negligible. The closest 10% tolerance capacitor value is 1.5 µF. The capacitor used should have an X7R dielectric and should be rated to 50V. The high voltage rating
ensures that CO will not be damaged if the LED fails open
circuit and a load dump occurs. Several manufacturers produce ceramic capacitors with these specifications in the 1206
case size. With only 4V of DC bias a 50V rated ceramic capacitor will have better than 90% of it's rated capacitance,
which is more than enough for this design.
In order to prevent inductor saturation the inductor’s peak
current rating must be above 1.8A. A 15 µH off-the shelf inductor rated to 2.4A (peak) and 2.2A (average) with a DCR of
47 mΩ will be used.
USING AN OUTPUT CAPACITOR
This application does not require high frequency PWM dimming, allowing the use of an output capacitor to reduce the
size and cost of the output inductor while still meeting the 20%
P-P (300 mA) target for LED ripple current. To select the proper
output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following:
RSNS
Using the expression for RSNS:
RSNS = 0.2 / IF
RSNS = 0.2 / 1.5 = 0.133Ω
The dynamic resistance, rD,of one LED can be calculated by
taking the tangent line to the VF vs. IF curve in the LED
datasheet. Figure 8 shows an example rD calculation.
Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.13Ω device is the closest value, and a 0.33W, 1210
size device will handle the power dissipation of 290 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% tolerance requirement. From the expression for average LED
current:
IF = 0.2 / 0.13 = 1.54A, 3% above the target current
INPUT CAPACITOR
Controlling input ripple current and voltage is critical in automotive applications where stringent conducted electromagnetic interference tests are required. ΔvIN(MAX) will be limited
to 300 mVP-P or less. The minimum required capacitance is
calculated for the largest tON, 1090 ns, which occurs at the
minimum input voltage. Using the equations from the Input
Capacitors section:
CIN(MIN) = (1.5 x 1.09 x 10-6) / 0.3 = 5.5 µF
30020324
As with the output capacitor, this required value is low enough
to use a ceramic capacitor, and again the effective capacitance will be lower than the rated value with 16V across CIN.
Reviewing plots of %C vs. DC Bias for several capacitors reveals that a 3.3 µF, 1210-size capacitor in X7R rated to 50V
loses about 22% of its rated capacitance at 16V, hence two
such caps are needed.
Input rms current is high in buck regulators, and the worstcase is when the duty cycle is 50%. Duty cycle in a buck
regulator can be estimated as D = VO / VIN, and when VIN
drops to 9V the duty cycle will be nearly 50%.
FIGURE 10. Calculating rD from the VF vs. IF Curve
Extending the tangent line to the ends of the plot yields values
for ΔVF and ΔIF of 0.7V and 2000 mA, respectively. Dynamic
resistance is then:
rD = ΔVF / ΔIF = 0.5V / 2A = 0.25Ω
The most filtering (and therefore the highest output capacitance) is needed when ΔIL is highest, which occurs at VINMAX. Inductor ripple current with one LED is 516 mAP-P. The
required impedance of CO is calculated:
IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA
Ripple current ratings for 1210 size ceramic capacitors are
typically higher than 2A, so two of them in parallel can tolerate
more than enough for this design.
ZC = [0.3 / (0.516 - 0.3] x 0.35 = 0.35Ω
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18
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = 0.752 0.003 = 2 mW (negligible)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 1.542 x 0.05 = 120 mW
Recirculating diode loss, PD = (1 - 0.3) x 1.54 x 0.4 = 430 mW
Current Sense Resistor Loss, PSNS = 293 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) =
6.3 / (6.3 + 1.6) = 80%
Temperature Rise in the LM3406 IC is calculated as:
D = 4.1 / 13.8 = 0.3
The estimated average diode current is then:
ID = (1 - 0.3) x 1.54 = 1.1A
TLM3406 = (PC + PG + PS) x θJA = (0.53 + 0.06 + 0.19) x 50 =
39°C
A 2A-rated diode will be used. To determine the proper case
size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD
for a case size such as SMB in a 60V, 2A Schottky diode at
1.5A is approximately 0.4V and the θJA is 75°C/W. Power dissipation and temperature rise can be calculated as:
Thermal Considerations During
Input Transients
The error amplifier of the LM3406 ensures that average LED
current is controlled even at the transient load-dump voltage
of 40V, leaving thermal considerations as a primary design
consideration during high voltage transients. A review of the
operating conditions at an input of 40V is still useful to make
sure that the LM3406 die temperature is not exceeded.
Switching frequency drops to 325 kHz, the on-time drops to
350 ns, and the duty cycle drops to 0.12. Repeating the calculations for conduction, gate charging and switching loss
leads to a total internal loss of 731 mW, and hence a die temperature rise of 37°C. The LM3406 should operate properly
even if the ambient temperature is as high a 85°C.
PD = 1.1 x 0.4 = 440 mW
TRISE = 0.44 x 75 = 33°C
CB, CC AND CF
The bootstrap capacitor CB should always be a 22 nF ceramic
capacitors with X7R dielectric. A 25V rating is appropriate for
all application circuits. The COMP pin capacitor CC and the
linear regulator filter capacitor CF should always be 100 nF
ceramic capacitors, also with X7R dielectric and a 25V ratings.
Layout Considerations
EFFICIENCY
To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. One calculation will be detailed for the nominal
input voltage of 13.8V, and these calculations can be repeated for other numbers of LEDs.
Total output power, PO, is calculated as:
The performance of any switching converter depends as
much upon the layout of the PCB as the component selection.
The following guidelines will help the user design a circuit with
maximum rejection of outside EMI and minimum generation
of unwanted EMI.
COMPACT LAYOUT
Parasitic inductance can be reduced by keeping the power
path components close together and keeping the area of the
loops that high currents travel small. Short, thick traces or
copper pours (shapes) are best. In particular, the switch node
(where L1, D1, and the SW pin connect) should be just large
enough to connect all three components without excessive
heating from the current it carries. The LM3406/06HV operates in two distinct cycles whose high current paths are shown
in Figure 11:
PO = IF x VO = 1.54 x 4.1 = 6.3W
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (1.542 x 0.75) x 0.3 = 530 mW
Gate charging and VCC loss, PG, in the gate drive and linear
regulator:
PG = (IIN-OP + fSW x QG) x VIN
PG = (600 x 10-6 + 4.5 x 105 x 9 x 10-9) x 13.8 = 64 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW
PS = 0.5 x 13.8 x 1.54 x 40 x 10-9 x 4.5 x 105 = 190 mW
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LM3406/LM3406HV
RECIRCULATING DIODE
To survive an input voltage transient of 40V the Schottky
diode must be rated to a higher voltage. The next highest
standard voltage rating is 60V. Selecting a 60V rated diode
provides a large safety margin for the ringing of the switch
node and also makes cross-referencing of diodes from different vendors easier.
The next parameters to be determined are the forward current
rating and case size. The lower the duty cycle the more thermal stress is placed on the recirculating diode. When driving
one LED the duty cycle can be estimated as:
LM3406/LM3406HV
as small as possible to maximize noise rejection. RSNS should
therefore be placed as close as possible to the CS and GND
pins of the IC.
REMOTE LED ARRAYS
In some applications the LED or LED array can be far away
(several inches or more) from the LM3406/06HV, or on a separate PCB connected by a wiring harness. When an output
capacitor is used and the LED array is large or separated from
the rest of the converter, the output capacitor should be
placed close to the LEDs to reduce the effects of parasitic
inductance on the AC impedance of the capacitor. The current
sense resistor should remain on the same PCB, close to the
LM3406/06HV.
Remote LED arrays and high speed dimming with a parallel
FET must be treated with special care. The parallel dimming
FET should be placed on the same board and/or heatsink as
the LEDs to minimize the loop area between them, as the
switching of output current by the parallel FET produces a
pulsating current just like the switching action of the LM3406's
internal power FET and the Schottky diode. Figure 12 shows
the path that the inductor current takes through the LED or
through the dimming FET. To minimize the EMI from parallel
FET dimming the parasitic inductance of the loop formed by
the LED and the dimming FET (where only the dark grey arrows appear) should be reduced as much as possible. Parasitic inductance of a loop is mostly controlled by the loop area,
hence making this loop as physically small (short) as possible
will reduce the inductance.
30020326
FIGURE 11. Buck Converter Current Loops
The dark grey, inner loop represents the high current path
during the MOSFET on-time. The light grey, outer loop represents the high current path during the off-time.
GROUND PLANE AND SHAPE ROUTING
The diagram of Figure 11 is also useful for analyzing the flow
of continuous current vs. the flow of pulsating currents. The
circuit paths with current flow during both the on-time and offtime are considered to be continuous current, while those that
carry current during the on-time or off-time only are pulsating
currents. Preference in routing should be given to the pulsating current paths, as these are the portions of the circuit most
likely to emit EMI. The ground plane of a PCB is a conductor
and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the
ground net can be routed on the system ground plane with
less risk of injecting noise into other circuits. The path between the input source and the input capacitor and the path
between the recirculating diode and the LEDs/current sense
resistor are examples of continuous current paths. In contrast,
the path between the recirculating diode and the input capacitor carries a large pulsating current. This path should be
routed with a short, thick shape, preferably on the component
side of the PCB. Do not place any vias near the anode of
Schottky diode. Instead, multiple vias in parallel should be
used right at the pad of the input capacitor to connect the
component side shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop
formed by the SW and BOOT pins and capacitor CB. To minimize this loop and the EMI it generates, keep CB close to the
SW and BOOT pins.
30020328
FIGURE 12. Parallel FET Dimming Current Loops
CURRENT SENSING
The CS pin is a high-impedance input, and the loop created
by RSNS, RZ (if used), the CS pin and ground should be made
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20
ID
Part Number
Type
Size
Parameters
Qty
Vendor
U1
LM3406
LED Driver
L1
SLF10145T-220M1R-PF
Inductor
eTSSOP-14
42V, 2A
1
NSC
10 x 10 x 4.5mm
22 µH, 1.9A, 59 mΩ
1
D1
CMSH2-40
Schottky Diode
TDK
SMB
40V, 2A
1
Central Semi
Cc, Cf
VJ0603Y104KXXAT
Cb
VJ0603Y223KXXAT
Capacitor
0603
100 nF 10%
2
Vishay
Capacitor
0603
22 nF 10%
1
Cin1
Cin2
Vishay
C4532X7R1H475M
Capacitor
1812
4.7 µF, 50V
2
TDK
Co
C2012X7R1E105M
Capacitor
0805
1.0 µF, 25V
1
TDK
Rsns
ERJ14RQFR13V
Resistor
1210
0.13Ω 1%
1
Panasonic
Ron
CRCW08051433F
Resistor
0805
143 kΩ 1%
1
Vishay
TABLE 2. BOM for Design Example 2
ID
Part Number
Type
Size
Parameters
Qty
Vendor
U1
LM3406
LED Driver
eTSSOP-14
42V, 2A
1
NSC
L1
SLF10145T-150M2R2-P
Inductor
10 x 10 x 4.5mm
15 µH, 2.2A, 47 mΩ
1
TDK
D1
CMSH2-60
Schottky Diode
SMB
60V, 2A
1
Central Semi
Cc, Cf
VJ0603Y104KXXAT
Capacitor
0603
100 nF 10%
2
Vishay
Cb
VJ0603Y223KXXAT
Capacitor
0603
22 nF 10%
1
Vishay
Cin1
Cin2
C3225X7R1H335M
Capacitor
1210
3.3 µF, 50V
2
TDK
Co
C3216X7R1H105M
Capacitor
1206
0.15 µF, 50V
1
TDK
Rsns
ERJ14RQFR13V
Resistor
1210
0.13Ω 1%
1
Panasonic
Ron
CRCW08051243F
Resistor
0805
124 kΩ 1%
1
Vishay
Rpd
CRCW08051002F
Resistor
0805
10 kΩ 1%
1
Vishay
TABLE 3. Bill of Materials for Efficiency Curves
ID
Part Number
Type
Size
Parameters
Qty
Vendor
U1
LM3406
Buck LED
Driver
eTSSOP-14
42V, 1.5A
1
NSC
Q1
Si3458DV
N-MOSFET
SOT23-6
60V, 2.8A
1
Vishay
D1
CMSH2-60M
Schottky Diode
SMA
60V, 2A
1
Central Semi
L1
VLF10045T-330M2R3
Inductor
10 x 10 x 4.5mm
33 µH, 2.3A, 70 mΩ
1
TDK
Cin1 Cin2
C4532X7R1H685M
Capacitor
1812
6.8 µF, 50V
2
TDK
Co
C3216X7R1H474M
Capacitor
1206
470 nF, 50V
1
TDK
Cf ,Cc
VJ0603Y104KXXAT
Capacitor
0603
100 nF 10%
2
Vishay
Cb
VJ0603Y223KXXAT
Capacitor
0603
22 nF 10%
1
Vishay
R3.5
ERJ6RQFR56V
Resistor
0805
0.56Ω 1%
1
Panasonic
R.7
ERJ6RQFR62V
Resistor
0805
0.62Ω 1%
1
Panasonic
R1
ERJ6RQFR30V
Resistor
0805
0.3Ω 1%
1
Panasonic
R1.5
ERJ6RQFR16V
Resistor
0805
0.16Ω 1%
1
Panasonic
Ron
CRCW08051433F
Resistor
0805
143kΩ 1%
1
Vishay
Rpd Rout
CRCW06031002F
Resistor
0603
10 kΩ 1%
2
Vishay
OFF*
DIM1
DIM2
160-1512
Terminal
0.062"
3
Cambion
VIN GND
CS/LEDVo/LED+
160-1026
Terminal
0.094"
2
Cambion
21
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LM3406/LM3406HV
TABLE 1. BOM for Design Example 1
LM3406/LM3406HV
Appendix
The following expressions provide the best accuracy for users
who wish to create computer-based simulations or circuit calculators:
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22
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Notes
25
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LM3406/06HV 1.5A Constant Current Buck Regulator for Driving High Power LEDs
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