A 65 W Adaptor with NCP1239 Fixed Frequency Controller

AND9296/D
A 65 W Adaptor with
NCP1239 Fixed Frequency
Controller
The NCP1239 is a fixed-frequency current-mode
controller featuring a high-voltage start-up current source to
provide a quick and lossless power-on sequence.
With a supply range up to 35 V, the controller hosts
a jittered 65 or 100 kHz switching circuitry operated in peak
current mode control. When the power on the secondary side
starts to decrease, the controller automatically folds back its
switching frequency down to minimum level of 26 kHz. As
the power further goes down, the part enters skip cycle while
limiting the peak current that insures excellent efficiency in
light load condition.
It features a timer-based fault detection circuitry that
ensures a quasi-flat overload detection, independent of the
input voltage.
This application note focuses on the experimental results
of a 65 W adaptor driven by the NCP1239 and on the general
behavior of this controller.
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APPLICATION NOTE
Table 1. EVALUATION BOARD SPECIFICATION
Parameter
Value
Minimum Input Voltage
85 Vrms
Maximum Input Voltage
265 Vrms
Output Voltage
19 V
Nominal Output Power
65 W
Figure 1. EVB Picture (Top View)
Figure 2. EVB Picture (Bottom View)
© Semiconductor Components Industries, LLC, 2016
April, 2016 − Rev. 1
1
Publication Order Number:
AND9296/D
+
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2
Figure 3. Evaluation Board Schematic
C3
220nF
J1
F1
2A / 250V
L1
10mH / 2A
R21
1.5M
R23
1.5M
IN
IC1
KBU4K
C6
1n
85−265 V rms
R22
1.5M
−
C4
100u / 400V
R7
NTC
IC3x
OptoBase
D2
MRA4007
D1
MRA4007
C7
1n
R5
2.7k
5
4
C8
220p
R9
10
R3
22
Q1
BC857
R11
47k
C10
47u
D3
1N4937
D4
1N4937
R2
47k
D6
MMSD4148
R24
0R
R1
47k
R12
1
R10
0R
.
.
R13
1
ON Semiconductor
R14
1
SPA07N60
M1
750314896
T1
.
R15
33
C1
2.2nF
Type = Y1
C11
100p
IC3
OptoDiode
C18
1n
R25
10k
SFH6156−2
35V
35V
D8
BAV21
C14
680uF
C12
220p
C13
680uF
D7
MBR20H200
NCP1239 demoboard 19 V / 65 W
R8
1.6k C9
100n
IC2
NCP1239B65
6
8
3
2
1
D5
18V
R6
2.7k
C5
10n
L2
1u
IC4
NCP431
R16
1k
35V
C16
47n
R19
39k
R18
27k
Gnd
R20
10k
Gnd
C15
220uF
744772010
D9
RED
R26
33k
SGND
J2
19 V / 3.4 A
Gnd
Vout
AND9296/D
Board Schematic
AND9296/D
Start-up
between two cycles can be larger than 15 ms and VCC
voltage has to be kept above VCC(off). Finally, the last
constraints regarding the VCC capacitor is the start-up time.
Generally, the power supply has to start in less than 3 s.
Taking in account these parameters, in ours application
board, we have successfully tested (Figure 4) a 47 mF value
for C10.
The start-up sequence is performed with an internal high
voltage current source in order to reduce standby power
consumption. The start-up time is directly linked to the VCC
capacitor value. Also, this capacitor has to be large enough
to maintain the VCC voltage above VCC(off) level in no load
condition. Indeed, in light load or no load condition, the
controller enters in deep skip cycle mode and the dead time
vDRV (t)
270 ms
vCC (t)
Figure 4. The Start-up Sequence is below 3 s
The start-up sequence also involves the internal 8-ms
soft-start depicted in Figure 5. During this time, the peak
current setpoint is linearly increased from a very low value
up to the allowable maximum. This soft-start circuitry is
activated upon a fresh start-up but also every time a restart
is attempted, e.g. in an auto-recovery fault mode.
vGate(t)
8 ms
vCS (t)
Figure 5. The Soft-start Sequence
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AND9296/D
Protections
input voltage recovers a normal level before
initiate a new start-up sequence.
The NCP1239 embeds several needed protections
required by ac-dc adapters. They are listed below:
1. Short Circuit Protection, SCP: the adapter must
sustain a short circuit or an overload on the output
voltage without any damages. When short circuit
is removed, the power supply must be able to
restart and work normally.
2. Over Voltage Protection, OVP: when a
component on feedback loop like optocoupler is
damaged, the output voltage can dramatically
grow up and the controller must be turn off
immediately to protect devices that can be
connected to the adapter.
3. Over Temperature Protection, OTP: if the
temperature of the adapter exceeds a certain
ambient value, there is a risk of destruction.
To avoid this from happening, a thermal sensor
permanently monitors the temperature and in case
it exceeds the limit set by the designer, the adapter
shuts down permanently. The adapter is reset when
the user cycles the input power and the
temperature has decreased.
4. Over Power Protection, OPP: for some power
supplies, it is important that the maximum output
current stays in control in worse case conditions,
e.g. when the load is drawing more current that
what it should, without being a real short-circuit.
In our design, the nominal output current is 3.4 A
and must stay below 4.5 A in all input voltage
conditions.
5. Brown-out, BO: when the adapter is unplugged
or if there is a default on the main input, to avoid
damages when bulk voltage is too low,
the controller has to stop operation and waits the
Let us know check how each requirement has been
separately addressed.
Short Circuit Protection
The protection is ensured by monitoring the current sense
(CS) signal on pin 3. When this voltage exceeds the
maximum internal current setpoint (i.e. 0.8 V), an internal
error flag is raised and starts a timer. If the flag is asserted
longer than its programmed value (64 ms typical), the
driving pulses are stopped. The timer is reset if the CS
voltage goes back below the maximum current sense
threshold for 8 consecutive pulses. When the fault is
validated, the IC consumption is reduced to 500 mA. Thanks
to this consumption, VCC decreases and touches the 10 V
VCC(min) level. Here, the HV current source is activated to
build up the voltage to VCC(on) (12 V). At this moment,
depending of the controller option, there are two possible
configurations:
• Auto-recovery: when the 64 ms timer elapses,
the 1 s auto-recovery timer starts. If the 1 s timer is not
finished when VCC crosses VCC(on), HV current source
is disabled, controller stays off and VCC decays due to
IC consumption. Once auto-recovery timer elapses, the
controller initiates a new fresh sequence with soft-start
at next VCC(on) as shown in Figure 6 and Figure 7.
• Latching Off: when the 64 ms timer elapses,
the controller enters in endless hiccup mode meaning
that VCC will be charged and discharged between
VCC(on) and VCC(min) thanks to the HV current source
(Figure 8). The only ways to reset the controller and
have a new start-up sequence is to unplug to PSU
(VCC(reset) or BO even will be detected).
vDRV (t)
iOUT (t)
vFB (t)
64 ms
Figure 6. 64-ms Over-current Timer
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AND9296/D
vDRV (t)
vCC (t)
iOUT (t)
Figure 7. Auto-recovery Mode
vDRV (t)
vCC (t)
iOUT (t)
Figure 8. Latching Mode
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AND9296/D
Over Voltage Protection
perform this function, a Zener diode is usually connected
between the VCC pin (pin 5) and fault pin. The level is given
by VAUX(OVP) = VZ + VFault(OVP) where VAUX(OVP) is the
voltage on Auxilary winding during the off time and
VFault(OVP) is the 3-V threshold. Also, Auxiliary voltage is
linked to the output voltage with the transformer turns ratio:
VAUX = (NAux / NSec) * VOUT. We can deduct from these
equations the needed Zener diode value following the
wanted maximum output voltage in fault mode. Typical
waveforms are shown on Figure 9 and Figure 10.
When the optocoupler is broken or when the TL431
divider network is affected by a severe drift (or one of its
resistor is missing or features a wrong value), then the output
voltage can escape from the limits imposed by the
specifications: this is an over voltage condition. To protect
the converter, the controller has a dedicated fault pin (pin 1)
that combines the OVP detection and also the Over
Temperature Protection (see next section). The OVP
detection is made when the fault pin voltage exceeds 3 V
during 4 consecutives pulses, the controller is latched off. To
1
2
3
4
vDRV (t)
vOUT (t)
vFault(t)
Figure 9. 4 Consecutives Pulses to Validate the OVP Fault
vDRV (t)
VOUT(max) = 28.6 V
vOUT (t)
vFault(t)
Figure 10. OVP Event on Fault Pin
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AND9296/D
Finally, if the OVP function on the fault pin is not used,
this protection can be implemented on the VCC pin with
a fixed threshold (25.5 V). The level protected the controller
itself. The Figure 11 depicts this function. This protection
can be auto-recovery or latched depending of the controller
version.
vdrv(t)
25.5-V Threshold
vcc(t)
Figure 11. OVP Event on VCC Pin
Over Temperature Protection
the primary over pin 2 of the NCP1239. As detailed in the
datasheet, the current setpoint inside the circuit depends on
pin 2 level divided by 4. In fault conditions, when the loop
is lost, the feedback level can go up to 4.3 V. To avoid any
current runaway, the maximum voltage setpoint is safely
clamped to 0.8 V. In that case, the maximum peak current in
the inductor cannot exceed:
Due to the confined environment for adapter application,
a protection against run away temperature is highly
recommended. The fault pin has another lower threshold
(0.4 V) in order to connect a Negative Temperature
Coefficient resistance (NTC) with ground reference. In this
position, when the temperature increases, the NTC
resistance starts to decrease and lifts down the pin 1 voltage.
When the level reaches 0.4 V, the part simply latches off and
requires a reset before restart. Reset occurs when the user
cycles the input voltage.
Since we would like the adapter to enter over temperature
protection when ambient reaches 90°C, what will be the
needed pull down resistor the trigged the 0.4 V threshold?
Assuming the 45 mA internal OTP current source and 0.4 V
OTP detection level, at 90°C, NTC resistor should be .
R NTC_100 +
0.4 V
45 mA
I pk_max +
V Limit
R 12ńńR 13ńńR 14
(eq. 1)
With three paralleled 1 W resistances, we expect
a maximum peak current to be:
I pk_max + 0.8 + 2.4 A
0.33
(eq. 2)
The combination of two factors affects the maximum
output power delivery: the total propagation delay plays an
important role on the primary peak current and the operating
mode change between high line and low line.
The propagation delay tprop is the total time taken by the
control loop to bring the MOSFET gate down when the peak
current limit on CS pin (i.e. 0.8 V) has been reached.
+ 8.89 kW
Vishay NTC (NTCLE100E3104JB0W) matches pretty
well with the above calculation. The maximum ambient
temperature allowed by the demonstration board is around
87°C, so close to the expectation.
I pk_max +
Over Power Protection
V
V Limit
) bulk t prop
Lp
R 12ńńR 13ńńR 14
(eq. 3)
The control chip, alone, is rather fast: 50 ns typically.
However, the drive capability and the series drive resistance
naturally hamper the turn-off time. Typical total propagation
delays are therefore in the vicinity of 250−300 ns. Back to
Equation 3 and considering a rectified voltage Vbulk of
375 V dc (265 V rms input), the inductor peak current
becomes:
A current-mode power supply works by setting the
inductor peak current according to the output power
demand. The inductor current is transformed into a voltage
by a sense resistor, R12, R13 and R14 in our adapter. The peak
current setpoint depends on the error voltage delivered on
the feedback loop pin. In our adapter, this is the current
forced by the TL431 on the secondary side and reflected to
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AND9296/D
I pk_max + 0.8 ) 375 300n + 2.6 A
0.33 600m
obviously not acceptable and NCP1239 has a dedication
function to fight again this derivation.
NCP1239 senses the input voltage via HV pin. This line
voltage is transformed into a current information further
applied to the current sense pin. A resistor placed in series
from the sense resistance to the CS pin will create an offset
voltage proportional to the input voltage variation. Assume
we need to reduce the maximum peak current setpoint by
210 mV to reduce the maximum power at the 260 V input.
In that case, we will need to generate a 210 mV offset across
ROPP. With a 130 mA IOPP current, ROPP should be equal to:
(eq. 4)
This 200 mA difference represents a theoretical 15%
output power increase compared to the original calculation.
As said above, the other parameter that plays a role on the
maximum power delivery is the operating mode. At low
line, the power supply operates in deep Continuous
Conduction Mode (CCM) and the energy store in the
transformer is:
E p + 1 L pǒI pk_max * I valley
2
2
2
Ǔ
(eq. 5)
R OPP + 210m + 1.6 kW
130m
However, at high line, the peak current is indeed slightly
increased due to the propagation delay but because the
off-time has expanded, the valley current Ivalley is much
smaller than at low line: we are going into the Discontinuous
Conduction Mode (DCM). If Ivalley2 also goes down in
Equation 5, you naturally store more energy into the
inductor and the output power runs away. This situation is
(eq. 6)
With this OPP resistor, the over current limits from 85 V
rms to 265 V rms is between 3.9 A and 4.5 A. The maximum
output current evolution depending of the input voltage is
described in Figure 12.
4.5
4.4
IOCP (A)
4.3
4.2
4.1
4
3.9
3.8
85
100
115
130
145
160
175
190
205
220
235
250
265
VIN (V rms)
Figure 12. OCP Current vs Input Voltage
Brown-out Protection
Please note that different BO level options are available
upon request. Please contact sales to have more information.
There are two difference cases where BO event can be
detected. The first one is before start-up. The controller
starts to wake-up when VCC crosses VCC(min). At this
moment, the HV pin level is monitored. If, for any reason,
the input voltage is abnormally low, below VBO(on)
threshold, the controller do not turn-on the DRV pin and
VCC enters in hiccup mode until HV pin recovers a normal
level. This typical behavior is described on Figure 13.
The brown-out function is highly recommended to protect
the adapter against the low input voltage. Thanks to the HV
pin, we have an easy and no-consuming way to implement
this function. The brown-out thresholds are fixed:
• Line increasing, VBO(on): the controller is enable when
HV pin reaches 110 V dc
• Line decreasing, VBO(off): the controller is disable when
HV pin drops below 101 V dc
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AND9296/D
vDRV(t)
Wait the next VCC(on)
vCC (t)
vHV(t)
VHV < VBO(on)
Figure 13. BO Event before Start-up
The second case is when there is a line dropout. Assume
the converter operates normally. At a moment, the input
voltage drops below the VBO(off) level. A 68 ms timer starts.
During timer counting, the controller continues to work. If
the line comes back above VBO(on) level, the timer is reset
and PSU works normally. This behavior is depicted in
Figure 14. If BO timer elapses, DRV pulses are stopped, and
VCC enters in hiccup mode thanks to the HV current source.
In hiccup mode, when the line recovers its normal level, the
controller waits the next VCC(on) to initiate a fresh start-up
sequence with soft-start like shown in Figure 15.
vDRV(t)
vCC (t)
vHV(t)
tdrop-out < tBO
Figure 14. Line Drop-out Duration Shorter than BO Timer
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AND9296/D
vDRV(t)
vCC (t)
vHV(t)
Figure 15. BO Event During Operation
Efficiency Results
The output voltage and output current were measured
using digital multimeter embedded on dc electronic load
66103 from Chroma.
The average efficiency was calculated from the efficiency
measurements at 25%, 50%, 75% and 100% of the nominal
output power.
All measurements have been done after a 30 min warm-up
phase at full load and an additional 5 min at the load under
consideration.
The input power was measured with the power meter
66202 from Chroma.
Table 2. EFFICIENCY @ 115 V RMS AND 230 V RMS
Input Voltage
Pout (%)
Pout (W)
Pin (W)
Efficiency (%)
115 V rms
100
64.72
72.48
89.29
75
48.55
54.06
89.82
50
32.41
35.99
90.05
25
16.25
18.15
89.52
Average
−
−
89.67
No Load*
−
32 m
−
100
64.73
71.62
90.38
75
48.59
53.88
90.19
50
32.43
36.12
89.78
25
16.27
18.24
89.18
Average
−
−
89.88
No Load*
−
44 m
−
230 V rms
*Without the LED D9 and with 4.5 MW for X2 discharge resistor.
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AND9296/D
92
91
Efficiency (%)
90
89
88
87
115 V rms
86
230 V rms
85
0
10
20
30
40
50
60
70
80
90
100
Output Load (%)
Figure 16. Efficiency (%) vs Output Power (% of max) at 115 V rms and 230 V rms
Please note that the efficiency variation at 230 V rms
around 60−70% of the load is due to the DCM mode. Indeed,
if the MOSFET is turned on when the drain voltage is in the
valley, the efficiency will be better.
If we expand our view on the light-load power
consumption, in the range of 1 W output power, we can see
that we can deliver more than 0.78 W on the output and keep
the input consumption below 1 W.
1.4
1.3
115 V rms
1.2
230 V rms
1.1
1-W Input Power
Input Power (W)
1
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
Output Power (W)
Figure 17. Low Power Consumption
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0.8
0.78
0.9
1
AND9296/D
Stand-by Performance
Thanks to these two points, the standby consumption is
below 50 mW regardless the input voltage like shown in the
Table 3. Also, if we consider a LED connected on the output
voltage through a 33 kW resistance, the input power is still
below 60 mW @ 230 V rms.
The stand-by consumption is a key parameter for this kind
of application. Thanks to the HV startup current, the
resistance needed to build up the VCC voltage can be saved
and so the power dissipation. Moreover, the input power in
no load condition is highly impacted by the IC consumption
itself so this parameter has been optimized for the NCP1239
controller.
Table 3. STAND-BY CONSUMPTION
Input Voltage
Without LED D9
With LED D9
85 Vrms
30 mW
43 mW
115 Vrms
32 mW
44 mW
230 Vrms
44 mW
55 mW
265 Vrms
49 mW
61 mW
We can improve even more the standby performance by
playing with some components like the optocoupler or the
bridge divider on the NCP431 reference pin. These all
methods are explained and tested around the NCP1256
controller on the AND9208 application note.
the NCP1239 controller can operate in several modes. From
fixed frequency to skip mode passing by the frequency
foldback or frequency clamp mode, all these modes are
explained and illustrated in the following section. Also, the
NCP1239 operation can be illustrated versus the FB pin
voltage (Figure 18).
Typical Waveforms
The feedback voltage on the primary side is an image of
the load on the secondary side. Depending on the FB level,
Frequency
Peak Current Setpoint
FSW
VCS
V fold(end)
FB
max
65 kHz
max
0.8 V
[0.47 V
min
26 kHz
[0.25 V
skip
0.8 V
V skip
1.5 V
1.9 V
3.2 V
min
VFB
V fold
0.8 V
1.0 V
1.9 V
V skip
V freeze
V fold
3.2 V
V FB
Figure 18. By Observing the Voltage on the Feedback Pin, the Controller Reduces its Switching Frequency
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AND9296/D
Fixed Frequency Mode
decreased, the frequency will remain unchanged, 65 kHz
here, but the primary peak current will be reduced to transfer
less energy on the secondary side.
When the output load is close to the maximum, the
controller operates in fixed frequency mode. If the load
vFB(t)
VFB = 2.5 V
vCS (t)
vDrain(t)
65 kHz
Figure 19. Fixed Frequency Operation @ 65 W/140 V dc
vFB(t)
VFB = 2.0 V
vCS(t)
vDrain(t)
65 kHz
Figure 20. Fixed Frequency Operation @ 47 W/140 V dc
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Frequency Foldback mode
If while operating in fixed frequency, the load further
decreases, the NCP1239 will operate in Frequency Foldback
(FF) mode. Practically, the circuit enters in FF mode when
FB voltage drops below 1.9 V. In this mode, both frequency
and primary peak current vary according to the feedback
voltage as shown in Figure 21 and Figure 22.
VFB = 1.8 V
vFB(t)
vCS(t)
vDrain(t)
53 kHz
Figure 21. Frequency Foldback Operation @ 34 W/140 V dc
vFB(t)
VFB = 1.6 V
vCS(t)
vDrain(t)
34 kHz
Figure 22. Frequency Foldback Operation @ 18 W/140 V dc
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Frequency Clamp Mode
The switching frequency is clamped to 26 kHz in order to
avoid acoustic noise frequency range. The regulation is
made by varying the primary peak current (Ipeak reduces if
the power demand diminishes). This operation mode is
depicted at two different output powers in Figure 23 and
Figure 24.
vFB(t)
VFB = 1.45 V
vCS (t)
vDrain(t)
26 kHz
Figure 23. Frequency Clamp Mode @ 11.6 W/140 V dc
vFB(t)
VFB = 1.1 V
vCS(t)
vDrain(t)
26 kHz
Figure 24. Frequency Clamp Mode @ 6 W/140 V dc
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Skip Mode
typically), the power delivery cannot be continuously
controlled down to zero. Instead, the circuit stops pulsing
when the FB voltage drops below 800 mV and recovers
operation when VFB exceeds 830 mV (30 mV hysteresis).
Figure 25 shows controller operation this skip mode.
If the load continues to decrease and FB voltage drops
below 1 V, the primary peak current will be frozen to 31.25%
of its maximum value. Since the NCP1239 forces
a minimum peak current and a minimum frequency (26 kHz
vFB(t)
VFB ≈ 0.8 V
vCS(t)
vDrain(t)
Figure 25. Skip Cycle Mode in Light Load (3 W @ 140 V dc)
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Transient Load
The step load response is ±220 mV or ±1.2% of the output
voltage.
Figure 26 and Figure 27 show an output transient load
step from 10% to 100% of the maximum output power at low
line and high line. The slew rate is 1 A/ms and the frequency
is 20 Hz.
iOUT(t)
(1 A/div)
vOUT(t) - AC coupled
(200 mV/div)
Figure 26. Step Load Response between 10% to 100% @ 115 V rms
iOUT(t)
(1 A/div)
vOUT(t) - AC coupled
(100 mV/div)
Figure 27. Step Load Response between 10% to 100% @ 230 V rms
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AND9296/D
Table 4. BILL OF MATERIAL (BOM)
Designator
Quantity
Description
Value
Tolerance
Manufacturer
Part Number
C1
1
Y1 Capacitor, 250 V
2.2 nF
250 V
440LD22
C3
1
X2 Capacitor, 305 V
220 nF
305 V
B32922C3224M289
C4
1
Electrolytic Capacitor, 400 V
100 mF
400 V
400TXW100MEFC18X30
C5
1
Film Capacitor, 200 V
10 nF
200 V
Standard
C6, C7, C18
3
Ceramic Capacitor, SMD, 50 V
1 nF
10%, 50 V
Standard
C8
1
Ceramic Capacitor, SMD, 50 V
220 pF
10%, 50 V
Standard
C9
1
Ceramic Capacitor, SMD, 50 V
100 nF
10%, 50 V
Standard
C10
1
Electrolytic Capacitor, 35 V
47 mF
20%, 35 V
Standard
C11
1
Ceramic Capacitor, Axial, 1000 V
100 pF
10%, 1000 V
DEBB33A101KC1B
C12
1
Ceramic Capacitor, SMD, 50 V
220 pF
10%, 50 V
Standard
C13, C14
2
Electrolytic Capacitor, 35 V
680 mF
35 V
35ZL680M12.5X20
C15
1
Electrolytic Capacitor, 35 V
220 mF
35 V
Standard
C16
1
Ceramic Capacitor, SMD, 50 V
47 nF
10%, 50 V
Standard
D1, D2
2
Diode, Axial, 1 A, 1000 V
MRA4007
1 A, 1000 V,
SMA
MRA4007T3G
D3, D4
2
Fast Recovery Diode, Axial, 1 A, 600 V
D1N4937
1 A, 600 V,
DO−35
1N4937G
D5
1
18 V Zener Diode, Axial
Zener
18 V, DO−35
Standard
D6
1
Diode, SMD, 100 V
D1N4148
100 V
MMSD4148
D7
1
Schottky Diode, TO−220, 20 A, 150 V
MBR20H200
20 A, 200 V,
TO−220
MBR20200CTG
D8
1
Diode, Axial, 200 mA, 250 V
BAV21
200 mA,
250 V,
DO−35
Standard
D9
1
LED Rouge
HS1, HS2
2
Heatsink, 13°C/W, For M1 & D7
HSC1,
HSC2
2
Heatsink Clip for TO−220, For M1 & D7
IC1
1
Diode Bridge, 4 A, 800 V
KBU4K
KBU4K
IC2
1
QR Controller
NCP1239B65
NCP1239B65
IC3
1
Optocoupler SFH6156−2, SMD
SFH6156−2
SFH6156−2T
IC4
1
Shunt Regulator, 2.5−36 V, 1−100 mA
NCP431
NCP431AVSNT1G
F1
1
Fuse, 2 A, 250 V
2 A, 250 V
.0034.6618
J1
1
Input Connector, 2.5 A, 260 V
2.5 A, 260 V
JR−201S(PCB)
J2
1
Output Connector
10 A, 300 V
PM5.08/2/90
Jumper
1
L1
1
Common Mode Choke, 2*10 mH, 1.2 A
10 mH
1.2 A
RN114−1.2/02
L2
1
Radial Coil, 1 mH, 7.5 A, 20%
1 mH
7.5 A, 20%
744772010
M1
1
MOSFET, 650 V, 8 A
IPA65R190
8 A, 650 V
IPA65R190C7
Q1
1
PNP Transistor, SMD
BC857
R1, R2
2
Resistor, Axial, 3 W, 5%
47 kW
3 W, 5%
Standard
R3
1
Resistor, Axial, 1 W, 1%
22 W
1%
Standard
R5, R6
2
Ceramic Resistor, SMD, 0.25 W, 50 V
2.7 kW
5%
Standard
Standard
13°C/W
5901
www.onsemi.com
18
SW25−2
BC857ALT1G
AND9296/D
Table 4. BILL OF MATERIAL (BOM) (continued)
Designator
Quantity
Description
Value
Tolerance
Manufacturer
Part Number
R7
1
NTC, 100 kW at 25°C, Beta = 4190
100 kW
@ 25°C
0.05
NTCLE100E3104JB0
R8
1
Ceramic Resistor, SMD, 0.25 W, 200 V
1.6 kW
5%
Standard
R9
1
Ceramic Resistor, SMD, 0.25 W, 200 V
10 W
5%
Standard
R10, R24
2
Ceramic Resistor, SMD, 0.25 W, 200 V
0W
5%
Standard
R11
1
Ceramic Resistor, SMD, 0.25 W, 200 V
47 kW
5%
Standard
R12, R13,
R14
3
Ceramic Resistor, SMD, 1 W, 1%, 50 V
1W
1 W, 1%
Standard
R15
1
Ceramic Resistor, SMD, 0.25 W, 200 V
33 W
5%
Standard
R16
1
Ceramic Resistor, SMD, 0.25 W, 200 V
1 kW
5%
Standard
R18
1
Ceramic Resistor, SMD, 0.25 W, 50 V
27 kW
5%
Standard
R19
1
Ceramic Resistor, SMD, 0.25 W, 50 V
39 kW
5%
Standard
R20, R25
2
Ceramic Resistor, SMD, 0.25 W, 50 V
10 kW
5%
Standard
R21, R22,
R23
3
Ceramic Resistor, SMD, 0.25 W, 200 V
1.5 MW
5%
Standard
R26
1
Ceramic Resistor, SMD, 0.25 W, 50 V
33 kW
5%
Standard
S1, S3
2
Strap
400
Standard
S2, S4
2
Strap
700
Standard
SP1
1
Jumper400h
D3082F05
T1
1
Transformer, PQ26/25
750314896
TP2, TP3,
TP4, TP5,
TP6, TP7,
TP8, TP9,
TP10
9
Test Point
5010
X1, X2, X3,
X4
4
Support à riveter
SFCBS−M4−12M−01
Conclusion
This application note has described the results obtained
for a 65 W Fixed Frequency flyback topology with
NCP1239 controller. Thanks to the frequency foldback
mode, the middle and light load consumption have been
improved. The controller offers all necessary protections
needed to safe power supply.
The author wishes to thank Wurth Elektronik for kindly
providing samples for the transformer.
ON Semiconductor and the
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