dsPIC SMPS AC-DC Reference Design User Guide

SMPS AC/DC Reference Design
User’s Guide
© 2008 Microchip Technology Inc.
DS70320B
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Trademarks
The Microchip name and logo, the Microchip logo, Accuron,
dsPIC, KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro,
PICSTART, rfPIC, SmartShunt and UNI/O are registered
trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
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SEEVAL, SmartSensor and The Embedded Control Solutions
Company are registered trademarks of Microchip Technology
Incorporated in the U.S.A.
Analog-for-the-Digital Age, Application Maestro, CodeGuard,
dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN,
ECONOMONITOR, FanSense, In-Circuit Serial
Programming, ICSP, ICEPIC, Mindi, MiWi, MPASM, MPLAB
Certified logo, MPLIB, MPLINK, mTouch, PICkit, PICDEM,
PICDEM.net, PICtail, PIC32 logo, PowerCal, PowerInfo,
PowerMate, PowerTool, REAL ICE, rfLAB, Select Mode, Total
Endurance, WiperLock and ZENA are trademarks of
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countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
All other trademarks mentioned herein are property of their
respective companies.
© 2008, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
Printed on recycled paper.
Microchip received ISO/TS-16949:2002 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
DS70320B-page ii
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Table of Contents
Preface ........................................................................................................................... 1
Chapter 1. Introduction
1.1 System Specifications .................................................................................... 7
1.2 Block Diagram ................................................................................................ 8
1.3 Multi-Phase Synchronous Buck Converter ................................................... 19
1.4 Listing of I/O Signals for Each Block, Type of Signal and
Expected Signal Levels .......................................................................... 21
Chapter 2. Hardware Design
2.1 PFC Boost Converter ................................................................................... 25
2.2 Full-Bridge ZVT Converter ........................................................................... 30
2.3 Single-Phase Synchronous Buck Converter ................................................ 41
2.4 Three-Phase Synchronous Buck Converter ................................................. 43
2.5 Auxiliary Power Supply ................................................................................. 45
Chapter 3. Software Design
3.1 Overview ...................................................................................................... 51
3.2 Structure of the Control Software ................................................................. 52
3.3 Primary Side Control Software (PFC_ZVT) .................................................. 54
3.4 Secondary Side Control Software (DC_DC) ................................................ 61
3.5 Auxiliary Software Routines ......................................................................... 64
Chapter 4. System Operation
4.1 System Setup ............................................................................................... 67
4.2 System Operation ......................................................................................... 72
Appendix A. Board Layouts and Schematics
A.1 Introduction .................................................................................................. 75
A.2 SMPS AC/DC Reference Design Layout ..................................................... 75
A.3 SMPS AC/DC Reference Design Schematics ............................................. 77
© 2008 Microchip Technology Inc.
DS70320B-page iii
SMPS AC/DC Reference Design User’s Guide
Appendix B. Test Results
B.1 Soft-Start and Overshoot ............................................................................. 93
B.2 Dynamic Load Response ............................................................................. 95
B.3 Output Voltage Ripple .................................................................................. 99
B.4 Input Current .............................................................................................. 101
B.5 Efficiency .................................................................................................... 103
B.6 Input Current Total Harmonic Distortion (ITHD) ......................................... 103
B.7 Power Factor .............................................................................................. 103
B.8 Test Results Table ..................................................................................... 104
Appendix C. References
Index ...........................................................................................................................111
Worldwide Sales and Service ...................................................................................114
DS70320B-page iv
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Preface
NOTICE TO CUSTOMERS
All documentation becomes dated, and this manual is no exception. Microchip tools and
documentation are constantly evolving to meet customer needs, so some actual dialogs
and/or tool descriptions may differ from those in this document. Please refer to our web site
(www.microchip.com) to obtain the latest documentation available.
Documents are identified with a “DS” number. This number is located on the bottom of each
page, in front of the page number. The numbering convention for the DS number is
“DSXXXXXA”, where “XXXXX” is the document number and “A” is the revision level of the
document.
For the most up-to-date information on development tools, see the MPLAB® IDE on-line help.
Select the Help menu, and then Topics to open a list of available on-line help files.
INTRODUCTION
This chapter contains general information that will be useful to know before using the
SMPS AC/DC Reference Design. Items discussed in this chapter include:
•
•
•
•
•
•
•
•
Document Layout
Conventions Used in this Guide
Warranty Registration
Recommended Reading
The Microchip Web Site
Development Systems Customer Change Notification Service
Customer Support
Document Revision History
DOCUMENT LAYOUT
This document describes how to use the SMPS AC/DC Reference Design as a
development tool to emulate and debug firmware on a target board. The manual layout
is as follows:
• Chapter 1. “Introduction” – This chapter introduces the SMPS AC/DC
Reference Design and provides an overview of its features and background
information.
• Chapter 2. “Hardware Design” – This chapter provides a functional overview of
the SMPS AC/DC Reference Design and identifies the major hardware
components.
• Chapter 3. “Software Design” – This chapter provides a functional overview of
the software used in the hardware design and identifies the major software
components.
• Chapter 4. “System Operation” – This chapter provides information on the
system operation and setup for the SMPS AC/DC Reference Design.
© 2008 Microchip Technology Inc.
DS70320B-page 1
SMPS AC/DC Reference Design User’s Guide
• Appendix A. “Board Layouts and Schematics” – This appendix provides
detailed technical drawings and schematic diagrams of the SMPS AC/DC
Reference Design.
• Appendix B. “Test Results” – This appendix provides information on obtaining
the source code referenced in this document.
• Appendix C. “References” – This appendix provides detailed information on all
external references used throughout this document.
DS70320B-page 2
© 2008 Microchip Technology Inc.
Preface
CONVENTIONS USED IN THIS GUIDE
This manual uses the following documentation conventions:
DOCUMENTATION CONVENTIONS
Description
Arial font:
Italic characters
Initial caps
Quotes
Underlined, italic text with
right angle bracket
Bold characters
N‘Rnnnn
Text in angle brackets < >
Courier New font:
Plain Courier New
Represents
Referenced books
Emphasized text
A window
A dialog
A menu selection
A field name in a window or
dialog
A menu path
MPLAB® IDE User’s Guide
...is the only compiler...
the Output window
the Settings dialog
select Enable Programmer
“Save project before build”
A dialog button
A tab
A number in verilog format,
where N is the total number of
digits, R is the radix and n is a
digit.
A key on the keyboard
Click OK
Click the Power tab
4‘b0010, 2‘hF1
Italic Courier New
Sample source code
Filenames
File paths
Keywords
Command-line options
Bit values
Constants
A variable argument
Square brackets [ ]
Optional arguments
Curly brackets and pipe
character: { | }
Ellipses...
Choice of mutually exclusive
arguments; an OR selection
Replaces repeated text
Represents code supplied by
user
© 2008 Microchip Technology Inc.
Examples
File>Save
Press <Enter>, <F1>
#define START
autoexec.bat
c:\mcc18\h
_asm, _endasm, static
-Opa+, -Opa0, 1
0xFF, ‘A’
file.o, where file can be
any valid filename
mcc18 [options] file
[options]
errorlevel {0|1}
var_name [,
var_name...]
void main (void)
{ ...
}
DS70320B-page 3
SMPS AC/DC Reference Design User’s Guide
WARRANTY REGISTRATION
Please complete the enclosed Warranty Registration Card and mail it promptly.
Sending in the Warranty Registration Card entitles users to receive new product
updates. Interim software releases are available at the Microchip web site.
RECOMMENDED READING
This user's guide describes how to use SMPS AC/DC Reference Design. Other useful
documents are listed below. The following Microchip documents are available and
recommended as supplemental reference resources.
Readme Files
For the latest information on using other tools, read the tool-specific Readme files in
the Readmes subdirectory of the MPLAB® IDE installation directory. The Readme files
contain update information and known issues that may not be included in this user’s
guide.
Application Notes
The following related SMPS application notes are available for download from the
Microchip website:
• AN1106 “Power Factor Correction in Power Conversion Applications Using the
dsPIC® DSC” (DS01106)
• AN1114 “Switch Mode Power Supply (SMPS) Topologies (Part I)” (DS01114)
• AN1207 “Switch Mode Power Supply (SMPS) Topologies (Part II)” (DS01207)
THE MICROCHIP WEB SITE
Microchip provides online support via our web site at www.microchip.com. This web
site is used as a means to make files and information easily available to customers.
Accessible by using your favorite Internet browser, the web site contains the following
information:
• Product Support – Data sheets and errata, application notes and sample
programs, design resources, user’s guides and hardware support documents,
latest software releases and archived software
• General Technical Support – Frequently Asked Questions (FAQs), technical
support requests, online discussion groups, Microchip consultant program
member listing
• Business of Microchip – Product selector and ordering guides, latest Microchip
press releases, listing of seminars and events, listings of Microchip sales offices,
distributors and factory representatives
DS70320B-page 4
© 2008 Microchip Technology Inc.
Preface
DEVELOPMENT SYSTEMS CUSTOMER CHANGE NOTIFICATION SERVICE
Microchip’s customer notification service helps keep customers current on Microchip
products. Subscribers will receive e-mail notification whenever there are changes,
updates, revisions or errata related to a specified product family or development tool of
interest.
To register, access the Microchip web site at www.microchip.com, click on Customer
Change Notification and follow the registration instructions.
The Development Systems product group categories are:
• Compilers – The latest information on Microchip C compilers and other language
tools. These include the MPLAB C18 and MPLAB C30 C compilers; MPASM™
and MPLAB ASM30 assemblers; MPLINK™ and MPLAB LINK30 object linkers;
and MPLIB™ and MPLAB LIB30 object librarians.
• Emulators – The latest information on Microchip in-circuit emulators.This
includes the MPLAB ICE 2000 and MPLAB ICE 4000.
• In-Circuit Debuggers – The latest information on the Microchip in-circuit
debugger, MPLAB ICD 2.
• MPLAB® IDE – The latest information on Microchip MPLAB IDE, the Windows®
Integrated Development Environment for development systems tools. This list is
focused on the MPLAB IDE, MPLAB SIM simulator, MPLAB IDE Project Manager
and general editing and debugging features.
• Programmers – The latest information on Microchip programmers. These include
the MPLAB PM3 and PRO MATE II device programmers and the PICSTART®
Plus and PICkit™ 1 development programmers.
CUSTOMER SUPPORT
Users of Microchip products can receive assistance through several channels:
•
•
•
•
Distributor or Representative
Local Sales Office
Field Application Engineer (FAE)
Technical Support
Customers should contact their distributor, representative or field application engineer
(FAE) for support. Local sales offices are also available to help customers. A listing of
sales offices and locations is included in the back of this document.
Technical support is available through the web site at: http://support.microchip.com
DOCUMENT REVISION HISTORY
Revision A (February 2008)
Initial release of this document.
Revision B (November 2008)
This revision of the document includes the following updates:
• Extensive updates have been made throughout the document to reflect the
redesign of the software and the hardware to accommodate the 3.3V SMPS
dsPIC® DSC device (dsPIC33FJ16GS504)
• Modified contents of Appendix B. “Test Results” (previously named “Source
Code”)
• Updated all board layout diagrams and schematics in Appendix A. “Board
Layouts and Schematics”
© 2008 Microchip Technology Inc.
DS70320B-page 5
SMPS AC/DC Reference Design User’s Guide
NOTES:
DS70320B-page 6
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Chapter 1. Introduction
This chapter provides an introduction to the SMPS AC/DC Reference Design and
includes the following major topics:
•
•
•
•
1.1
System Specifications
Block Diagram
Multi-Phase Synchronous Buck Converter
Listing of I/O Signals for Each Block, Type of Signal and Expected Signal Levels
SYSTEM SPECIFICATIONS
This reference design describes the design of an off-line Switch Mode Power Supply
(SMPS) design using an SMPS dsPIC® DSC (dsPIC33FJ16GS504).
The SMPS AC/DC Reference Design works with universal input voltage range and
produces three output voltages (12V, 3.3V and 5V). The continuous output rating of the
reference design is 300 Watts. This reference design is based on a modular structure
having three major block sets as shown in Figure 1-1. Figure 1-2 shows a more
detailed block diagram with all functional blocks as implemented on the SMPS AC/DC
Reference Design.
The Power Factor Circuit (PFC) converts the universal AC input voltage to constant
high-voltage DC, and maintains the sinusoidal input current at high power factor. The
Phase-Shift Zero Voltage Transition circuit converts high-voltage DC to intermediate
low-voltage DC with isolation from the input AC mains, at high efficiency. The
Multi-Phase Synchronous and Single-Phase Synchronous Buck circuit converts
intermediate low-voltage DC to very low-voltage DC at high current at high efficiency.
The input and output specifications are as follows:
• Input:
- Input voltage: 85 VAC-265 VAC
- Input frequency: 45 Hz-65 Hz
• Outputs (individually loaded):
- Output voltage 1 (Vo1) = 12V
- Output load 1 (Io1) = 0A-30A
- Output voltage 2 (Vo2) = 3.3V
- Output load 2 (Io2) = 0A-69A
- Output voltage 3 (Vo3) = 5V
- Output load 3 (Io3) = 0A-23A
• Outputs (simultaneously loaded):
- Output voltage 2 (Vo2) = 3.3V
- Output load 2 (Io2) = 0A-56A
- Output voltage 3 (Vo3) = 5V
- Output load 3 (Io3) = 0A-23A
© 2008 Microchip Technology Inc.
DS70320B-page 7
SMPS AC/DC Reference Design User’s Guide
1.2
BLOCK DIAGRAM
A conventional SMPS must implement PFC if it draws more than 75 watts from the AC
Mains. The PFC circuitry draws input current in phase with the input voltage, and the
Total Harmonic Distortion (THD) of the input current should be less than 5% at full load.
The PFC provides a fixed DC high-output voltage, which needs to be converted to a
lower Direct Current (DC) output voltage and isolated with an input mains supply.
Figure 1-2 shows a high-level block diagram of the SMPS AC/DC Reference Design.
Figure 1-2 shows a detailed block diagram.
The SMPS AC/DC Reference Design operates on universal input voltage and
produces multiple DC output voltages. The front-end PFC Boost circuit converts
universal AC input voltage to 420 VDC bus voltage. The Phase-Shift Zero Voltage
Transition (ZVT) circuit produces 12 VDC output voltage from a 420 VDC bus. The
Phase-Shift ZVT converter also provides output voltage isolation from the input AC
mains. The Multi-Phase Synchronous Buck converter produces 3.3 VDC @ 69 Amps
from the 12 VDC bus. The Single-Phase Buck converter produces 5 VDC @ 23 Amps
from the 12 VDC bus.
The following sections in this chapter provide an overview and background of the main
power conversion blocks implemented in the SMPS AC/DC Reference Design.
FIGURE 1-1:
HIGH-LEVEL SMPS AC/DC REFERENCE DESIGN BLOCK DIAGRAM
AC Input
Voltage
85-265V
45-65 Hz
Input Stage
AC-DC Converter
FIGURE 1-2:
Intermediate Stage
DC-DC Converter
Point of Load
DC-DC Converter
Multiple
DC Outputs
DETAILED SMPS AC/DC REFERENCE DESIGN BLOCK DIAGRAM
12 VDC
30A
Isolation
Barrier
Rectified
Sinusoidal
Voltage
420 VDC
Phase-Shift ZVT Converter
EMI Filter
and Bridge
Rectifier
PFC
Boost
Converter
ZVT
Full-Bridge
Converter
Multi-Phase
Buck Converter
3.3 VDC
69A
Synchronous
Rectifier
85-265 VAC
45-65 Hz
dsPIC33FJ16GS504
DS70320B-page 8
Optocoupler
Single-Phase
Buck Converter
5 VDC
23A
dsPIC33FJ16GS504
© 2008 Microchip Technology Inc.
Introduction
1.2.1
Power Factor Correction (PFC)
Most power conversion applications consist of an AC-to-DC conversion stage
immediately following the AC source. The DC output obtained after rectification is
subsequently used for further stages. Current pulses with high peak amplitude are
drawn from a rectified voltage source with sine wave input and capacitive filtering.
Regardless of the load connected to the system, the current drawn is discontinuous
and of short duration. Because many applications demand a DC voltage source, a
rectifier with a capacitive filter is necessary. However, this results in discontinuous,
short duration current spikes.
1.2.1.1
OVERVIEW AND BACKGROUND INFORMATION
Two factors that provide a quantitative measure of the power quality in an electrical
system are Power Factor (PF) and Total Harmonic Distortion (THD). The amount of
useful power being consumed by an electrical system is predominantly decided by the
PF of the system.
To understand PF, it is important to know that power has two components:
• Working (or Active Power)
Working Power is the power that is actually consumed and registered on the
electric meter at the consumer's location. Working power is expressed in
kilowatts (kW), which register as kilowatt hour (kWh) on an electric meter.
• Reactive Power
Reactive Power is required to maintain and sustain the electromagnetic field associated with the industrial inductive loads such as induction motors driving pumps
or fans, welding machines and many more. Reactive Power is measured in kilovolt ampere reactive (kVAR) units. The total required power capacity, including
Working Power and Reactive Power, is known as Apparent Power, expressed in
kilovolt ampere (kVA) units.
Power Factor is a parameter that gives the amount of working power used by any
system in terms of the total apparent power. Power Factor becomes an important
measurable quantity because it often results in significant economic savings. Typical
waveforms of current with and without PFC are shown in Figure 1-3.
© 2008 Microchip Technology Inc.
DS70320B-page 9
SMPS AC/DC Reference Design User’s Guide
FIGURE 1-3:
INPUT CURRENT WAVEFORM WITH AND WITHOUT PFC
Input Voltage
0
DC Bus Output Voltage
Without
PFC
Input Current
0
DC Bus Output Voltage
With PFC
Input Current
These waveforms illustrate that PFC can improve the input current drawn from the
mains supply and reduce the DC bus voltage ripple. The objective of PFC is to make
the input to a power supply look like a simple resistor. The PFC circuitry provides a
power factor that is nearly equal to unity with very low current THD (< 5%).
Figure 1-4 shows a block diagram of the AC-to-DC converter stage, which converts the
AC input voltage to a DC voltage and maintains sinusoidal input current at a high input
Power Factor.
The input rectifier converts the alternating voltage at power frequency into
unidirectional voltage. This rectified voltage is fed to the chopper circuit to produce a
smooth and constant DC output voltage to the load. The chopper circuit is controlled
by the PWM switching pulses generated by the dsPIC DSC device, based on three
measured feedback signals:
• Rectified input voltage
• DC bus current
• DC bus voltage
DS70320B-page 10
© 2008 Microchip Technology Inc.
Introduction
FIGURE 1-4:
BLOCK DIAGRAM OF THE COMPONENTS FOR POWER FACTOR CORRECTION
Load
AC Input
Rectifier
Chopper
Switching pulses
Rectified Voltage
Bus Current
DC Voltage
dsPIC® Digital Signal Controller (DSC)
1.2.1.2
PFC TOPOLOGIES
The Power Factor can be achieved with various basic topologies such as Buck, Boost
and Buck/Boost.
1.2.1.2.1
Buck PFC Circuit
In a Buck PFC circuit, the output DC voltage is less than the input rectified voltage.
Large filters are needed to suppress switching ripples and this circuit produces
considerable Power Factor improvement. The switch (MOSFET) is rated to VIN in this
case. Figure 1-5 shows the circuit for the Buck PFC stage. Figure 1-6 shows the Buck
PFC input current shape.
FIGURE 1-5:
BUCK PFC
L
D
© 2008 Microchip Technology Inc.
+
Co
-
DS70320B-page 11
SMPS AC/DC Reference Design User’s Guide
FIGURE 1-6:
BUCK PFC INPUT CURRENT SHAPE
Input Voltage
t
Input Current
t
1.2.1.2.2
Boost PFC Circuit
The Boost converter produces a voltage higher than the input rectified voltage;
therefore, the switch (MOSFET) rating should be rated higher than VOUT. Figure 1-7
shows the circuit for the Boost PFC stage. Figure 1-8 shows the Boost PFC input
current shape.
FIGURE 1-7:
BOOST PFC
L
D
+
Co
-
FIGURE 1-8:
BOOST PFC INPUT CURRENT SHAPE
Input Voltage
t
Input Current
t
1.2.1.2.3
Buck/Boost PFC Circuit
In the Buck/Boost PFC circuit, the output DC voltage may be either less or greater than
the input rectified voltage. High Power Factor can be achieved in this case. The switch
(MOSFET) is rated to (VIN + VOUT). Figure 1-9 shows the circuit for the Buck/Boost
PFC stage. Figure 1-10 shows the Boost PFC input current shape.
DS70320B-page 12
© 2008 Microchip Technology Inc.
Introduction
FIGURE 1-9:
BUCK/BOOST PFC
D
L
FIGURE 1-10:
Co
+
BUCK/BOOST PFC INPUT CURRENT SHAPE
Input Voltage
t
Input Current
t
Regardless of the input line voltage and output load variations, input current drawn by
the Buck converter and the Buck/Boost converter is always discontinuous. However,
when the Boost converter operates in Continuous Conduction mode, the current drawn
from the input voltage source is always continuous and smooth as shown in Figure 1-8.
This feature makes the Boost converter an ideal choice for the Power Factor Correction
(PFC) application. In PFC, the input current drawn by the converter should be
continuous and smooth enough to meet Total Harmonic Distortion (THD) specifications
for the input current (ITHD) such that it is close to unity. In addition, input current should
follow the input sinusoidal voltage waveform to meet displacement factor such that it is
close to unity.
1.2.2
Phase-Shift ZVT Converter
A Full-Bridge converter is a transformer isolated Buck converter. The basic schematic
and switching waveform is shown in Figure 1-11. The transformer primary is connected
between the two legs formed by switches Q1,Q2 and Q4,Q3. Switches Q1,Q2 and Q4,
Q3 create a pulsating AC voltage at the transformer primary. The transformer is used
to step down the pulsating primary voltage, as well as to provide isolation between the
input voltage source and the output voltage VOUT. A Full-Bridge converter configuration
retains the voltage properties of the Half-Bridge topology, and the current properties of
push-pull topology. The diagonal switch pairs, Q1,Q3 and Q4,Q2, are switched
alternately at the selected switching period. Since the maximum voltage stress across
any switch is VIN, and with the complete utilization of magnetic core and copper, this
combination makes the Full-Bridge converter an ideal choice for high input voltage,
high-power range SMPS applications.
© 2008 Microchip Technology Inc.
DS70320B-page 13
SMPS AC/DC Reference Design User’s Guide
FIGURE 1-11:
FULL-BRIDGE CONVERTER
Q1
+
Q4
COSS1
COSS4
D3
+
L
VL
LLKG
(A)
- VOUT
CO
VIN
-
Q3
COSS2
COSS3
D4
Q2
TS
Q1PWM
Q3PWM
TON
TOFF
(B)
Q4PWM
Q2PWM
(C)
VIN
VIN
(D)
(A) = Full-Bridge/Half-Bridge Phase-Shift ZVT converter
(B) = PWM gate pulse waveform for Full-Bridge switches
(C) = Voltage across the transformer primary
(D) = Output inductor and rectifier diode current
In the Full-Bridge converter, four switches are used, thereby increasing the amount of
switching device loss. The conduction loss of a MOSFET can be reduced by using a
MOSFET with a low RDS(ON) rating. Switching losses can be reduced by using Zero
Voltage Transition (ZVT), Zero Current Switching (ZCS), or both techniques. At high
power output and high input voltage, the ZVT technique is preferred for the MOSFET.
In a Phase-Shift ZVT converter, the output is controlled by varying the phase of switch
Q4 with respect to Q1.
In this topology, the parasitic output capacitor of the MOSFETs, and the leakage
inductance of the switching transformer, are used as a resonant tank circuit to achieve
zero voltage across the MOSFET at the turn-on transition. There are two major
differences in the operation of a Phase-Shift ZVT and simple Full-Bridge topology. In a
Phase-Shift ZVT converter, the gate drive of both diagonal switches is phase shifted.
In addition, both halves of the bridge switch network are driven through the
complementary gate pulse with a fixed 50% duty cycle. The phase difference between
the two half-bridge switching network gate drives control the power flow from primary
to secondary, which results in the effective duty cycle.
DS70320B-page 14
© 2008 Microchip Technology Inc.
Introduction
Power is transferred to the secondary only when the diagonal switches are ON. If the
top or bottom switches of both legs are ON simultaneously, zero voltage is applied
across the transformer primary. Therefore, no power is transferred to the secondary
during this period. When the appropriate diagonal switch is turned OFF, primary current
flows through the output capacitor of the respective MOSFETs causing switch drain
voltage to move toward to the opposite input voltage rail. This creates zero voltage
across the MOSFET to be turned ON next, which creates zero voltage switching when
it turns ON. This is possible when enough circulating current is provided by the
inductive storage energy to charge and discharge the output capacitor of the respective
MOSFETs. Figure 1-12 shows the gate pulse required, and the voltage and current
waveform across the switch and transformer.
The operation of the Phase-Shift ZVT can be divided into different time intervals.
Assuming that the transformer was delivering the power to the load, the current flowing
through primary is IPK, and the diagonal switch Q1,Q3 was ON, at t = t0, the switch Q3
is turned OFF as shown in Figure 1-12.
FIGURE 1-12:
REQUIRED GATE PULSES AND VOLTAGE AND CURRENT ACROSS
PRIMARY
Q1PWM
Q2PWM
(A)
Q3PWM
Q4PWM
(B)
Vprimary
IPK
(C)
Ip
t0 t1 t2 t3 t4
(A) = Gate pulse for all switches for Phase-Shift ZVT converter
(B) = Voltage across primary
(C) = Current across primary
© 2008 Microchip Technology Inc.
DS70320B-page 15
SMPS AC/DC Reference Design User’s Guide
1.2.2.1
TIME INTERVALS
• Interval1: t0 < t < t1
Switch Q3 is turned OFF, beginning the resonant transition of the right leg. Primary current is maintained constant by the resonant inductor LLK. The primary
current charges the output capacitor of switch Q3 (COSS3) to the input voltage
VIN, which results in the output capacitance of Q4 (COSS4) being discharged to
zero potential. This creates zero potential across switch Q4 prior to turn-on, resulting in zero voltage switching. During this transition period, the transformer primary
voltage decreases from VIN to zero, and the primary no longer supplies power to
the output. Inductive energy stored in the output inductor, and zero voltage across
the primary, cause both output MOSFETs to share the load current equally.
• Interval2: t1 < t < t2
After charging COSS3 to VIN, the primary current starts flowing through the body
diode of Q4. Q4 can then be turned on any time after t1 and have a zero voltage
turn-on transition.
• Interval3: t2 < t < t3
At t = t2, Q1 was turned OFF and the primary was maintained by the resonant
inductor LLK. In addition, at t = t2, IP is slightly less than the primary peak current
IPK because of finite losses. The primary resonant current charges the output
capacitor of switch Q1 (COSS1) to input voltage VIN, which discharges the output
capacitor of Q2 (COSS2) to zero potential, thus preparing for zero voltage turn-on
for switch Q2. During this transition, the primary current decays to zero. ZVS of
the left leg switches depending on the energy stored in the resonant inductor,
conduction losses in the primary switches and the losses in the transformer
winding. Since the left leg transition depends on leakage energy stored in the
transformer, it may require an external series inductor if the stored leakage energy
is not enough for ZVS. When Q2 is then turned ON in the next interval, voltage VIN
is applied across the primary in the reverse direction.
• Interval: t3 < t < t4
The two diagonal switches Q4, Q2 are ON, applying full input voltage across the
primary. During this period, the magnetizing current, plus the reflected secondary
current into the primary, flows through the switch. The exact diagonal switch-on
time depends on the input voltage, the transformer turns ratio and the output
voltage. After the switch-on time period of the diagonal switch, Q4 is turned OFF.
One switching cycle is completed when the switch Q4 is turned OFF. The primary
current charges COSS4 to a potential of input voltage VIN, and discharges COSS3
to zero potential, thereby enabling ZVS for switch Q3. The identical analysis is
required for the next half cycle.
In the Phase-Shift ZVT converter shown in Figure 1-11, the maximum transition time
occurs for the left leg at minimum load current and maximum input voltage, and
minimum transition time occurs for the right leg at maximum load current and minimum
input voltage. Therefore, to achieve ZVT for all switches, enough inductive energy must
be stored to charge and discharge the output capacitance of the MOSFET in the
specified allocated time. Energy stored in the inductor must be greater than the
capacitive energy required for the transition. The MOSFET output capacitance varies
as applied drain-to-source voltage varies. Thus, the output capacitance of the
MOSFET should be multiplied by a factor of 4/3 to calculate the equivalent output
capacitance.
DS70320B-page 16
© 2008 Microchip Technology Inc.
Introduction
1.2.3
Buck Converter Description and Background
A Buck converter, as its name implies, can only produce lower average output voltage
than the input voltage. The basic schematic of a Buck converter is shown in
Figure 1-13. The switching waveforms for a Buck converter are shown in Figure 1-14.
FIGURE 1-13:
BUCK CONVERTER
IIN
Q1
L
VIN
IOUT
+ IL -
D1
VOUT
In a Buck converter, a switch (Q1) is placed in series with the input voltage source VIN.
Input source VIN feeds the output through the switch and a low-pass filter, implemented
with an inductor and a capacitor.
In a steady state of operation, when the switch is ON for a period of TON, the input
provides energy to the output as well as to the inductor (L). During the TON period, the
inductor current flows through the switch and the difference of voltages between VIN
and VOUT is applied to the inductor in the forward direction, as shown in Figure 1-13.
Therefore, the inductor current IL rises linearly from its present value IL1 to IL2.
During the TOFF period, when the switch is OFF, the inductor current continues to flow
in the same direction as the stored energy within the inductor, which continues to
supply the load current. Diode D1 completes the inductor current path during the Q1
OFF period (TOFF); thus, it is called a freewheeling diode. During the TOFF period, the
output voltage VOUT is applied across the inductor in the reverse direction, as shown in
Figure 1-14. Therefore, the inductor current decreases from its present value IL2 to IL1.
FIGURE 1-14:
BUCK CONVERTER SWITCHING WAVEFORM
Q1GATE
t
VIN - VOUT
VL
t
-VOUT
(VIN - VOUT)/L
IIN
IL
t
IL2
-VOUT/L
IL1
t
© 2008 Microchip Technology Inc.
DS70320B-page 17
SMPS AC/DC Reference Design User’s Guide
The inductor current is continuous and never reaches zero during one switching period
(TS); therefore, this mode of operation is known as Continuous Conduction mode. In
Continuous Conduction mode, the relation between the output and input voltage is
given by Equation 1-1. The duty cycle is given by Equation 1-2.
EQUATION 1-1:
V OUT = D ⋅ VIN
where D is the duty cycle
EQUATION 1-2:
t on
D = -----TS
where ton is the ON time and TS is the switching time period
When the output current requirement is high, the excessive power loss inside
freewheeling diode D1 limits the minimum output voltage that can be achieved. To
reduce the loss at high current, and to achieve lower output voltage, the freewheeling
diode is replaced by a MOSFET with a very low ON state resistance (RDS(ON)). This
MOSFET is turned on and off synchronously with the Buck MOSFET. Therefore, this
topology is known as a Synchronous Buck converter. A gate drive signal, which is the
complement of the Buck switch gate drive signal, is required for this synchronous
MOSFET.
A MOSFET can conduct in either direction; which means the synchronous MOSFET
should be turned off immediately if the current in the inductor reaches zero because of
a light load. Otherwise, the direction of the inductor current will reverse (after reaching
zero) because of the output LC resonance. In such a scenario, the synchronous
MOSFET acts as a load to the output capacitor, and dissipates energy in the RDS(ON)
of the MOSFET, resulting in an increase in power loss during the discontinuous mode
of operation (inductor current reaches zero in one switching cycle). This may happen if
the Buck converter inductor is designed for a medium load, but needs to operate at no
load and/or a light load. In this case, the output voltage may fall below the regulation
limit if the synchronous MOSFET is not switched off immediately after the inductor
reaches zero.
DS70320B-page 18
© 2008 Microchip Technology Inc.
Introduction
1.3
MULTI-PHASE SYNCHRONOUS BUCK CONVERTER
If the load current requirement is more than 35-40 amps, more than one converter is
connected in parallel to deliver the load. To optimize the input and output capacitors, all
the parallel converters operate on the same time base and each converter starts
switching after a fixed time/phase from the previous one. This type of converter is called
a Multi-Phase Synchronous Buck converter, which is shown in Figure 1-15. Figure 1-16
shows gate pulse timing relation of each leg and the input current drawn by the
converter. The fixed time/phase is given by Time period/n (or 360/n), where “n” is the
number of the converters connected in parallel.
The design of input and output capacitors is based on the switching frequency of each
converter multiplied by the number of parallel converters. The ripple current seen by
the output capacitor reduces by “n” times. As shown in Figure 1-16, the input current
drawn by a Multi-Phase Synchronous Buck converter is continuous, with less ripple
current as compared to a single converter. Therefore, a smaller input capacitor meets
the design requirement in the case of a Multi-Phase Synchronous Buck converter.
FIGURE 1-15:
MULTI-PHASE SYNCHRONOUS BUCK CONVERTER
+
IQ1
Q3
Q1
IQ5
IQ3
Q5
L3
VIN
IL3
L2
IL2
L1
IL1
VOUT
CO
Q2
Q4
Q6
-
FIGURE 1-16:
SWITCHING WAVEFORM OF SYNCHRONOUS BUCK
CONVERTER
IL1
Q1PWM
t
IL2
Q3PWM
t
IL3
Q5PWM
t
IQ5 + IQ1
IQ1
IQ1 + IQ3
IQ5 + IQ1
IQ3 + IQ5
IQ3
IQ5
IIN
t
© 2008 Microchip Technology Inc.
DS70320B-page 19
SMPS AC/DC Reference Design User’s Guide
1.3.1
Auxiliary Supply Description
The auxiliary power supply is based on the flyback topology, where it generates a
voltage source for the control circuitry and MOSFET drivers on both sides of the
isolation boundary. The multiple output flyback converter is controlled by a TNY277G
switch; the block diagram is shown in Figure 1-17. The auxiliary power supply
generates four isolated outputs, where on each side of the isolation barrier, the auxiliary
transformer will generate a voltage source for the MOSFET drivers and a voltage
source for the control circuitry.
A flyback converter is a transformer-isolated converter based on the basic Buck
topology. In a flyback converter, a switch is connected in series with the transformer
primary. The transformer is used to store energy during the ON period of the switch,
and provides isolation between the input voltage source VIN and the output voltage
VOUT. During the TOFF period, the energy stored in the primary of the flyback
transformer transfers to secondary through the flyback action. This stored energy
provides energy to the load, and charges the output capacitor. Since the magnetizing
current in the transformer cannot change instantaneously when the switch is turned
OFF, the primary current transfers to the secondary, and the amplitude of the
secondary current will be the product of the primary current and the transformer turns
ratio.
FIGURE 1-17:
AUXILIARY POWER SUPPLY BLOCK DIAGRAM
HV Bias Supply
+13V
LIVE_GND
High-Voltage
Bus (400V)
Live Drive
Supply
+7V
Live Digital
Supply
LIVE_GND
+HV_BUS
+17V
Drive Supply
C
R
GND
D
+7V
Digital Supply
GND
Energy Efficient
Switching Converter
D
TNY277
S
-HV_BUS
F/B
Bias Supplies
At the end of the ON period, when the switch is turned OFF, there is no current path to
dissipate the stored leakage energy in the magnetic core of the flyback transformer.
There are many ways to dissipate this leakage energy. One such method is shown in
Figure 1-17 as a snubber circuit consisting of D, R, and C. In this method, the leakage
flux stored inside the magnetic core induces positive voltage at the non-dot end primary
winding, which forward-biases diode D and provides the path to the leakage energy
stored in the core, and clamps the primary winding voltage to a safe value. Because of
the presence of the secondary reflected voltage on the primary winding and the
leakage stored energy in the transformer core, the maximum voltage stress VDS of the
switch is approximately 1.6 times the input voltage (i.e., 400•1.6 = 660V).
DS70320B-page 20
© 2008 Microchip Technology Inc.
Introduction
1.4
LISTING OF I/O SIGNALS FOR EACH BLOCK, TYPE OF SIGNAL AND
EXPECTED SIGNAL LEVELS
1.4.1
PFC Boost Converter
As indicated in the block diagram in Figure 1-18, three input signals are required to
implement the control algorithm. The only output from the dsPIC DSC device is firing
pulses to the Boost converter switch to control the nominal voltage on the DC bus in
addition to presenting a resistive load to the AC line. Table 1-1 shows the dsPIC DSC
resources used by the PFC application.
FIGURE 1-18:
RESOURCES REQUIRED FOR DIGITAL PFC
IPFC
VHV_BUS
|VAC|
k1(1)
k3(1)
VAC
k2
(1)
FET Driver
ADC Channel PWM Output
ADC Channel
ADC Channel
dsPIC33FJ16GS504
Note 1:
K1, K2 and K3 are feedback gain circuits. See A.3 “SMPS AC/DC Reference Design Schematics” for detailed
schematics.
TABLE 1-1:
RESOURCES REQUIRED FOR DIGITAL PFC
Type of Signal
dsPIC® DSC Resources
Used
Output Voltage (VHV_BUS)
Analog
AN5
3.01V (nominal)
PFC Current (IPFC)
Analog
AN4
2.5V (maximum)
Analog
AN3
1.9V (maximum)
PFC Drive Output, Digital
PWM4L
—
Description
AC Input Voltage (VAC)
PFC Gate Drive
© 2008 Microchip Technology Inc.
Expected Signal Level
DS70320B-page 21
SMPS AC/DC Reference Design User’s Guide
1.4.2
Phase-Shift ZVT Converter
As indicated in the block diagram in Figure 1-19, three input signals are required to
implement the control algorithm. The only outputs from the dsPIC DSC device are firing
pulses to the Full-Bridge Phase-Shift ZVT and synchronous MOSFETs to control the
nominal voltage on VOUT.
FIGURE 1-19:
RESOURCES REQUIRED FOR DIGITAL PHASE-SHIFT ZVT CONVERTER
Isolation
Barrier
VHV_BUS
VOUT
IZVT
FET
Driver
k4(1)
ADC
Channel
PWM
PWM
PWM
PWM
FET
Driver
k5(1)
FET
Driver
ADC
Channel
ADC
Channel
PWM
dsPIC33FJ16GS504
UART
RX
Note 1:
dsPIC33FJ16GS504
PWM
UART
TX
K4 and K5 are feedback gain circuits. See A.3 “SMPS AC/DC Reference Design Schematics” for detailed
schematics.
Table 1-2 shows the dsPIC DSC resources used by Phase-Shift ZVT application.
TABLE 1-2:
RESOURCES REQUIRED FOR DIGITAL PHASE-SHIFT ZVT
Description
Type of Signal
dsPIC® DSC
Resources Used
Expected Signal Level
ZVT CURRENT 1 (IZVT1)
Analog
AN0
1.5V (maximum)
ZVT CURRENT 2 (IZVT2)
Analog
AN2
1.5V (maximum)
Voltage Sense (VOUT)
Analog
AN5 (secondary side)
2.79V (maximum)
Full-Bridge Drive Outputs,
Digital
PWM1H, PWM1L,
PWM2H, PWM2L
—
Sync FET Drive Outputs, Digital
PWM3H, PWM3L
—
ZVT Gate Drive
Synchronous Rectifier
Gate Drive
DS70320B-page 22
© 2008 Microchip Technology Inc.
Introduction
Table 1-3 shows the common resources used on the Primary side.
TABLE 1-3:
PRIMARY COMMON RESOURCES
Type of Signal
dsPIC® DSC Resources
Used
Expected Signal Level
Live_MCLR
Digital
MCLR
—
Live_PGC
Digital
PGC
—
Live_PGD
Digital
PGD
—
Live_Fault
Digital
RC6
—
Live_RS232_TX
Digital
UART1 Transmit
—
Live_RS232_RX
Digital
UART1 Receive
—
Live_Temp_Sense
Analog
AN10
1.4V
Signal Name
1.4.3
Secondary Side Synchronous Buck Converters
Figure 1-20 shows the input signals required to implement the control algorithms for the
synchronous Buck converters. The output from the dsPIC DSC device is firing pulses
to the Multi-Phase as well as Single-Phase Synchronous Buck converters.
FIGURE 1-20:
RESOURCES REQUIRED FOR DIGITAL SYNCHRONOUS BUCK CONVERTERS
3.3V Output
12V Input
I3.3V_1
5V Output
I3.3V_2
I5V
FET
Driver
FET
Driver
(1)
Analog
Comp.
I3.3V_3
(1)
k11
k7
ADC
Channel
PWM
PWM
(1)
k6
PWM
PWM
ADC
Channel
FET
Driver
PWM
PWM
(1)
k5
PWM
PWM
FET
Driver
(1)
Analog Comparator
k8
Analog Comparator
k9
Analog Comparator
k10
(1)
dsPIC33FJ16GS504
(1)
UART
TX
ADC Channel
Note 1:
K5 through K11 are feedback gain circuits. See A.3 “SMPS AC/DC Reference Design Schematics” for
detailed schematics.
© 2008 Microchip Technology Inc.
DS70320B-page 23
SMPS AC/DC Reference Design User’s Guide
Table 1-4 shows the dsPIC DSC resources used by Multi-Phase as well as
Single-Phase Synchronous Buck converters.
TABLE 1-4:
RESOURCES REQUIRED FOR SECONDARY SIDE SYNCHRONOUS BUCK
CONVERTERS
Type of Signal
dsPIC® DSC Resources
Used
Expected Signal Level
5V Buck Current
Analog
AN0, CMP1A
1.25V (maximum)
5V Buck Output
Analog
AN1
2.7V (nominal)
3.3V Buck Current 1
Analog
AN2, CMP2A
1V (maximum)
3.3V Buck Current 2
Analog
AN4, CMP3A
1V (maximum)
3.3V Buck Current 3
Analog
AN6, CMP4A
1V (maximum)
3.3V Buck Output
Analog
AN3
1.65V (nominal)
Single-Phase Synchronous
Buck Drive
PWM4H, PWM4L
—
Multi-Phase Synchronous
Buck Drive
PWM1H, PWM1L,
PWM2H, PWM2L,
PWM3H, PWM3L,
—
12V Bus Sense
Analog
AN5
2.79V
12V Digital Feedback
Digital
UART1 Transmit
—
RS232_RX
Digital
UART1 Receive
—
Temperature Sense
Analog
AN8
1.4V
MCLR
Digital
MCLR
—
PGC
Digital
PGC
—
PGD
Digital
PGD
—
Fault_Reset
Digital
RC6
—
Description/Signal Name
5V Buck Gate Drive
3.3V Buck Gate Drive
DS70320B-page 24
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Chapter 2. Hardware Design
This chapter provides a functional overview of the SMPS AC/DC Reference Design
and identifies the major hardware components. Topics covered include:
•
•
•
•
•
2.1
PFC Boost Converter
Full-Bridge ZVT Converter
Single-Phase Synchronous Buck Converter
Three-Phase Synchronous Buck Converter
Auxiliary Power Supply
PFC BOOST CONVERTER
The conventional single-phase power factor correction circuit is a standard Boost
converter topology operating from the full wave rectified mains input, as shown in
Figure 2-1.
The converter controller has an inner current control loop and outer voltage control
loop. The current reference waveform is the input rectified mains voltage, so that the
resultant current drawn from the mains is essentially sinusoidal and in-phase with the
mains voltage. The amplitude of the current is controlled by the duty cycle of the fixed
frequency PWM of the MOSFET, and is controlled by the PWM reference, which is the
product of the current reference and the output of the DC link voltage error amplifier.
Refer to Chapter 1. “Introduction” for details on the operation of this converter.
FIGURE 2-1:
PFC POWER CONVERTER
VDC
VAC
EQUATION 2-1:
D(t ) =1−
Vac ( t )
Vdc
EQUATION 2-2:
THD = 100
I%ac2 − I%12
I%
1
© 2008 Microchip Technology Inc.
DS70320B-page 25
SMPS AC/DC Reference Design User’s Guide
2.1.1
Power-Train Design
The target specification for the PFC converter is as follows:
•
•
•
•
•
•
Input voltage, VIN = 85-265 Vrms
Input frequency, fin = 45-65 Hz
Switching frequency, fsw = 125 kHz
Maximum Output voltage, VOUT = 420 VDC
Maximum Output power, POUT = 450 W
Current THD < 5%
EMC standards for conducted, radiated and line current harmonics:
•
•
•
•
•
FCC Class B
EN55022 (CISPR 22) Class B
EN61000-3-2 A14 Class A
EN61000-3-3
IEEE519
2.1.1.1
MOSFETS AND GATE DRIVE
MOSFETs are the preferred technology for the Boost converter power switch because
of the high operating frequency. The rms current in the MOSFET switch can be
approximated using Equation 2-3.
EQUATION 2-3:
i%mos =
Po
Vac 2
2−
16Vac 2
3π Vdc
Psw = 12 CossVdc2 f sw
The maximum rms current occurs at minimum mains voltage, so the maximum normal
operating MOSFET current is 4.6 Arms. Therefore, two TO-220 Infineon CoolMOS™
SPP11N60CFD 500V, 0.44Ω MOSFETs are connected in parallel, with each
dissipating 2.3W of conduction loss. The MOSFET output capacitance is 390 pF so the
switching loss is estimated at about 0.4W each. The actual loss in practice will be layout
dependent and will probably be a factor of 2 higher, but still low enough to achieve high
system efficiency.
The gate drive circuitry is a low-side Microchip TC1412N gate-drive IC, which drives
the MOSFET gates directly. A single dsPIC DSC PWM module pin interfaces with the
gate-drive IC via an inverting open-collector transistor stage which provides immunity
against noise voltage differences between the Boost converter common and dsPIC
DSC signal common (ground bounce).
2.1.1.2
OUTPUT DIODE
The output diode must be rated for the mean output current, which is given by
Equation 2-4.
EQUATION 2-4:
idiode =
DS70320B-page 26
Po
Vdc
© 2008 Microchip Technology Inc.
Hardware Design
In this design, the diode must be rated for 1.2A, so a STMicroelectronics STTH5R06D
600V, 5A TO-220 ultra-fast high-voltage rectifier has been selected. The typical forward
voltage drop at high junction temperature is 1.4V, which means that the device will run
cool since the dissipation is only 1.7 Watts. There will be additional switching losses
due to the high switching frequency and diode recovery characteristics. For a lower
cost solution, a smaller axial diode may be used. Alternatively, if switching losses are
an issue, then the recently introduced SiC Schottky diode would be an attractive option.
2.1.1.3
PFC CHOKE
The target THD of the input current is 5%, which means the non-fundamental (50 Hz
nominal) rms current component must be only 1% of input rms current. This component
is the high-frequency ripple current in the Boost inductor, and is dependent on the
inductance. If it is assumed that, on average, the duty cycle is 0.5, the ripple current
rms of a triangular waveform is given by Equation 2-5.
EQUATION 2-5:
I rms =
I 2pk − pk
12
Therefore, for a 5.3 Arms input current, we can only allow a maximum of 0.2A
peak-to-peak, which will entail a large inductor size. However, the high frequency
capacitor placed across the output terminals of the bridge rectifier will shunt-off most of
the high frequency current so that a larger component of ripple can be tolerated in a
smaller inductor. Note that too large a capacitance will cause distortion in the current
waveform, so a design compromise must be reached. The inductor current
peak-to-peak ripple in a PFC Boost converter varies over the whole mains cycle and
depends on the input voltage, as shown in Equation 2-6.
EQUATION 2-6:
iripple
)
DVac
=
Lf sw
However, the absolute maximum value is independent of input voltage and is
calculated from Equation 2-7.
EQUATION 2-7:
V
iˆripple = dc
4 Lfsw
In this design, the ripple current is chosen to be 25% of the minimum voltage peak
mains current; therefore, inductance of about 400 μH is required. The Boost choke
uses a Kool Mu 77548 core, which has an outside diameter of 33 mm. The AL value for
this core is 127. A single layer of 58 turns of 0.9 mm (19 AWG) enameled copper fits
on the core giving an unsaturated inductance of 427 μH. From the Magnetics Inc.
published wire-core tables, this results in a predicted winding resistance of 77 mΩ at
100ºC. The variation of ripple current for a selection of input voltages is shown in
Figure 2-2 and Figure 2-3.
The core loss can be roughly estimated from the mean flux density over a complete
mains cycle. The worst case condition occurs at roughly 180 Vrms, where the mean
flux density is 180 mT.
© 2008 Microchip Technology Inc.
DS70320B-page 27
SMPS AC/DC Reference Design User’s Guide
FIGURE 2-2:
INPUT VOLTAGE RIPPLE CURRENT VARIATION
FIGURE 2-3:
CORE LOSSES FOR PFC CHOKE
2.1.1.4
PFC OUTPUT CAPACITOR
The PFC output capacitor provides bulk capacitance on the output of the PFC Boost
converter and smooths the DC voltage input to the ZVT Full-Bridge converter. The size
of the capacitor is dictated by the hold-up requirements of the SMPS, its AC ripple
current and thermal lifetime under normal operating conditions.
DS70320B-page 28
© 2008 Microchip Technology Inc.
Hardware Design
The capacitance must be high enough to maintain the PFC output voltage within
acceptable bounds under normal peak power operating conditions and when a mains
brown-out occurs. The required hold-up time, thold, at the minimum mains frequency is
22 ms, therefore the conditions of Equation 2-8 must be met.
EQUATION 2-8:
C>
2thold Po
(V −Vdc2 (min) )
2
dc
For a minimum DC link voltage of 300V, a 330 μF is required. The actual capacitor
selected is a Panasonic EET-ED2W331EA 35 x 40 mm electrolytic capacitor rated to
450 VDC and 105ºC. The ESR at 20 kHz is 0.181Ω, and the maximum ripple current
rating at 105ºC is 2.64 Arms.
2.1.1.5
EMI FILTER
The SMPS AC/DC Reference Design has been designed to meet international
standards for conducted EMC. The EMI filter between the mains input terminals and
the PFC is a two-stage design because of the high switching frequency of the different
stages in the SMPS. The circuit is shown in Figure 2-4. The two common-mode chokes
are rated to 6 Arms, and the 2.2 mH inductance forms a filter with the capacitors to
Earth Ground for common-mode noise. The leakage inductance of the chokes together
with the capacitors across the live and neutral terminals, filter the differential-mode
noise.
The six capacitors connected to Earth Ground are 2.2 nF Y2-class capacitors meeting
the CATII overvoltage category. The two X2-class capacitors are 220 nF. A transient
spike voltage protection MOV is also fitted across the mains input, and a 470 kΩ
discharging resistor is fitted across the input to the SMPS to ensure that the filter
capacitors discharge within one second.
Note:
The EMI/EMC filter value has been chosen based on switching frequencies
and expected noise levels in the system. This value may be changed based
on the final test results of EMI/EMC.
FIGURE 2-4:
L
EMI FILTER
L
Mains
N
N
E
E
© 2008 Microchip Technology Inc.
DS70320B-page 29
SMPS AC/DC Reference Design User’s Guide
2.2
FULL-BRIDGE ZVT CONVERTER
The main power circuit for a ZVT Full-Bridge converter is shown in Figure 2-5. It is a
standard Full-Bridge converter, but with additional series resonant inductance, which
limits the rise rate of current at switching transitions and can eliminate turn-off switching
power dissipation in the MOSFETs. The stray leakage inductance of the transformer
forms part of the series resonant inductor and, in this particular design, is large enough
to ensure quasi-resonant operation over 80% of the operating power range without the
need for an additional inductor. The secondary-side high-frequency rectification is
normally done by using ultrafast recovery rectifiers or Schottky diodes. Alternatively,
lower loss rectification can be achieved by using MOSFETs operating as synchronous
rectifiers with primary-side commutation control, and this is the preferred solution in this
reference design.
ZVT operation occurs when the stored energy in the inductor is transferred to the
capacitor in parallel with the MOSFET. In this design, the stray output capacitance of
the MOSFET is large enough to not require additional capacitors in parallel. From
Reference 3 (see Appendix C. “References”), the equation relating energy in the
MOSFET output capacitance and the series inductance for ZVT operation is given by
Equation 2-9.
EQUATION 2-9:
1
4
LR I 2pri ≥ C RVin2
2
3
This ensures that there is more than enough energy to charge the MOSFET output
capacitance and maintain ZVT operation. Note that at low output power, there will be
far less energy stored in the resonant inductance so ZVT operation will be lost. The
inductor is therefore selected based on the minimum operating output power for ZVT
switching.
The modulation control scheme required for ZVT operation of a Full-Bridge converter
is phase-shifted PWM. The ideal power stage waveforms for the circuit are shown in
Figure 2-6. The ZVT transition in the switch is short in comparison with the primary
current transition time. This time, Δt, is dictated by the resonant inductance, LR, which
is given by Equation 2-10.
EQUATION 2-10:
Δt = 2
LR I pri
Vin
The control duty cycle is limited in a ZVT due to the time taken for the current to rise/fall
during switching transitions. The maximum duty cycle, Dmax, is achievable under the
ZVT operating conditions given by Equation 2-11.
EQUATION 2-11:
Dmax = 1 −
DS70320B-page 30
2Δt
T
© 2008 Microchip Technology Inc.
Hardware Design
The transformer turns ratio, n, for the current doubling synchronous rectifier topology
can then be selected for the required operating input and output voltages using the
following ideal relationship shown in Equation 2-12.
EQUATION 2-12:
n = Dmax
Vin
2Vo
The previous equations governing the ZVT operation and resonant circuit component
selection are also dependent on the peak primary current. If the output inductor
magnetizing current is ignored, the primary peak current is given by Equation 2-13.
EQUATION 2-13:
I pri =
FIGURE 2-5:
I oVin
n 2Vo
ZVT FULL-BRIDGE POWER CONVERTER WITH
SYNCHRONOUS RECTIFICATION
CR
CR
VPRI
Q1
Q4
IPRI
VIN
LR
Q2
CR
CR
Q3
VSEC
Q5
VOUT
Q6
© 2008 Microchip Technology Inc.
DS70320B-page 31
SMPS AC/DC Reference Design User’s Guide
FIGURE 2-6:
ZVT WAVEFORMS
Q1
t
Q3
t
Q2
t
Q4
t
VPRI
t
IPRI
t
VSEC
t
Δt
ton
T
DS70320B-page 32
© 2008 Microchip Technology Inc.
Hardware Design
2.2.1
Full-Bridge ZVT Power-Train Design
The target specification for the ZVT Full-Bridge converter is as follows:
•
•
•
•
Input voltage, VIN = 390-420V
Switching frequency, fsw = 250 kHz
Maximum output voltage, VOUT = 12V
Maximum output current, IOUT = 33A
2.2.1.1
MOSFETS AND GATE DRIVE
Care must be taken when selecting the MOSFET switch for the ZVT Full-Bridge since
there are potential failure modes associated with the diode characteristic and timing
control at light loads (see Reference 4 in Appendix C. “References”). For this
reference design, an Infineon CoolMOS CFD device has been selected because of its
optimized diode characteristic. The SPA11N60CFD is a 600V, 0.44Ω MOSFET in a
TO-220 package, and is a good compromise between cost and efficiency for this output
power rating. The output capacitance, COSS, is 45 pF and will form the resonant
capacitor for ZVT operation.
Gate driving is typically achieved with either a proprietary high-side and low-side
high-voltage driver IC, or using a small transformer. These circuit techniques provide
level-shifting of the dsPIC DSC gate firing signals. Adequate voltage creepage and
clearance distances are maintained in the layout. Given the high switching frequency
in this application, the transformer isolated gate drive approach has been adopted. This
is because of thermal concerns in standard gate driver ICs, although there are potential
candidates from a number of manufacturers available on the market.
A single drive transformer with two secondary windings manufactured by Sirio
Elettronica is used for each half-limb, and the turn-on switching time is controlled by a
single resistor in each MOSFET gate. Turn-off is much faster due to the diode across
the gate resistor. The drive for each transformer primary is provided by a Microchip
TC1404, which is a dual high-speed CMOS driver IC. The dead-time for each MOSFET
half-limb is inserted by the dsPIC DSC PWM peripheral module and is selected to avoid
any possible shoot-through condition based on the timing delays inherent in the
transformer gate drive circuitry.
2.2.1.2
TRANSFORMER
The following section describes a basic procedure for designing the ZVT Full-Bridge
transformer. The optimum choice of ferrite core and winding turns/construction is
dependent on many factors in the overall converter and may well involve a number of
design optimization iterations.
The transformer turns ratio must be selected to ensure that voltage regulation is
maintained at the maximum duty limit. As a starting point, Dmax is assumed to be 0.85,
so for the minimum DC link voltage (390V) and the output voltage (12.5V), which
includes the voltage drop across the synchronous rectifiers and output chokes, the
required transformer turns ratio is 13.3 or less (see Equation 2-12).
An ungapped ETD29 ferrite core pair is selected for the transformer. Table 2-1 lists the
various parameters for ETD29 cores made of N87 material.
TABLE 2-1:
TRANSFORMER CORE DATA
AL
ETD29
© 2008 Microchip Technology Inc.
(nH/Turn2)
(mm2)
Ae
(mm3)
Ve
w
h
lm
Rth
(mm)
(mm)
(mm)
(ºC/W)
2200
76
5350
19.4
4.85
52.8
28
DS70320B-page 33
SMPS AC/DC Reference Design User’s Guide
To compute the operating flux density and decide on the number of turns, it is assumed
that the maximum allowable transformer temperature rise is 80ºC. The power converter
will have forced air cooling from a lid mounted fan so the actual thermal resistance will
be about a third less at around 10ºC/W. This means that the total power dissipation can
be as much as 8W, with the losses split equally in the copper windings and the ferrite
cores. From the core loss curves shown in Figure 2-7, the operating AC peak-to-peak
flux density can be as high as 150 mT. The minimum number of turns at the given
maximum operating duty is given by Equation 2-14.
EQUATION 2-14:
N min =
Vin DmaxT
2 Ae Bmax
Therefore, Nmin is 58, which means that the secondary winding must have either four
or five turns to give the required turns ratio computed above. Based on this, a good
solution is 64 turns on the primary and five turns on the secondary with a turns ratio, n,
of 12.8.
FIGURE 2-7:
N87 POWER LOSS CURVES
With the selected turns ratio from Equation 2-12, the actual operating duty, D, is 0.82.
Therefore, the operating peak-to-peak flux density is 130 mT at 250 kHz. The core loss
in a N87 core can be computed by Equation 2-15.
EQUATION 2-15:
Pcore = 1.36 × 10−4 fsw1.59 B 2.74Ve
where fsw is in kHz and B is in mT.
DS70320B-page 34
© 2008 Microchip Technology Inc.
Hardware Design
Therefore, the predicted core loss is actually 2.9W. The next stage is now to optimize
the winding designs to minimize the losses, especially the high-frequency AC losses
due to skin-effect and the proximity effect in multilayer windings (see Reference 5 and
Reference 6 in Appendix C. “References”). The available winding width, bw, must be
reduced to accommodate a 3 mm creepage border on each side of the bobbin, leaving
around 13 mm available for the windings. The total height of the two windings must be
less than 4.5 mm which takes into account the layers of 0.05 mm inter-winding tape.
The secondary transformer winding rms current for the current-doubler synchronous
rectifier, ignoring inductor ripple current, is given by the relationship shown in
Equation 2-16.
EQUATION 2-16:
I
I%sec = o
2
The secondary rms current is therefore 16.5A, but will be slightly higher in practice due
to the magnetizing ramp component of current in the output inductors. For high current
windings, copper foil is better suited to utilize the available winding area, and minimize
AC copper losses. The secondary winding is a 5 turn strip of copper, and the ideal foil
height, hid, in mm is given by Equation 2-17.
EQUATION 2-17:
hid =
9.74 × 103
Ns f sw
So hid at 250 kHz is 0.088 mm. The resistance factor, FR, is given by Equation 2-18.
EQUATION 2-18:
1⎛ h ⎞
FR = 1 + ⎜ ⎟
3 ⎝ hid ⎠
when
4
h
< 1.4
hid
Therefore, for a practical foil thickness, h, of 0.1 mm, FR = 1.56. The total resistance
including AC effects is given by Equation 2-19.
EQUATION 2-19:
r=
FR lm
45 × 10 6 bwh
where lm is the mean turn length
and bw is the foil width.
The realistic foil width for the ETD29 is 13.0 mm. This means that the secondary
resistance is 1.4 mΩ, which leads to a secondary winding copper loss of about 0.5 W.
The current density is actually 14A/mm2 and, although very high, the power loss is
acceptable.
© 2008 Microchip Technology Inc.
DS70320B-page 35
SMPS AC/DC Reference Design User’s Guide
There is no requirement to reduce leakage inductance in the transformer design so the
primary winding can be a single winding block. This may also reduce the inter-winding
capacitance between primary and secondary and have an impact on EMC. For the low
current primary with the large number of turns, round conductors are preferred.
Equation 2-20 can be used to identify the ideal wire diameter at the operating
frequency, which takes into account skin and proximity effects, and was derived
through experimental work by Dowell in the 1960s (see Reference 7 in Appendix
C. “References”).
EQUATION 2-20:
⎛ 17.1bw ⎞
did = 1.01⎜
⎟
⎝ SNfsw ⎠
1
3
where S is the number of strands and N is number of turns in the winding portion.
The resistance factor for round conductors can then be computed from Equation 2-21.
EQUATION 2-21:
1⎛ d ⎞
FR = 1 + ⎜
⎟
2 ⎝ did ⎠
6
The best fill factor is achieved by using seven-stranded Litz wire, so this is a starting
point. Also, assume four layers as an initial starting point with 16 turns per layer.
Therefore, the ideal optimum wire diameter is 0.2 mm (32 AWG), giving a resistance
factor of 1.5. A commercially available Litz wire has eight strands of 0.2 mm with an OD
of 0.75 mm. The DC resistance per meter for single 0.2 mm strand at 100ºC is 0.7074
Ω/m, so the resistance for one mean turn of eight strands is 4.7 mΩ. Therefore, the total
adjusted AC resistance of the primary winding is 0.45Ω. The primary current is 1.3
Arms for the estimated secondary current and selected transformer turns ratio, which
leads to a primary winding loss of 0.8W. Therefore, the total transformer losses are
estimated to be 4.2W in the ETD29 transformer, and will limit the total temperature rise
to less than 80ºC with forced air-cooling.
The primary Litz wire has a total OD of around 0.7 mm, so the four-layer primary
winding height is about 3 mm. The secondary five-layer foil winding height, including
inter-turn insulation, will be around 0.75 mm height, and this design will easily fit in the
available 4.85 mm maximum winding window height. The final winding construction is
shown in Figure 2-8. The primary winding start and finish are terminated to bobbin pins,
while the secondary winding has flying leads which are soldered directly into the PCB.
DS70320B-page 36
© 2008 Microchip Technology Inc.
Hardware Design
FIGURE 2-8:
ZVT FULL-BRIDGE TRANSFORMER WINDING
CONSTRUCTION
3 layers 0.05 mm Melinex
3 mm
margin
5 turns of 0.1 mm x 13 mm foil
3 mm
margin
3 layers 0.05 mm Melinex
3 mm
margin
64 turns of 8 x 0.2 mm Litz wire
3 mm
margin
The last check is to assess whether an additional inductor is needed for ZVT operation.
The leakage inductance (see Reference 8 in Appendix C. “References”) for a
standard construction transformer is given by Equation 2-22.
EQUATION 2-22:
LL =
4π × 10−4 lm N p2 ⎛
hp + hs ⎞
⎜c+
⎟ µH
3 ⎠
bw
⎝
where c is the space between the primary and secondary.
All dimensions are in mm.
The ideal computed leakage inductance is 32 µH, which meets the criteria of
Equation 2-9 with the MOSFET output capacitance without requiring an additional
resonant inductor. Provision has been made on the reference design PCB for an
external inductor and parallel capacitors across the MOSFETs if the ZVT operation
needs tuning.
2.2.2
Synchronous Rectifier Design
Synchronous rectification is the technique used for reducing the power loss in the
output stage of switched mode power supplies (see Reference 9 in Appendix
C. “References”). The conventional diode is replaced by a MOSFET and is controlled
so that current will flow into the third quadrant in the source-to-drain direction when the
equivalent RDS(ON) voltage drop is lower than the intrinsic diode voltage drop. This
means that with very low RDS(ON) MOSFETs, the power supply efficiency can be
significantly increased. This is especially true in low output voltage power supplies.
© 2008 Microchip Technology Inc.
DS70320B-page 37
SMPS AC/DC Reference Design User’s Guide
In push-pull type power converters, there are a number of synchronous rectifier
topologies. In this particular reference design, a current-doubler form has been used
(see Reference 10 in Appendix C. “References”). Figure 2-9 illustrates the current
paths for the four operating modes of the rectifier. The MOSFET commutation is
synchronized to the ZVT Full-Bridge switching and gate control signals are generated
by the primary-side dsPIC DSC device and fed to the secondary-side via high-speed
opto-isolators.
The operating waveforms for the synchronous rectifier are shown in Figure 2-10. As
shown in the figure, switch Q6 is gated when the primary current is positive, which
coincides with the gating of switch Q2, and Q5 is synchronized with the primary bridge
MOSFET Q4 gate signal.
FIGURE 2-9:
CURRENT-DOUBLER SYNCHRONOUS RECTIFIER
OPERATING MODES
ISEC
I1
IOUT
Q5
VOUT
VSEC
Q6
I2
ISEC
I1
IOUT
Q5
VOUT
VSEC
Q6
I2
ISEC
I1
Q5
IOUT
VOUT
VSEC
Q6
I2
Note: Dotted lines with arrows indicate current polarity.
DS70320B-page 38
© 2008 Microchip Technology Inc.
Hardware Design
FIGURE 2-10:
SYNCHRONOUS RECTIFIER WAVEFORMS
Q5 (Q4)
t
Q6 (Q2)
t
VSEC
t
ISEC
t
Iout
2
t
I1
Iout
2
t
I2
ton
T
2.2.2.1
MOSFET SYNCHRONOUS RECTIFIERS AND GATE DRIVES
The MOSFET rectifiers selected for the synchronous rectifier are International Rectifier
IRF2804SPBF 40V, 2 mΩ devices. They are packaged in a D2PAK and mounted
directly onto the PCB. The minimum blocking voltage required is equal to the peak
applied transformer secondary voltage. With the turns ratio of 12:8 and at the maximum
input voltage of 410V, this is 32V. There will be very little overshoot voltage due to the
compact layout achieved through mounting on the PCB, and the RC snubber across
the secondary winding. The junction to ambient thermal resistance is 40ºC/W when
mounted on a 25.4 mm2 (1 inch2) PCB copper pad (see Reference 11 in Appendix
C. “References”).
The MOSFET’s internal diode voltage characteristic is 0.6V at 30A, 175ºC, which is
significantly above the voltage drop across the 2 mΩ ON state resistance, and the
benefits of synchronous rectification are maintained over the whole operating power
region.
The gate-drive circuit employs a Microchip MCP1403 gate driver. The gate control
signals are generated by the primary-side dsPIC DSC device so that they are
synchronized with the ZVT Full-Bridge commutation. Isolation is achieved by
high-speed opto-isolators.
© 2008 Microchip Technology Inc.
DS70320B-page 39
SMPS AC/DC Reference Design User’s Guide
2.2.2.2
OUTPUT CHOKE
There are two output chokes in the current-doubler synchronous rectifier. Each
inductor's mean current is half the output current, and the fluxing voltage period occurs
in only half of the cycle. Therefore the ripple current is given by Equation 2-23.
EQUATION 2-23:
I ripple =
DT ⎛ Vin
⎞
− Vo ⎟
⎜
2L ⎝ n
⎠
The choke is designed to have a 20% current ripple component, and the inductance is
selected to give 3.3A at the nominal input voltage 400V. Therefore, the inductance
needs to be 9 μH and the winding rated for a current of around 18A.
The Magnetics Kool Mu core iron loss can be computed from Equation 2-24.
EQUATION 2-24:
2
⎛ DT ⎛ Vin
⎞ ⎞ 1.46
Pc = ⎜
⎜ − Vo ⎟ ⎟ f sw
⎠⎠
⎝ 2NAe ⎝ n
W/m3
The peak flux density is computed from Equation 2-25.
EQUATION 2-25:
B pk =
LI pk
NAe
T
Taking the 20.3 mm OD core No. 77848 with an AL = 32 with Ae = 22.6 mm2, the
required number of turns, N, is 17. Therefore, the peak-to-peak AC flux density is 77
mT and the core loss is 0.5W. According to the single-layer winding data, 1.6 mm OD
wire (14 AWG) will fit on the core so that the 13 x 0.315 mm will fit. The copper
cross-sectional area is 1.01 mm2, so the current density is 17.7A/mm2. The resistance
is estimated to be 7 mΩ (double the 14 AWG resistance) and so the copper loss is
2.2W. With forced air-cooling this is acceptable.
2.2.2.3
OUTPUT CAPACITOR
The main output capacitor is a 2200 μF, 25V electrolytic with two 10 μF, 25V multilayer
ceramic capacitors in parallel. The larger bulk capacitance provides the main energy
storage while the small ceramic capacitors with very low ESR provide the
high-frequency ripple current decoupling capability. The capacitor ripple current is the
combined AC components of the two output inductors plus the AC component of the
synchronous Buck loads. In the case of the inductor currents, this is lower compared
to a conventional output rectifier due to the 50% phase shift in the inductor currents,
and is dominated by current supplied to the synchronous Buck regulators. The
capacitor must also provide enough energy during a step load transient to maintain the
12V output voltage within the required regulated limits.
DS70320B-page 40
© 2008 Microchip Technology Inc.
Hardware Design
2.3
SINGLE-PHASE SYNCHRONOUS BUCK CONVERTER
The Single-Phase Synchronous Buck converter uses the same basic topology as the
standard step-down Buck converter, but replaces the free-wheel diode with a MOSFET.
Figure 2-11 shows the main power circuitry. The two switches are operated as a
complementary pair with a dead-time inserted by the PWM controller to avoid
shoot-through. The low-side MOSFET is operated in the third quadrant with current
flowing from source to drain when the current is required to free-wheel, and, due to the
very lower ON state resistance of the MOSFET, higher efficiency is achieved compared
with a conventional Schottky diode.
FIGURE 2-11:
SINGLE-PHASE SYNCHRONOUS BUCK CONVERTER
Vbus
2.3.1
Vout
Single-Phase Buck Converter Power-Train Design
The target specification for the Single-Phase Buck converter is as follows:
•
•
•
•
•
•
Input voltage, VIN = 12 VDC
Switching frequency, fsw = 500 kHz
Output voltage, VOUT = 5 VDC
Output current, IOUT = 23A
Voltage ripple < 2%
Output slew rate > 5 A/μs
2.3.1.1
MOSFETS AND GATE DRIVE
The equation governing the duty cycle of a Buck converter is shown in Equation 2-26.
EQUATION 2-26:
D=
Vo
Vbus
The rms currents in the high-side and low-side MOSFETs, assuming a low inductor
ripple current is as follows in Equation 2-27 and Equation 2-28.
EQUATION 2-27:
i%high = I o D
© 2008 Microchip Technology Inc.
DS70320B-page 41
SMPS AC/DC Reference Design User’s Guide
EQUATION 2-28:
i%low = I o 1 − D
The nominal duty cycle is 0.42 ignoring stray voltage drops and inductor current ripple.
This equates to a high-side MOSFET on-time, ton, of 840 ns. Therefore, the high-side
MOSFET is rated for 14.9 Arms and the low-side MOSFET is rated for 17.5A.
The selected MOSFETs are International Rectifier IRLR7833PBF, 30V, 4.5 mΩ devices
in a DPAK package, and are mounted directly on the PCB. The intrinsic reverse diode
of the MOSFET is relatively slow in switching, so a fast Schottky diode is placed in
parallel across the MOSFET to reduce switching loss. The conduction losses of the
high-side and low-side MOSFETs are estimated at 0.85W and 1.2W respectively. It is
possible in some designs to optimize the efficiency performance by selecting different
devices for the high-side and low-side switches.
The switching loss in the MOSFETs can be estimated by assuming ideal linear
switching transitions using Equation 2-29.
EQUATION 2-29:
Psw = 16 Vbus I otr fsw
The estimated switching transition time is 50 ns, so the switching loss for each device
is 2.4W.
The gate drive circuitry is a dual Microchip MCP1404 gate-drive IC, which drives the
MOSFET gates directly. The maximum gate threshold of each MOSFET is 4V, so the
drive circuit for the high-side MOSFET is provided by the auxiliary SMPS power rail,
which is higher than the gate threshold and source voltage combined. The PWM
module pin of the dsPIC DSC device interfaces with the gate-drive IC via an inverting
open-collector transistor stage, which provides immunity against ground bounce.
2.3.1.2
OUTPUT CHOKES
The ripple current is given by the relationship shown in Equation 2-30.
EQUATION 2-30:
Δi =
ton (Vbus − V0 )
L
The ripple current needs to be less than 25% of the output current so each output choke
is a 1 μH Coilcraft SER1360-102KL surface mount. A smaller inductor would improve
the output transient response, but at the expense of higher ripple current and,
consequently, higher output voltage ripple. The DC resistance is 2.5 mΩ and the power
loss is 1.3W for the maximum 23A output current. The peak-to-peak ripple current is
5.9A.
2.3.1.3
OUTPUT CAPACITOR
The output capacitor ripple current is fairly low in a Buck converter due to the
continuous inductor current, and is only dependent on the amplitude of ripple current
in the inductor. The relationship for capacitor rms current is given by Equation 2-31.
DS70320B-page 42
© 2008 Microchip Technology Inc.
Hardware Design
EQUATION 2-31:
Δi
i%cap =
12
Therefore, the total capacitor ripple current is 1.7 Arms. Another important design
consideration for the bulk capacitors is the transient load requirements, so low
ESR/ESL parts are required to meet the specification (see Reference 12 in Appendix
C. “References”). Two Rubycon 10ZL1500M10X23 1500 μF, 10V electrolytic
capacitors, plus four 10 μF, 16V multi-layer ceramics in parallel were selected. The
electrolytic capacitors are each rated to 2.15 Arms at 105ºC, which can easily handle
the ripple current on their own.
2.4
THREE-PHASE SYNCHRONOUS BUCK CONVERTER
Multiple synchronous Buck converters can be connected in parallel to increase the
power handling of a step-down voltage stage. Performance improvements and a
reduction in output capacitor size can be achieved by phase-shifting the PWM
modulation in each stage. In this reference design, a Three-Phase Synchronous Buck
converter has been designed. The power circuit is shown in Figure 2-12 and illustrates
the switching cycle 120 degree PWM phase-shifting in the output choke currents.
FIGURE 2-12:
VBUS
© 2008 Microchip Technology Inc.
THREE-PHASE SYNCHRONOUS BUCK CONVERTER
VOUT
DS70320B-page 43
SMPS AC/DC Reference Design User’s Guide
2.4.1
Multi-Phase Buck Converter Power-Train Design
The target specification for the Multi-Phase Buck converter is as follows:
•
•
•
•
•
•
Input voltage, VIN = 12 VDC
Switching frequency, fsw = 500 kHz
Output voltage, VOUT = 3.3 VDC
Output power, IOUT = 69 A
Voltage ripple < 2%
Output slew rate > 50A/μs
2.4.1.1
MOSFETS AND GATE DRIVE
The output current for each phase Buck stage is 23A and nominal duty cycle is 0.275
ignoring stray voltage drops and inductor current ripple. This equates to a high-side
MOSFET on-time, ton, of 550 ns. Therefore, the high-side MOSFET is rated for 12
Arms, and the low-side MOSFET is rated for 19.6A.
The selected MOSFETs are International Rectifier IRLR7833PBF, 30 V, 4.5 mΩ
devices in a DPAK package, and are mounted directly on the PCB. The conduction
losses of the high and low-side MOSFETs are estimated at 0.55W and 1.5W
respectively. The estimated switching transition time is 50 ns, so the switching loss for
each device is 1.2W.
The gate drive circuitry is a dual Microchip MCP1404 gate-drive IC, which drives the
MOSFET gates directly. The maximum gate threshold of each MOSFET is 4V, so the
drive circuit for the high-side MOSFET is provided by the auxiliary SMPS power rail,
which is higher than the gate threshold and source voltage combined. The PWM
module pin of the dsPIC DSC device interfaces with the gate-drive IC via an inverting
open-collector transistor stage which provides immunity against ground bounce.
EQUATION 2-32:
D=
Vo
Vbus
ihigh = I o D
ilow = I o 1 − D
Psw = 16 Vbus I otr fsw
2.4.1.2
OUTPUT CHOKES
The ripple current in each inductor is given by the relationship shown in Equation 2-33.
EQUATION 2-33:
Δi =
ton (Vbus − V0 )
L
The ripple current needs to be about 20% of the output current; therefore, each output
choke is a 1 μH Coilcraft SER1360-102KL surface mount. The DC resistance is 2.5 mΩ
and the power loss is 1.3W. The peak-to-peak ripple current is 4.8A.
DS70320B-page 44
© 2008 Microchip Technology Inc.
Hardware Design
2.4.1.3
OUTPUT CAPACITOR
The output capacitor ripple current is very low due to the continuous inductor currents
with phase shifting. The relationship for capacitor rms current is given by
Equation 2-34.
EQUATION 2-34:
i%cap =
Δi
3 12
Therefore, the total capacitor ripple current is 1.0 Arms. The output capacitor is made
up of three Rubycon 6.3ZL1500M10X20 1500 μF, 6.3V electrolytic capacitors plus
three 10 μF, 16V multi-layer ceramics in parallel. The electrolytic capacitors are each
rated to 1.82 Arms at 105ºC, and can easily handle the ripple current on their own.
2.5
AUXILIARY POWER SUPPLY
The supply voltages for the primary and secondary side dsPIC DSC device and gate
drive circuitry are generated by a simple low-power flyback SMPS. Figure 2-13
illustrates the main components in the design. The integrated high-voltage proprietary
controlled switch feeds a high-frequency transformer, and multiple secondary windings
tapped-off via a diode/capacitor circuit. The SMPS operates in discontinuous flyback
mode, so that energy is stored in the magnetizing inductance of the transformer during
the switch on-period, and transferred to the secondary circuits during the off-period. A
snubber arrangement is required across the transformer primary to dissipate the
transformer parasitic leakage energy and ensure that the switch voltage does not
exceed its maximum rating. The typical flyback MOSFET waveforms are shown in
Figure 2-14.
FIGURE 2-13:
AUXILIARY FLYBACK SMPS
SEC_17V
SEC_7V
Snubber/
Clamp
SEC_0V
PRI_13V
Vdc
Control
© 2008 Microchip Technology Inc.
PRI_7V
DS70320B-page 45
SMPS AC/DC Reference Design User’s Guide
FLYBACK SMPS MOSFET WAVEFORMS
700
0.7
600
0.6
500
0.5
400
0.4
300
0.3
200
0.2
100
0.1
0
0
2.5.1
2.5
5
7.5
Time ( s)
10
12.5
MOSFET Current (A)
MOSFET Voltage (V)
FIGURE 2-14:
0
15
Basic Design Methodology
The target specification for the auxiliary flyback SMPS is as follows:
•
•
•
•
•
Input voltage, VIN = 120-400 VDC
Primary Output Rail 1 = 7V @ 0.3A
Primary Output Rail 2 = 13V @ 0.15A
Secondary Output Rail 1 = 7V @ 0.3A
Secondary Output Rail 2 = 17V @ 0.45A
2.5.1.1
HVIC TECHNOLOGY
The total output power rating of the auxiliary power supply is 13.8W; therefore, a Power
Integrations TinySwitch (see Reference 13 in Appendix C. “References”) was
selected for the main power switch. These devices are intended specifically for
low-cost high-efficiency designs and operate with simple ON/OFF control rather than
more sophisticated PWM common to higher power SMPS. The TNY277G is a suitably
rated part for this wide input voltage application, and can be mounted directly on the
PCB without any additional cooling requirements.
2.5.1.2
TRANSFORMER
Based on the operating frequency and power throughput requirement, an EF20 ferrite
core pair is selected for the transformer design. The first step in the design is to select
the number of primary turns and transformer air-gap. Discontinuous flyback SMPS
output power is dictated by the energy stored in the primary magnetizing inductance of
the transformer and the switching frequency. The basic equation, ignoring losses, is
shown in Equation 2-35.
EQUATION 2-35:
Po = 12 L p I p2 fsw
DS70320B-page 46
© 2008 Microchip Technology Inc.
Hardware Design
The peak current in the primary is fixed for a given output power and the switch on-time
varies as a function of the DC input voltage, as shown in Equation 2-36.
EQUATION 2-36:
ton =
Lp I p
VDC
From the TNY277 data sheet, the maximum switching frequency is 140 kHz, and a
sensible maximum on-time is 4.5 μs. The minimum DC voltage is 120V, and the power
at the transformer primary is 17W, assuming ~ 80% efficiency. It is worth noting that this
is only a transient requirement, since once the dsPIC DSC device rails are established,
the PFC will boost the DC input voltage to 400V. Equation 2-35 and Equation 2-36 can
be rearranged to find the required primary magnetizing inductance of the transformer,
as shown in Equation 2-37.
EQUATION 2-37:
2
2
ton
VDC
f sw
Lp =
2 Po
Therefore, the target primary inductance is 1.2 mH and the peak primary current is
0.45A. The primary turns must be selected to ensure that the ferrite core losses are
around 0.5W for thermal reasons. The EF20 core has a cross-sectional area of 32 mm2
and a volume of 1490 mm3.
The core loss can be computed from Equation 2-38.
EQUATION 2-38:
⎛
⎞
Pcore
B =⎜
⎟
−4 1.59
⎝ 1.36 × 10 f sw Ve ⎠
0.365
where fsw is in kHz and B is in mT.
Therefore, the peak flux density is about 150 mT. The required number of primary turns
can be computed using Equation 2-39.
EQUATION 2-39:
B=
Lp I p
N p Ae
Using the above formula, Np is set at 110 turns, which requires a 0.5 mm air-gap in the
EF20 core to give 1.2 mH inductance. To minimize the leakage inductance of the
transformer, and hence the loss in the snubber/clamp, the primary winding is split into
two layers of 55 turns each. The rms current in the primary is given by Equation 2-40.
EQUATION 2-40:
i%p =
© 2008 Microchip Technology Inc.
D
Ip
3
DS70320B-page 47
SMPS AC/DC Reference Design User’s Guide
From Equation 2-36, the on-time for the switch at 400V is 1.35 μs and the duty cycle is
0.19. Therefore, the rms current in the primary is 0.11 Arms. A suitable winding wire
diameter is 0.16 mm with 5.5 Amm-2. The 100ºC resistance of this wire is 1.1 Ωm-1, and
from the mean turn length of the bobbin (41.2 mm), the predicted primary resistance is
5Ω. Therefore, the primary copper loss is 60 mW.
The secondary turns must be selected to ensure that the referred voltage across the
TNY277 does not exceed its maximum blocking voltage rating and for discontinuous
current operation given the operating duty cycle. The peak voltage across the TNY277
when energy is transferred to the secondary is given by Equation 2-41.
EQUATION 2-41:
V p = V DC +
Np
Ns
Vo
A sensible limit on the maximum switch voltage is 540V, so for the main 17V secondary
output, including a 0.8V diode forward voltage drop, the required turns ratio is 7.86.
Therefore, the secondary turns on the 17V winding is 14. A separate tapping at six
turns on this winding can be used for the 7V secondary. The secondary winding is
constructed using TEX-E wire to give the required 2500 Vrms galvanic voltage
isolation.
The time taken for the flyback transformer to be de-fluxed is given by Equation 2-42.
EQUATION 2-42:
toff =
Ns Lp I p
N pVo
Therefore, the off period is 3.9 μs, which is lower than the maximum available period
to ensure discontinuous operation under normal operating conditions.
The secondary peak and rms currents in the winding are given by the following formula
in Equation 2-43 and Equation 2-44.
EQUATION 2-43:
Is =
2TIo
toff
EQUATION 2-44:
i%s = 2
T
Io
3toff
Therefore, the secondary current in the main 17V output is 0.7 Arms. The secondary
winding resistance is 0.1Ω and the total copper loss in the secondary winding is
estimated to be 50 mW. Keeping a similar winding current density as the primary
means that a 0.4 mm wire gauge can be chosen for the secondary windings. The two
auxiliary supply windings on the primary side follow directly from the chosen
transformer turns ratio. The total transformer power dissipation is dominated by the iron
loss and is roughly 0.8W. This will lead to a temperature rise of 40ºC, based on the
published thermal characteristics of an EF20 transformer. Figure 2-15 illustrates the
flyback transformer construction.
DS70320B-page 48
© 2008 Microchip Technology Inc.
Hardware Design
FIGURE 2-15:
FLYBACK TRANSFORMER CONSTRUCTION
3 layers 0.05 mm Melinex
3 layers 0.05 mm Melinex
8 turns of 0.4 mm
6 turns of 0.4 mm
3 layers 0.05 mm Melinex
5 turns of 0.5 mm
6 turns of 0.5 mm
3 layers 0.05 mm Melinex
55 turns of 0.16 mm
2.5.1.3
OUTPUT CAPACITORS
The capacitors selected for outputs are 100 μF, 25V and 470 μF, 10V, for gate drive
voltage rails and the dsPIC DSC device voltage rails, respectively. The ripple current
rating for United Chemi-Con electrolytic capacitors are 0.34 Arms and 0.64 Arms,
respectively.
© 2008 Microchip Technology Inc.
DS70320B-page 49
SMPS AC/DC Reference Design User’s Guide
NOTES:
DS70320B-page 50
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Chapter 3. Software Design
3.1
OVERVIEW
Note:
Any libraries and source files associated with SMPS AC/DC Reference
Design are available by request from your local Microchip sales office. See
the Microchip Web site, or the last page of this document for contact
information.
The SMPS AC/DC Reference Design is controlled using two dsPIC DSCs as shown in
the block diagram in Figure 3-1.
The dsPIC DSC on the primary side (on the left of the isolation barrier in Figure 3-1)
controls the PFC Boost Converter and the Phase-Shift ZVT Converter. The dsPIC DSC
on the secondary side (on the right of the isolation barrier in Figure 3-1) controls the
Multi-Phase Buck Converter and the Single-Phase Buck Converter.
The secondary side dsPIC DSC also performs the function of measuring the output
voltage of the Phase-Shift ZVT Converter, and feeding back to the primary side dsPIC
DSC as a digital feedback signal.
FIGURE 3-1:
BLOCK DIAGRAM OF SMPS AC/DC REFERENCE DESIGN
12 VDC
30A
Isolation
Barrier
Rectified
Sinusoidal
Voltage
420 VDC
Phase-Shift ZVT Converter
EMI Filter
and Bridge
Rectifier
PFC
Boost
Converter
ZVT
Full-Bridge
Converter
Multi-Phase
Buck Converter
3.3 VDC
69A
Synchronous
Rectifier
85-265 VAC
45-65 Hz
dsPIC33FJ16GS504
© 2008 Microchip Technology Inc.
OptoCoupler
Single-Phase
Buck Converter
5 VDC
23A
dsPIC33FJ16GS504
DS70320B-page 51
SMPS AC/DC Reference Design User’s Guide
3.2
STRUCTURE OF THE CONTROL SOFTWARE
The control software for the SMPS AC/DC Reference Design essentially follows a
single basic structure as shown in Figure 3-2.
FIGURE 3-2:
FLOWCHART OF CONTROL SOFTWARE
Start
Initialization
Enable Peripherals
Soft-Start
Idle Loop
(Normal Operation)
Yes
No
Fault Present?
Fault Loop
ADC Interrupt
The control software uses a mixture of C programming and Assembly programming. All
time-critical functions are written in Assembly language. The main loop, peripheral
setup routines, initialization routines and non-critical functions are all written using the
C programming language.
The SMPS AC/DC Reference Design comprises of two separate projects, namely:
• Primary: This project contains the complete code for the primary side of the
SMPS AC/DC Reference Design. This includes the control software for the PFC
Boost Converter and the Phase-Shift ZVT Converter.
• Secondary: This project contains the complete code for the secondary side of the
SMPS AC/DC Reference Design. This includes the control software for the
Single-Phase Buck Converter and Multi-Phase Buck Converter, and also includes
code for the ZVT output voltage measurement and digital voltage feedback.
All of the functional blocks shown in Figure 3-2 are common to both projects. Brief
descriptions of each functional block are provided in subsequent sections.
DS70320B-page 52
© 2008 Microchip Technology Inc.
Software Design
3.2.1
Initialization Routine
The initialization routines are called from the main program at the start of execution. All
peripherals including PWM, ADC, analog comparator, UART, I2C™ and Timers are
configured in this step. It is important that none of the peripherals are enabled before
the entire peripheral configuration is completed.
In addition to configuring the peripherals, all required interrupts and interrupt priorities
are configured in the initialization step. Memory allocation for control loop variables is
also done during the initialization stage.
3.2.2
Peripheral Enable Routine
After configuring all peripheral modules that are used in the project, we enable them in
the correct sequence. Since the PWM output directly affects the output of the system,
it is important to enable it after all other peripherals have been enabled.
3.2.3
Soft-Start Routine
Each individual stage of the SMPS AC/DC Reference Design employs a controlled
soft-start routine. This routine ramps individual output stages to the desired output
voltage.
3.2.4
Fault Check Routine
The fault check routine is used to check for faults that have occurred in the system. If
a fault has occurred, the system has to be shut down. To do this, the fault loop is called
and used to disable all active modules, such as the ADC and PWM, and to visually
display the fault on the LEDs.
The PWM module on the dsPIC33FJ16GS504 has built-in fault inputs that ensure a
fast PWM shutdown in order to prevent damage to the system and downstream
electronics. After the PWM is shutdown, the program execution jumps to the fault loop.
3.2.5
ADC Interrupt Service Routine
The ADC Interrupt Service Routine (ISR) is the heart of the control software. All control
loops are executed in the ISR. Since faster control loop execution is desired for the best
system performance, functions executed in this routine are written in Assembly
language. The ADC ISR has the highest priority of execution.
The ADC module is configured to generate interleaved interrupt requests in order to
execute multiple control loops within the same ISR.
The implementation of the control software for each stage of the SMPS AC/DC
Reference Design contains all the blocks described above. However, there are subtle
differences in the implementation for each stage.
Specific details for each stage of the design are covered in subsequent sections of this
user’s guide.
© 2008 Microchip Technology Inc.
DS70320B-page 53
SMPS AC/DC Reference Design User’s Guide
3.3
PRIMARY SIDE CONTROL SOFTWARE (PFC_ZVT)
The PFC Boost Converter and Phase-Shift ZVT Converter follow a similar control
scheme. However, there are significant differences in the operation of these two
converters. These differences will be explained in the description of the control
software for each converter.
3.3.1
PFC Boost Converter Control Software
3.3.1.1
PFC CONTROL SCHEME
The control scheme implemented for the PFC Boost Converter is shown in Figure 3-3.
FIGURE 3-3:
PFC CONTROL SCHEME
Rectified
AC Mains
Voltage
Voltage
Reference
Voltage Error
Compensator
Σ
+
Calculated
Current
Reference
χ
+
-
VOUT
Current Error
Compensator
Σ
-
PWM
PFC
Choke
1
|VAC|MEAN
PFC
Current
Sense
V2
Current
Feedback
1011001010
S&H
1001011011
Voltage Feedback
VOUT
Sense
ADC
S&H
The PFC Boost Converter uses an outer voltage loop and inner current loop control
scheme. The output of the voltage error compensator is multiplied by a function of the
rectified AC mains voltage to generate a sinusoidal current reference.
An additional feed-forward term is introduced, |VAC|MEAN, at the output of the voltage
error compensator to make the control loop immune to fluctuations in the AC input
voltage. This feed-forward term ensures that the PFC Boost Converter always delivers
the correct output power for the entire input voltage range.
The PFC voltage and current error compensators are both implemented as
Proportional-Integral (PI) systems with excess error compensation. The compensator
functions are math intensive routines and utilize the DSP engine of the dsPIC DSC.
The output of the PFC Current compensator modifies the PWM duty cycle to maintain
a constant output voltage and also a sinusoidal input current waveform.
Both the current and voltage compensators are executed in the ADC ISR. The current
control loop is executed at a much faster rate compared to the voltage control loop.
DS70320B-page 54
© 2008 Microchip Technology Inc.
Software Design
3.3.1.1.1
Digital PFC Implementation Using the dsPIC DSC
Figure 3-4 shows the hardware resources utilized on the primary side dsPIC DSC for
Power Factor Correction.
FIGURE 3-4:
dsPIC® DSC RESOURCE ALLOCATION FOR PFC BOOST CONVERTER
IPFC
VHV_BUS
|VAC|
k1
k3
VAC
FET Driver
k2
ADC
Channel
ADC Channel
PWM
Output
ADC
Channel
dsPIC33FJ16GS504
Table 3-1 lists the resources used on the dsPIC DSC for implementing the PFC control
scheme shown in Figure 3-3.
TABLE 3-1:
dsPIC® DSC RESOURCES FOR PFC BOOST CONVERTER
Description
VHV_BUS
Type of Signal
dsPIC® DSC Resource Used
Analog Input
AN5
PFC Current (IPFC)
Analog Input
AN4
|VAC| Sense
Analog Input
AN3
MOSFET Gate Drive
Drive Output
PWM4L
The control of the PFC Boost Converter is obtained by varying the duty cycle of the
PWM signal. Only one pin of the PWM is utilized for the PFC control scheme, and
therefore the PWM module is configured for independent output mode. The frequency
of the PWM is determined by the hardware design. It is configured to be approximately
125 kHz.
The Analog Inputs AN4 and AN5 are configured to sample simultaneously. A
conversion is triggered on both AN4 and AN5 once every 3 PWM cycles, and the
current loop is executed on every conversion. The voltage loop is executed only once
in 15 current loop executions.
The Analog Input AN3 measures the AC Input voltage, which is used for generating a
sinusoidal current reference. In the SMPS AC/DC Reference Design, the current
reference value is calculated when the current loop is executed.
© 2008 Microchip Technology Inc.
DS70320B-page 55
SMPS AC/DC Reference Design User’s Guide
3.3.2
Phase-Shift ZVT Control Scheme
3.3.2.1
ZVT RESOURCE ALLOCATION
The control scheme for the Phase-Shift ZVT Converter is shown in Figure 3-5. A
schematic of the Phase-Shift ZVT Converter is shown in Figure 3-6.
FIGURE 3-5:
PHASE-SHIFT ZVT CONVERTER CONTROL SCHEME
VOUT
Rectifier
Voltage
Reference
Voltage Error
Compensator
Σ
+
Update
PWM
Phase
-
ZVT
Transformer
S&H
Overcurrent 1011001010
Protection
1001011011
ADC
ZVT Current Sense
for overcurrent
protection
S&H
Optocoupler
Voltage
Feedback
UART
RX
UART
TX
1011001010
ADC
S&H
VOUT
Sense
The ZVT converter uses voltage mode control to maintain a constant output voltage.
The circuit configuration of the ZVT converter has the output voltage (the parameter to
be controlled) on the secondary side of the isolation barrier (refer to Figure 3-8).
The SMPS AC/DC Reference Design implements the ZVT voltage feedback by
measuring the output voltage using the secondary side dsPIC DSC. The Most
Significant 8 bits of data are transmitted back to the primary side dsPIC DSC through
a UART communication channel.
The data received by the primary side dsPIC DSC is right-shifted by two bits (for 10-bit
data) and is compared with the voltage reference to produce the voltage error. The
voltage error compensator is then executed in the ADC ISR.
The output of the voltage error compensator is feed into the phase-shifted PWM. The
Phase-Shift ZVT converter uses the unique phase-shifting capability of the PWM
module in the dsPIC33FJ16GS504. The phase of the PWM signal can be modified by
simply writing the new value in the appropriate Special Function Register in the PWM
module.
DS70320B-page 56
© 2008 Microchip Technology Inc.
Software Design
FIGURE 3-6:
ZVT PHASE-SHIFT CONVERTER
CR
CR
VPRI
Q1
Q4
IPRI
VIN
LR
Q2
CR
CR
Q3
VSEC
Q5
VOUT
Q6
Current sensing for the Phase-Shift ZVT Converter uses two dedicated analog inputs
to measure the primary transformer current in both directions. The two analog inputs
are tied to the same current signal, but are sampled at opposite current peaks. The
precise triggering instants are in Figure 3-7 along with the expected current waveform.
The current in the positive and negative directions must be measured and checked for
any imbalance. If the currents in the two directions are not balanced, it may cause a
phenomenon called “flux walking” in the ZVT transformer. Flux walking must be
prevented since it may lead to transformer saturation and subsequent damage to the
system hardware.
© 2008 Microchip Technology Inc.
DS70320B-page 57
SMPS AC/DC Reference Design User’s Guide
FIGURE 3-7:
ZVT CURRENT SAMPLING INSTANTS
Q1PWM
(1)
(1)
Q2PWM
(1)
Q3PWM
Q4PWM
Ipk
Ip
ADC Trigger generated
by PWM2 at beginning
of PWM cycle
Note 1:
ADC Trigger generated
by PWM1 at duty cycle
minus phase shift
The shaded regions represent the times when power is transferred from the primary side to the
secondary side. This region is sometimes referred to as the “effective” ZVT duty cycle.
The voltage compensator is implemented as a Proportional-Integral-Derivative
(PID) function. The voltage compensator is executed in the ADC ISR.
DS70320B-page 58
© 2008 Microchip Technology Inc.
Software Design
3.3.2.2
PHASE-SHIFT ZVT IMPLEMENTATION USING THE dsPIC DSC
dsPIC® DSC RESOURCE ALLOCATION FOR PHASE-SHIFT ZVT CONVERTER
FIGURE 3-8:
Isolation
Barrier
VHV_BUS
VOUT
IZVT
FET
Driver
FET
Driver
PWM
PWM
PWM
PWM
k1
ADC
Channel
k2
FET
Driver
ADC
Channel
ADC
Channel
PWM
dsPIC33FJ16GS504
dsPIC33FJ16GS504
PWM
UART
RX
UART
TX
Table 3-2 lists the resources used on the primary side dsPIC DSC for the different
feedback and control signals required for the Phase-Shift ZVT Converter control
scheme. Table 3-3 lists resources used on the secondary side dsPIC DSC, also
required for the Phase-Shift ZVT Converter control scheme.
TABLE 3-2:
PRIMARY SIDE dsPIC® DSC RESOURCES FOR PHASE-SHIFT
ZVT CONVERTER
Type of Signal
dsPIC® DSC Resource
Used
ZVT Current 1 (IZVT1)
Analog Input
AN0
ZVT Current 2 (IZVT2)
Analog Input
AN2
VOUT Feedback
UART Input
U1RX
Full-Bridge Gate Drive
Drive Output
PWM1H, PWM1L,
PWM2H, PWM2L
Synchronous Rectifier Gate Drive
Drive Output
PWM3H, PWM3L
Description
TABLE 3-3:
SECONDARY SIDE dsPIC® DSC RESOURCES FOR
PHASE-SHIFT ZVT CONVERTER
Type of Signal
dsPIC® DSC Resource
Used
Voltage Sense (VOUT)
Analog Input
AN5
VOUT Feedback
UART Output
U1TX
Description
The Phase-Shift ZVT Converter is controlled by modifying the phase of the PWM drive
signals of one leg of the Full-Bridge relative to the drive signals of the other leg of the
Full-Bridge.
© 2008 Microchip Technology Inc.
DS70320B-page 59
SMPS AC/DC Reference Design User’s Guide
As specified in Table 3-2, PWM1H, PWM1L, PWM2H, and PWM2L are the PWM
signals used for switching the Full-Bridge MOSFETs. PWM1H and PWM1L control one
leg of the Full-Bridge, while PWM2H and PWM2L control the second leg of the
Full-Bridge.
PWM1 and PWM2 are configured to operate in the complementary PWM mode and
approximately 250 kHz switching frequency. The duty cycle of these PWM signals is
fixed at 50%. Some dead time is also inserted to prevent shoot-through.
PWM3 is used for driving the synchronous rectifier MOSFETs on the secondary side
of the ZVT Transformer. PWM3 is also configured as a complementary mode PWM
signal with dead time. The PWM3 signal is configured identically to that of PWM1. The
output of the control loop directly modifies the phase of PWM2 to accomplish the
control of the output.
AN0 and AN2 both measure the ZVT current, but each input is sampled on opposite
peaks of the current signal. The conversion result of AN0 is used for the ZVT
overcurrent fault protection. The voltage feedback is received on the U1RX pin of the
primary side dsPIC DSC.
The voltage loop is executed every two PWM periods, but the measured voltage is only
updated when data is received by the UART. This UART data reception is
asynchronous to the PWM drive signals.
3.3.3
Primary Side Software Time Management
Both the PFC and ZVT converters are controlled using a single dsPIC DSC. The
execution rates are carefully chosen to effectively utilize the available processing
bandwidth of the dsPIC DSC. The flexible PWM-ADC trigger feature of the
dsPIC33FJ16GS504 enables precise sampling of analog signals and interleaved
control loop execution.
Figure 3-9 shows the interleaved control loop execution as implemented on the primary
side control software on the SMPS AC/DC Reference Design.
FIGURE 3-9:
INTERLEAVED CONTROL LOOP EXECUTION
PFC Current Trigger:
ZVT Trigger:
Once every 3 PFC Cycles Once every 2 ZVT Cycles
PFC Current Trigger:
Once every 3 PFC Cycles
PWM1H
PWM1L
Phase
Shift
PWM2H
PWM2L
Sample and
Convert
AN4, AN5
PWM4L
Execute
PFC
Current
Loop
Sample and Convert
AN4, AN5
Idle Loop
Sample and Convert
AN0, AN1
DS70320B-page 60
Execute
PFC
Current
Loop
Execute
ZVT
Voltage
Loop
Execute
ZVT
Voltage
Loop
Execute
ZVT
Voltage
Loop
ADC Pair 0
has highest
priority
© 2008 Microchip Technology Inc.
Software Design
3.4
SECONDARY SIDE CONTROL SOFTWARE (DC_DC)
3.4.1
Single-Phase Buck Converter
3.4.1.1
SINGLE-PHASE BUCK CONVERTER CONTROL SCHEME
The Single-Phase Buck Converter on the SMPS AC/DC Reference Design uses peak
current mode control. The control scheme is shown in Figure 3-10.
The control loop is implemented by utilizing the analog comparator module. The Buck
MOSFET current is sensed using a current transformer and fed directly to the analog
comparator input.
FIGURE 3-10:
SINGLE-PHASE BUCK CONVERTER CONTROL SCHEME
Calculated
Current
Reference
VREF
+
Analog
Comparator
Voltage
Reference
+
Current-Limit
Shutdown
Voltage Error
Compensator
Σ
Buck
Current
Sense
VOUT
Buck
Inductor
PWM
-
1001011011
Voltage Feedback
V OUT
Sense
ADC
S&H
The measured output voltage is compared with the reference to produce the voltage
error. The voltage error compensator is then executed and a current reference value is
obtained. The current control loop is implemented on the dsPIC DSC using the analog
comparator by varying the programmable threshold in software.
The analog comparators on the dsPIC33FJ16GS504 have built-in programmable
Digital-to-Analog Converters (DACs) that determine the comparator threshold. The
calculated current reference is used to set a new threshold for the analog comparator.
When the inductor current signal exceeds the programmed threshold, the comparator
terminates the PWM pulse. This termination of the PWM pulse effectively modifies the
ON time for the PWM signal to control the output voltage.
The Voltage Error Compensator is implemented as a PI function in the ADC ISR.
3.4.1.2
SINGLE-PHASE BUCK CONVERTER IMPLEMENTATION USING THE
dsPIC DSC
The resources used on the secondary side dsPIC DSC for the Single-Phase Buck
Converter are summarized in Table 3-4.
TABLE 3-4:
dsPIC® DSC RESOURCE ALLOCATION FOR SINGLE-PHASE
BUCK CONVERTER
Description
Buck Current
Type of Signal
dsPIC® DSC Resource Used
Analog Comparator Input
CMP1A, AN0
Buck Voltage (VOUT)
Analog Input
AN1
Single-Phase Synchronous
Buck Gate Drive
Drive Output
PWM4H, PWM4L
© 2008 Microchip Technology Inc.
DS70320B-page 61
SMPS AC/DC Reference Design User’s Guide
The output voltage is measured using the analog input AN1. The analog comparator
input CMP1A is connected to the output of the current transformer. The output voltage
is controlled by varying the duty cycle of PWM4.
The PWM4 pair is operated in Complementary mode with dead time. The switching
frequency is approximately 500 kHz. The duty cycle is controlled directly by the built-in
Cycle-by-Cycle Current-Limit mode and the analog comparator.
When the current-sense signal at the input of the analog comparator exceeds the
programmed comparator threshold, the PWM output is immediately terminated for the
remainder of the PWM cycle.
The Single-Phase Buck Converter circuitry is designed to operate in continuous
conduction mode at load currents greater than approximately 3A. If the Single-Phase
Buck Converter is operated in Discontinuous Conduction mode, the freewheeling
MOSFET is disabled through software.
At no load and light load current (< 3A), the PWM output may enter a “burst” mode. This
is caused by a low demand for load current by the converter in this range load current.
The voltage control loop is executed in the ADC ISR every two PWM cycles.
3.4.2
Multi-Phase Buck Converter
3.4.2.1
MULTI-PHASE BUCK CONVERTER CONTROL SCHEME
Voltage mode control is used for controlling the output of the Multi-Phase Buck
Converter on the SMPS AC/DC Reference Design. As shown in Figure 3-11, the
control scheme only implements a single control loop.
FIGURE 3-11:
MULTI-PHASE BUCK CONVERTER CONTROL SCHEME
Voltage
Reference
Σ
+
-
Voltage
Error
Compensator
PWM
Update
PWM
Duty
Cycle
Phase 1
Inductor
VOUT
PWM
Phase 2
Inductor
PWM
Phase 3
Inductor
1001011011
VOUT
Sense
Voltage Feedback
ADC
S&H
The output voltage is compared with the reference and results in a voltage error, which
is fed as an input to the voltage error compensator. The output of the voltage error
compensator modifies the duty cycle of all phases of the Multi-Phase Buck Converter.
The voltage error compensator is implemented as a PID function that is implemented
in the ADC ISR.
The Multi-Phase converter comprises of three individual phases, but the output is
controlled by a single duty cycle that identically drives the three phases. The PWM
drive signals for each phase are phase shifted by 120 degrees using the built-in PWM
phase-shifting feature available on the dsPIC33FJ16GS504. The PWM drive signals
for the Multi-Phase Buck Converter are shown in Figure 3-12.
DS70320B-page 62
© 2008 Microchip Technology Inc.
Software Design
FIGURE 3-12:
MULTI-PHASE BUCK CONVERTER PWM DRIVE SIGNALS
Drive Signals are
Phase-Shifted by 120º
3.3V Output
12V Input
Q1
Q2
120º 120º 120º
Q1
Q3
Q4
Q3
Q5
Q5
Q6
3.4.2.2
GND
MULTI-PHASE BUCK CONVERTER IMPLEMENTATION USING THE
dsPIC DSC
Table 3-5 summarizes the resource allocation for the Multi-Phase Buck Converter.
TABLE 3-5:
dsPIC® DSC RESOURCE ALLOCATION FOR MULTI-PHASE
BUCK CONVERTER
Type of Signal
dsPIC® DSC Resource Used
Buck 1 Current
Analog Comparator Input
CMP2A
Buck 2 Current
Analog Comparator Input
CMP3A
Buck 3 Current
Analog Comparator Input
CMP4A
Description
Buck Voltage
Analog Input
AN3
Multi-Phase Synchronous
Gate Drive
Drive Outputs
PWM1H, PWM1L,
PWM2H, PWM2L,
PWM3H, PWM3L
The output voltage is measured from the analog input AN3. As this converter uses
voltage mode control, there is no need for current measurement. However, overcurrent
protection must be provided for each individual phase. Overcurrent protection is
implemented using the analog comparators on the dsPIC33FJ16GS504. CMP2A,
CMP3A and CMP4A are used for the overcurrent sensing for the Multi-Phase Buck
Converter.
Each of the three phases is driven by a pair of complementary PWM signals. The PWM
module on the dsPIC33FJ16GS504 provides a built-in mode to generate a pair of
complementary PWM outputs with dead time insertion. The PWM module also has a
feature to generate a Master Period and Master Duty Cycle for multiple outputs. PWM1,
PWM2, and PWM3 are all configured for a PWM switching frequency of approximately
500 kHz, and complementary mode operation with dead time.
PWM2 is phase advanced by 120 degrees from PWM1, and PWM3 is phase advanced
by 120 degrees from PWM2 (or 240 degrees from PWM1). The voltage control loop is
executed every two PWM cycles. The control loop is called from the ADC ISR.
The output of the voltage control loop is used to directly modify the PWM Master Duty
Cycle to control the output voltage.
© 2008 Microchip Technology Inc.
DS70320B-page 63
SMPS AC/DC Reference Design User’s Guide
3.4.3
Secondary Side Software Time Management
The Single-Phase Buck Converter and Multi-Phase Buck Converter are both controlled
digitally by the same dsPIC DSC. Both converters operate at the same switching
frequency.
The two control loops are executed in an interleaved manner as shown in Figure 3-13.
The execution rate for each control loop is once every other PWM cycle. The execution
rate is determined by the execution times of each control loop.
The ADC ISR assumes the highest priority of all user software. Other interrupts are
assigned lower priority than the ADC interrupt, and all auxiliary software functions are
performed in the “Idle loop” (when the ADC interrupt is not being serviced).
FIGURE 3-13:
INTERLEAVED CONTROL LOOP EXECUTION FOR SINGLE-PHASE AND
MULTI-PHASE BUCK CONVERTERS
3.3V Buck
Voltage Trigger
5V Buck
Voltage Trigger
5V Buck
Voltage Trigger
5V Buck
Voltage Trigger
3.3V Buck
Voltage Trigger
3.3V Buck
Voltage Trigger
PWM1H
PWM1L
PWM2H
PWM2L
PWM3H
PWM3L
PWM4H
PWM4L
Execute
5V Buck
Control
Loop
Sample
and
Convert
AN0, AN1
Idle
Loop
Execute
3.3V
Buck
Control
Loop
Sample
and
Convert
AN2, AN3
3.5
Execute
5V Buck
Control
Loop
Execute
5V Buck
Control
Loop
Idle
Loop
Sample Idle
Loop
and
Convert
AN0, AN1
Sample
and
Convert
AN2, AN3
Execute
3.3V
Buck
Control
Loop
Idle
Loop
Sample
Idle
and
Convert Loop
AN0, AN1
Execute
3.3V
Buck
Control
Loop
Sample
and
Convert
AN2, AN3
AUXILIARY SOFTWARE ROUTINES
3.5.1
Output Sequencing
Many applications require specific turn-on and turn-off sequencing of power supplies
to ensure correct operation of the downstream electronics. The SMPS AC/DC
Reference Design implements power-on sequencing in the following order:
1.
2.
3.
4.
5.
DS70320B-page 64
Initial start-up delay (allows all circuitry to stabilize after power-up).
PFC Converter ramps to around 420V.
Phase-Shift ZVT Converter ramps to 12V.
Single-Phase Buck Converter ramps to 5V.
Multi-Phase Buck Converter ramps to 3.3V.
© 2008 Microchip Technology Inc.
Software Design
3.5.2
Soft-Start Routine
Each individual stage of the SMPS AC/DC Reference Design employs a controlled
soft-start routine. At power-up, all reference set points are configured to produce 0V
output. Once the power-on delay has lapsed, the outputs begin their soft-start where
the reference set point is incremented until the desired output voltage is reached.
3.5.3
Overtemperature Protection
Temperature sensors are provided on the SMPS AC/DC Reference Design in two
positions. Overtemperature protection must be enabled to prevent damage to the
system in the event of:
• Insufficient airflow in the system caused by a failure of the cooling fan
• Operation of the system at a high ambient temperature
The implementation detail for each sensor is described in the following sections.
3.5.3.1
PCB OVERTEMPERATURE PROTECTION
One of the PCB temperature sensors is located on the secondary side in the middle of
the four Buck phases. The other temperature sensor is located on the primary side just
below the Boost inductor (L4). An analog temperature sensor is chosen that outputs an
analog voltage proportional to the measured temperature.
The output of the temperature sensor is connected to analog input AN8 on the
secondary side dsPIC DSC, and AN10 on the primary side dsPIC DSC. The PCB
temperature is measured in the ADC ISR and checked for overtemperature protection
in the fault loop. The maximum temperature set point is configured to approximately
90°C. If the measured temperature exceeds the maximum set point, a fault is
generated and the PWM outputs are turned OFF.
3.5.4
Input AC Undervoltage/Overvoltage Protection
The PFC Boost Converter is designed to operate normally for input voltages in the
range 85V–265V. In the event of an undervoltage condition, the circuit components will
undergo excessive stress due to the additional current drawn by the system to deliver
maximum output power. Therefore, undervoltage and overvoltage faults must be
implemented to prevent damage to the system and load. In the event of an overvoltage
condition, the input voltage may exceed the device ratings.
There are instances when the power grid may exhibit momentary voltage fluctuations,
which must be ignored by the system.
The input AC voltage protection is implemented in software by calculating the average
input voltage in the ADC ISR. The average input voltage is calculated as a part of the
PFC control scheme, and is checked in the fault loop to detect a sustained
undervoltage/overvoltage condition.
The first time an undervoltage/overvoltage condition is detected, a counter is
incremented. If the undervoltage/overvoltage condition remains for an extended period
of time, a fault is generated and the outputs are turned OFF.
3.5.5
Fault Source Identification
If a Fault condition is detected, it is often required that the system be turned OFF to
prevent damage. The PWM module on the dsPIC33FJ16GS504 has a built-in latched
fault mode. Using the latched fault mode, certain faults will immediately disable the
PWM outputs with no software overhead.
Each fault is assigned a fault ID to indicate the source of the last fault that occurred. A
visual indication is also provided using LEDs on both the primary and secondary side
of the SMPS AC/DC Reference Design.
© 2008 Microchip Technology Inc.
DS70320B-page 65
SMPS AC/DC Reference Design User’s Guide
The LED flashes the same number of times as the fault ID of the source that caused
the fault. The source of the fault can be decoded using the values in Table 3-6 and
Table 3-7.
TABLE 3-6:
FAULT SOURCE INDICATION ON PRIMARY SIDE
Fault ID
TABLE 3-7:
1
PCB overtemperature
2
PFC output overvoltage
3
ZVT overcurrent
4
AC overvoltage
5
AC undervoltage
6
Secondary UART failure
7
System overload
FAULT SOURCE INDICATION ON SECONDARY SIDE
Fault ID
DS70320B-page 66
Source of Fault
Source of Fault
1
12V Buck overvoltage
2
12V Buck undervoltage
3
Multi-phase overcurrent
4
PCB overtemperature
5
Single-phase overcurrent
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Chapter 4. System Operation
This chapter describes the system setup and operation of the SMPS AC/DC Reference
Design.
4.1
SYSTEM SETUP
4.1.1
Recommended Test Equipment
The following list of test equipment is recommended for complete evaluation of the
SMPS AC/DC Reference Design and/or development of software.
•
•
•
•
•
Oscilloscope
High Voltage Probe (100:1 attenuation ratio) or Differential Probe
10A AC Current Probe
Power Quality Meter
DC Electronic Load (350W or higher, and should have capability to load at least
two outputs simultaneously)
• 0V-265V Variac or Programmable AC Source (500W or higher)
• Two Digital Multimeters
4.1.2
Functional Blocks of the System
Figure 4-1 shows a block diagram of the SMPS AC/DC Reference Design. Table 4-1
describes the inputs and outputs of the system and also displays the location of the
isolation barrier. To assist in identifying each functional block on the SMPS AC/DC
Reference Design, Figure 4-2 shows a top-view of the system with each block called
out with dotted lines.
FIGURE 4-1:
SMPS AC/DC REFERENCE DESIGN BLOCK DIAGRAM
12 VDC
30A
Isolation
Barrier
Rectified
Sinusoidal
Voltage
420 VDC
Phase-Shift ZVT Converter
EMI Filter
and Bridge
Rectifier
PFC
Boost
Converter
ZVT
Full-Bridge
Converter
Multi-Phase
Buck Converter
3.3 VDC
69A
Synchronous
Rectifier
85-265 VAC
45-65 Hz
dsPIC33FJ16GS504
© 2008 Microchip Technology Inc.
Optocoupler
Single-Phase
Buck Converter
5 VDC
23A
dsPIC33FJ16GS504
DS70320B-page 67
SMPS AC/DC Reference Design User’s Guide
FIGURE 4-2:
FUNCTIONAL BLOCKS OF SMPS AC/DC REFERENCE DESIGN
10
7
9
8
4
5
12
2
1
1.
2.
3.
4.
5.
6.
11
3
6
EMI Filter
Single-Phase Converter
Multi-Phase Converter
Primary Side Controller
Secondary Side Controller
Bridge Rectifier
TABLE 4-1:
7.
8.
9.
10.
11.
12.
PFC Boost Converter
ZVT Full-Bridge Converter
Synchronous Rectifier
12V Output (Intermediate Bus)
3.3V Output
5V Output
INPUT AND OUTPUT ELECTRICAL SPECIFICATIONS
Functional Block
Input
Output
EMI Filter
85-265V AC, 45-65 Hz
85-265V AC, 45-65 Hz
Bridge Rectifier
85-265V AC, 45-65 Hz
120-374V DC (unregulated)
PFC Boost Converter
120-374V DC (unregulated)
420V DC
Full-Bridge Converter
420V DC
N/A
Synchronous Rectifier
N/A
12V DC
Multi-Phase Buck Converter
12V DC
3.3V DC
Single-Phase Buck Converter
12V DC
5V DC
4.1.3
Safety Isolation Information
The SMPS AC/DC Reference Design provides safety isolation to protect users and
downstream electronics from the input AC Mains voltage. The location of the isolation
boundary on the SMPS AC/DC Reference Design is displayed with a dotted line in
Figure 4-3.
DS70320B-page 68
© 2008 Microchip Technology Inc.
System Operation
FIGURE 4-3:
ISOLATION BOUNDARY ON SMPS AC/DC REFERENCE
DESIGN
Isolation Boundary
+12V
GND
+3.3V
GND
+5V
GND
NOTICE
During testing and evaluation of the SMPS AC/DC Reference Design, no equipment
should be connected across the isolation boundary. This applies to oscilloscope
probes, multimeters, and programmers/debuggers. Under no circumstances should the
Live_GND (on the primary or “live” side) and GND (on the secondary or “isolated” side)
be tied together.
As a general rule, the primary (live) side and the secondary (isolated) side should
always be tested independently with no connections across the isolation boundary.
Before connecting oscilloscope probes to the SMPS AC/DC Reference Design, ensure
that the oscilloscope is isolated from the SMPS AC/DC Reference Design.
Table 4-2 lists the location where each functional block resides with respect to the
isolation barrier.
TABLE 4-2:
LOCATION OF FUNCTIONAL BLOCK WITH RESPECT TO
ISOLATION BARRIER
Live or Isolated Side?
Functional Block
Power Circuit
Control Circuit
EMI Filter
Primary (Live)
N/A
Bridge Rectifier
Primary (Live)
N/A
PFC Boost Converter
Primary (Live)
Primary (Live)
Full-Bridge Converter
Primary (Live)
Primary (Live)
Synchronous Rectifier
Secondary (Isolated)
Primary (Live)
Multi-Phase Buck Converter
Secondary (Isolated)
Secondary (Isolated)
Single-Phase Buck Converter
Secondary (Isolated)
Secondary (Isolated)
© 2008 Microchip Technology Inc.
DS70320B-page 69
SMPS AC/DC Reference Design User’s Guide
4.1.4
System Connections
4.1.4.1
INPUT CONNECTIONS
The AC input connector (J16) is shown in Figure 4-4. The SMPS AC/DC Reference
Design has a transparent lid (not shown in pictures) with a 12V fan mounted on it. There
are three holes provided in the lid to fasten the connection screws on the AC input
connector (J16).
Ensure that the power chord is not connected to the Variac or programmable AC source
(or AC Mains). Connect the AC cord with terminal lugs to the AC input connector in the
configuration shown in Figure 4-4.
FIGURE 4-4:
SMPS AC/DC REFERENCE DESIGN INPUT CONNECTIONS
Live
Earth
(No connection)
Neutral
This is the minimum connection required to power up the SMPS AC/DC Reference
Design. However, other connections are recommended for detailed testing and
evaluation of the system.
4.1.4.2
OUTPUT CONNECTIONS
Ensure that the SMPS AC/DC Reference Design is not powered. Connect DC
Electronic Loads (if available) to the 5V and 3.3V output terminals shown in Figure 4-5.
FIGURE 4-5:
SMPS AC/DC REFERENCE DESIGN OUTPUT CONNECTIONS
+12V
GND
+3.3V
GND
+5V
GND
A DC electronic load can be used to adjust the output power delivered by the SMPS
AC/DC Reference Design. If a DC electronic load is not available, a rheostat or
resistors with sufficient power ratings may also be used for loading the SMPS AC/DC
Reference Design.
DS70320B-page 70
© 2008 Microchip Technology Inc.
System Operation
4.1.4.3
PROGRAMMING CONNECTIONS
The SMPS AC/DC Reference Design comes pre-programmed with the control software
and does not require a programmer to be connected to the system. However, ICSP™
headers are provided on the primary and secondary side for software development and
testing.
Figure 4-6 shows the location of the primary side programming header (J2 on the
control board).
FIGURE 4-6:
PRIMARY SIDE PROGRAMMING HEADER
Primary Side
ICSP™ Header
CAUTION
The primary side ICSP header (J2) is not isolated from the AC Mains. An isolated USB
hub must be used to connect the programmer to the computer's USB port. If the power
supply of the SMPS AC/DC Reference Design is not isolated before programming the
device, ensure that the computer is isolated or removed from the grid.
Figure 4-7 shows the location of the secondary side programming header (J1 on the
control board).
FIGURE 4-7:
SECONDARY SIDE PROGRAMMING HEADER
Secondary Side
ICSP™ Header
The secondary side ICSP header is isolated from the AC Mains, so there are no special
precautions necessary when programming the device.
© 2008 Microchip Technology Inc.
DS70320B-page 71
SMPS AC/DC Reference Design User’s Guide
4.2
SYSTEM OPERATION
4.2.1
System Power-Up
Once the input and output connections as described in Section 4.1.4 “System
Connections” are completed, the mains voltage can be applied to the SMPS AC/DC
Reference Design.
There are three power-on indicator LEDs on the system. One LED (D38) on the power
board near the AC input terminals, and two more on the control board (one on the
primary side (LED1), and one on the secondary side (LED2)).
There will be a short delay before the 12V, 5V, and 3.3V outputs are ON because of the
soft-start routines and output sequencing scheme implemented on each stage of the
SMPS AC/DC Reference Design. Details of the soft-start routine and output
sequencing are provided in Chapter 3. “Software Design”.
As soon as the 12V output is ON, the cooling fan mounted on the lid will start running.
LEDs are provided near each output terminal to indicate that the output is ON.
Faults are indicated on the control board with two LEDs (one on the primary side (D34)
and one on the secondary side (D33)). Details of faults are provided in Chapter
3. “Software Design”.
4.2.2
System Evaluation and Testing
4.2.2.1
INPUT PERFORMANCE TESTING
The input specifications of the SMPS AC/DC Reference Design can be tested by using
a power meter. The voltage is measured directly across the Live and Neutral terminals
on the SMPS AC/DC Reference Design. The input current is measured by sensing the
current through either the Live or Neutral line. Figure 4-8 shows two separate power
meter connections depending on the type of power meter used.
FIGURE 4-8:
TYPICAL POWER METER CONNECTIONS
Series Ammeter Power Quality Meter
+
I
–
+
V
–
AC Mains
L
L
G
G
N
N
SMPS AC/DC
Reference Design
Clamp-on Ammeter Power Quality Meter
I
+
V
–
AC Mains
DS70320B-page 72
L
L
G
G
N
N
SMPS AC/DC
Reference Design
© 2008 Microchip Technology Inc.
System Operation
A power meter is capable of measuring the input Power Factor of the SMPS AC/DC
Reference Design and also the Total Harmonic Distortion (THD) on the current drawn
by the system.
Using a programmable AC source will enable the user to evaluate the system
performance over the entire range of input voltage (85V-265V, 45Hz-65Hz).
4.2.2.2
OUTPUT PERFORMANCE TESTING
The SMPS AC/DC Reference Design can be loaded using DC electronic loads. A
number of output parameters can be tested by connecting oscilloscope probes to the
output terminals. Some of the parameters that can be tested are:
•
•
•
•
•
Output Ripple Voltage
Load Regulation
Transient Response
Step-Load Response
Fault Shutdown Delay
© 2008 Microchip Technology Inc.
DS70320B-page 73
SMPS AC/DC Reference Design User’s Guide
NOTES:
DS70320B-page 74
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Appendix A. Board Layouts and Schematics
A.1
INTRODUCTION
This appendix contains the schematics and layouts for the SMPS AC/DC Reference
Design (Power board and Signal board).
SMPS AC/DC REFERENCE DESIGN LAYOUT
SMPS AC/DC REFERENCE DESIGN LAYOUT (POWER BOARD)
GND
+3.3V
GND
+12V
SMPS Power Rev C.1
FAN
+5V
FIGURE A-1:
GND
A.2
© 2008 Microchip Technology Inc.
DS70320B-page 75
SMPS AC/DC Reference Design User’s Guide
FIGURE A-2:
DS70320B-page 76
SMPS AC/DC REFERENCE DESIGN LAYOUT (SIGNAL BOARD)
© 2008 Microchip Technology Inc.
Board Layouts and Schematics
SMPS AC/DC REFERENCE DESIGN SCHEMATICS
EMI FILTER SCHEMATIC
© 2008 Microchip Technology Inc.
NEUTRAL
1
EARTH
2
3
LIVE
EARTH
EARTH
EMC_LIVE
FIGURE A-3:
EMC_NEUTRAL
A.3
DS70320B-page 77
DS70320B-page 78
LIVE_GND
+HV_BUS
EMC_NEUTRAL
GBU8
EMC_LIVE
BR1
HV_LINK_SENSE
D15
PFC_SHUNT_NEG
BAT54S_SOT23
PFC_FIRE
U5
EMC_NEUTRAL
EMC_LIVE
-HV_BUS
LIVE_DRIVE_SUPPLY
1N5408
BAS16
D10
BAS16
D12
DNP
|VAC|_SENSE
DO-220 Formed
STTH5R06D
D1
-HV_BUS
EARTH
+HV_BUS
FIGURE A-4:
NTC1
D23
SMPS AC/DC Reference Design User’s Guide
PFC CIRCUIT SCHEMATIC
© 2008 Microchip Technology Inc.
C31
D=21mm, P=7.5mm
© 2008 Microchip Technology Inc.
ZVT_LEFT_LOW_FIRE
ZVT_LEFT_HIGH_FIRE
DRIVE_B
DRIVE_A
D47
D45
8
1
-HV_BUS
6
5
4
3
LIVE_GND
BAS16
D4
BAS16
BRIDGE_A
C6
U4
DRIVE_B
DRIVE_A
LIVE_DRIVE_SUPPLY
Connect Resistor and Transformer Directly to Source
and then connect to -HV_BUS
C8
ZVT_RIGHT_LOW_FIRE
ZVT_RIGHT_HIGH_FIRE
DRIVE_D
DRIVE_C
8
1
D49
D48
6
5
4
3
LIVE_GND
BAS16
D6
BAS16
D3
C5
BRIDGE_B
U3
LIVE_DRIVE_SUPPLY
DRIVE_D
DRIVE_C
FIGURE A-5:
C7
D2
+HV_BUS
Board Layouts and Schematics
FULL-BRIDGE ZVT SCHEMATIC
DS70320B-page 79
LIVE_GND
ZVT_CT_|BURDEN|_POS
C25
SYNCH_RECT_#2_FIRE
SYNCH_RECT_#1_FIRE
BRIDGE_B
LIVE_GND
12V_BUS_ERROR_F/B
LIVE_GND
BAT54S_SOT23
D11
Q23
S
P
D13
S
P
BAT54S_SOT23
6
10
L3
4
2
7
4
10mm dia, 5mm pitch
SFH617-2
4
BRIDGE_A
C43
3
DS70320B-page 80
2N2 Y2
NEG_OUT
R14
+DIG
15R 3W AX P=20mm
12V_F/B_OPTO_CATHODE
12V_F/B_OPTO_ANODE
DNP
SR#2_FIRE
SR#1_FIRE
U2
L2
BAS16
D8
DRIVE_SUPPLY
9uH
L1
9uH
BAS16
D7
SR#2_FIRE
SR#1_FIRE
12V_F/B_OPTO_CATHODE
12V_F/B_OPTO_ANODE
12V_OUT
D9
1000pF
NEG_OUT
C20 4.7nF
C24
NEG_OUT
12V_OUT
100pF
C21
FIGURE A-6:
R13
+HV_BUS
SMPS AC/DC Reference Design User’s Guide
SYNCHRONOUS RECTIFIER AND ZVT CURRENT SENSE SCHEMATIC
© 2008 Microchip Technology Inc.
-HV_BUS
+HV_BUS
R62
3.9M .25W MetalGlazed
TNY277G
100nF
1K 0.6W P6KE150A
D22
R48
1N4007
© 2008 Microchip Technology Inc.
U10
10nF 1KV
100R 0.6W
5mm Pitch, 10mm Dia Disc
C34
BL01RN1A1D2B
L10
L11
BL01RN1A1D2B
LIVE_DRIVE_SUPPLY
LIVE_GND
UF4002
D32
LIVE_DIG_SUPPLY
1N5819
D35
Q22
3
4
SFH617-2
D37
L9
D25
UF4002
DRIVE_SUPPLY
BL01RN1A1D2B
BZx84C16
1N5819
D29
DIG_SUPPLY
L7
BL01RN1A1D2B
DRIVE_SUPPLY
FIGURE A-7:
D30
LIVE_DRIVE_SUPPLY
Board Layouts and Schematics
AUXILIARY POWER SUPPLY SCHEMATIC
DS70320B-page 81
R58
D19
NEG_OUT
12V_OUT
NEG_OUT
12V_OUT
NEG_OUT
7
8
1
CT_NEG
3.3V_OUT
3.3V_BUCK#3_SHUNT_POS
NEG_OUT
3.3V_BUCK#1_FW_GATE
3.3V_BUCK#1_CONTROL_GATE
3.3V_OUT
3.3V_BUCK#1_SHUNT_POS
NEG_OUT
3.3V_BUCK#3_FW_GATE
3.3V_BUCK#3_CONTROL_GATE
NEG_OUT
5V_BUCK_FW_GATE
5V_BUCK_CONTROL_GATE
CT_POS
L8
Coilcraft SER1360 1uH
L5
D31
D14
3
L14
D41
Coilcraft SER1360 1uH
15MQ040NPBF
15MQ040NPBF
15MQ040NPBF
DS70320B-page 82
NEG_OUT
NEG_OUT
CT_POS
CT_NEG
3.3V_OUT
3.3V_OUT_J3P5_IN
3.3V_OUT_J3P9_IN
NEG_OUT
3.3V_BUCK#1_SHUNT_POS_HDR
3.3V_OUT
NEG_OUT
D46
L6
NEG_OUT
NEG_OUT
3.3V_OUT_J3P7_IN
5V_BUCK_SHUNT_NEG_SENSE
5V_BUCK_SHUNT_POS_SENSE
3.3V_BUCK#2_SHUNT_POS_HDR
3.3V_OUT
NEG_OUT
47pF
Coilcraft SER1360 1uH
BAT54S_SOT23
3.3V_BUCK#2_SHUNT_POS
3.3V_OUT
3.3V_BUCK#2_SHUNT_POS
NEG_OUT
3.3V_BUCK#2_FW_GATE
3.3V_BUCK#2_CONTROL_GATE
3.3V_BUCK#3_SHUNT_POS_HDR
NEG_OUT
NEG_OUT
3.3V_BUCK#1_SHUNT_POS
3.3V_BUCK#3_SHUNT_POS
NEG_OUT
12V_OUT
NEG_OUT
5V_OUT
D20
FIGURE A-8:
15MQ040NPBF
12V_OUT
SMPS AC/DC Reference Design User’s Guide
BUCK CONVERTER STAGES SCHEMATIC
© 2008 Microchip Technology Inc.
DRIVE_SUPPLY
3.3V_BUCK#1_FW_FIRE
3.3V_BUCK#1_CONTROL_FIRE
PULLUP_SUPPLY
3.3V_BUCK#3_FW_FIRE
D17
© 2008 Microchip Technology Inc.
BZX84C5V1
D18
D36
PULLUP_SUPPLY
BAT54S_SOT23
BAT54S_SOT23
D33
DRIVE_SUPPLY
BAT54S_SOT23
BAT54S_SOT23
D16
DRIVE_SUPPLY
MCP1404-E/SN_SO8N
MCP1404-E/SN_SO8N
5V_BUCK_FW_FIRE
3.3V_BUCK#3_FW_GATE
NEG_OUT
PULLUP_SUPPLY
5V_BUCK_CONTROL_FIRE
3.3V_BUCK#3_CONTROL_GATE
NEG_OUT
3.3V_BUCK#1_FW_GATE
3.3V_BUCK#1_CONTROL_GATE
3.3V_BUCK#2_FW_FIRE
3.3V_BUCK#2_CONTROL_FIRE
PULLUP_SUPPLY
D26
D44
BAT54S_SOT23
BAT54S_SOT23
D43
DRIVE_SUPPLY
BAT54S_SOT23
BAT54S_SOT23
D24
DRIVE_SUPPLY
MCP1404-E/SN_SO8N
NEG_OUT
NEG_OUT
5V_BUCK_FW_GATE
5V_BUCK_CONTROL_GATE
MCP1404-E/SN_SO8N
3.3V_BUCK#2_FW_GATE
3.3V_BUCK#2_CONTROL_GATE
FIGURE A-9:
3.3V_BUCK#3_CONTROL_FIRE
PULLUP_SUPPLY
Board Layouts and Schematics
BUCK CONVERTER GATE DRIVE SCHEMATIC
DS70320B-page 83
PKSA
5V_OUT
PKSB
5V_REMOTE_POS
5V_REMOTE_NEG
U11
0.1uF
MCP9700_SC-70
+DIG
D39
3.3V_REMOTE_POS
3.3V_REMOTE_NEG
Remote Analog Inputs
TEMP_SENSE
NEG_OUT
D40
DS70320B-page 84
BAT54S_SOT23
+DIG
SDA
SCL
PKSB
+DIG
PKSA
3.3V_BUCK#3_CONTROL_FIRE
3.3V_BUCK#3_FW_FIRE
3.3V_BUCK#2_CONTROL_FIRE
3.3V_BUCK#2_FW_FIRE
3.3V_BUCK#1_CONTROL_FIRE
3.3V_BUCK#1_FW_FIRE
5V_BUCK_CONTROL_FIRE
5V_BUCK_FW_FIRE
3.3V_OUT
3.3V_REMOTE_POS
3.3V_REMOTE_NEG
3.3V_BUCK#1_SHUNT_POS_HDR
3.3V_OUT_J3P5_IN
3.3V_BUCK#2_SHUNT_POS_HDR
3.3V_OUT_J3P7_IN
3.3V_BUCK#3_SHUNT_POS_HDR
3.3V_OUT_J3P9_IN
TEMP_SENSE
5V_OUT
5V_REMOTE_POS
5V_REMOTE_NEG
5V_BUCK_SHUNT_POS_SENSE
5V_BUCK_SHUNT_NEG_SENSE
12V_OUT
DRIVE_SUPPLY
DIG_SUPPLY
NEG_OUT
12V_OUT
J2
NEG_OUT
5V_OUT
NEG_OUT
NEG_OUT
3.3V_OUT
3.3V_OUT
220R
LIVE_SDA
LIVE_SCL
NEG_OUT
12V_OUT
100nF
U8
BZX84C5V1
SYNCH_RECT_#1_FIRE
LIVE_GND
ZVT_LEFT_HIGH_FIRE
ZVT_LEFT_LOW_FIRE
ZVT_RIGHT_HIGH_FIRE
ZVT_RIGHT_LOW_FIRE
LIVE_GND
0.1uF
PFC_FIRE
SYNCH_RECT_#2_FIRE
MCP9700_SC-70
D34
LIVE_GND
LIVE_+DIG
TC74_TO-220_5PUP
U1
LIVE_+DIG
LIVE_DIG_SUPPLY
1
LIVE_GND
|VAC|_SENSE
HV_LINK_SENSE
LIVE_+DIG
LIVE_GND
LIVE_DIG_SUPPLY
LIVE_SCL
LIVE_SDA
LIVE_TEMP_SENSE
-HV_BUS
PFC_SHUNT_NEG
12V_BUS_ERROR_F/B
ZVT_CT_|BURDEN|_POS
TP
LIVE_DRIVE_SUPPLY
FIGURE A-10:
BAT54S_SOT23
12V_OUT
SMPS AC/DC Reference Design User’s Guide
CONTROL BOARD I/F AND TEMPERATURE SENSING/MISC. SCHEMATIC
© 2008 Microchip Technology Inc.
Board Layouts and Schematics
PRIMARY SIDE CONTROLLER SCHEMATIC
1R
R77
FIGURE A-11:
dsPIC33FJ16GS504
DNP
R46
IC_24LC128_SO8
LIVE_GND
1uF
R96
4K7
1R or Ferrite beads
4K7 R54
LIVE_+DIG
4K7 R56
470R
R35
470R
C74 0.1uF
R182
C28 1uF
© 2008 Microchip Technology Inc.
DS70320B-page 85
SMPS AC/DC Reference Design User’s Guide
FIGURE A-12:
PRIMARY ↔ SECONDARY COMMUNICATION SCHEMATIC
HCPL-2611#300
R24
270R
C8
100pF
HCPL-2611#300
R75
270R
R74
1K
C21
100pF
R120
HCPL-2611#300
270R
R109
1K
DS70320B-page 86
© 2008 Microchip Technology Inc.
Board Layouts and Schematics
C16
1nF
R47
2.7K
R78
1.3K
R147
C31
1nF
4K7
R148
100R
1nF
PRIMARY SIDE FEEDBACK CIRCUITS SCHEMATIC
C18
FIGURE A-13:
© 2008 Microchip Technology Inc.
DS70320B-page 87
© 2008 Microchip Technology Inc.
3.3K R13
R89 100K
R86
10K
R91
47K
R93
47K
R92
47K
R94
47K
R2 DNP
R7
100K
10K
10K
R95
10K
R85
10K
R84
10K
R81
R90
10K
R25
10K
R88
R32
10K
220pF C5
10K
220pF C6
R33
220pF C37
R3 10K
10K
220pF C36
R36
220pF C39
R106
R80
R83
R31
1M
1M
1M
1M
1M
R30
Under-Voltage AC
Over-Voltage AC
R87 4K7
DS70320B-page 88
R82 4K7
R29 4K7
R76
4K7
R28 4K7
R17 470R
R14
R18 470R
470R
R15 470R
R4 4K7
R19 470R
R16 470R
R1214K7
R122
1M
4K7
4K7
R190
R130
100pF C44
R188
R97
0R
4.7K
R129100K
R117100K
FIGURE A-14:
PRIMARY SIDE HARDWARE FAULT CIRCUITRY SCHEMATIC
SMPS AC/DC Reference Design User’s Guide
Board Layouts and Schematics
1R
IC_24LC128_SO8
dsPIC33FJ16GS504
R98
4K7
1uF
1uF
R53
4K7
GND
R52
+DIG
R51
4K7
R68
SECONDARY SIDE CONTROLLER SCHEMATIC
4K7
FIGURE A-15:
470R
R12
470R
0.1uF C23
R181
C711uF
© 2008 Microchip Technology Inc.
DS70320B-page 89
DS70320B-page 90
OPA4354AID
OPA4354AID
BAT54S_SOT23
D35
R140 3.3K
R69
1K
FIGURE A-16:
OPA4354AID
OPA4354AID
Remove R100,R105,R65, R60
Replace 0R at the following locations
R101,R102, R103, R104, R61, R62, R63, R64
When using the Maxim Part:
SMPS AC/DC Reference Design User’s Guide
SECONDARY SIDE FEEDBACK SCHEMATIC
© 2008 Microchip Technology Inc.
R45 6K8
R38 100K
R57 47K
R66 47K
R39 47K
R27 47K
R23 47K
47K
R6
R55 10K
R59 10K
R26
10K
10K
10K
R44
R37
10K
R34
C17
220pF
R67 10K
R49 10K
220pFC19
220pFC2
220pFC13
R21
R5
R58
R48
1M
1M
1M
1M
R1
4K7
R79
4K7
4K7
R150
1M
R123
R22 470R
R116
R10 470R
C53
220pF
10K R136
10K
470R
© 2008 Microchip Technology Inc.
R9
R11 470R
R135
47K
R114
4K7
R118
1M
0R
C58
100pF
R134 4K7
R126 4K7
R142
100K
R143
100K
FIGURE A-17:
R191
Over_Voltage DC Bus
Board Layouts and Schematics
SECONDARY SIDE HARDWARE FAULT CIRCUITS SCHEMATIC
DS70320B-page 91
R189 4.7K
R145 47K
R8 470R
R155 4k7
SMPS AC/DC Reference Design User’s Guide
NOTES:
DS70320B-page 92
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Appendix B. Test Results
This appendix provides information on the test procedures and results for the SMPS
AC/DC Reference Design.
The following equipment was used to test the SMPS AC/DC Reference Design:
•
•
•
•
•
•
B.1
Programmable DC Load (or resistive load)
Two- or Four-Channel Oscilloscope (100 MHz or higher)
Current probe and Differential probe
Programmable AC Source, or Variac, or AC Power Cord
Power Meter
True RMS Multimeter
SOFT-START AND OVERSHOOT
The SMPS AC/DC Reference Design has soft-start routines implemented for the PFC
Boost converter, ZVT converter, Single-Phase and Multi-Phase Buck converters. The
soft-start routines eliminate in-rush current, eliminates overshoot and provides control
of the output voltages during start-up. The soft-start scheme is sequenced as follows:
•
•
•
•
The PFC Boost converter ramps to 420V,
the ZVT converter ramps to 12V,
the Single-Phase Buck converter ramps to 5V,
and then the Multi-Phase converter ramps to 3.3V.
B.1.1
Test Procedure
1. Ensure that the system is off and that all probes are disconnected.
2. Connect the oscilloscope probes across the output terminals (J2, J4, and J17).
3. Set up the oscilloscope for normal trigger mode and on the rising edge of the
scope connected to (J2) with a time scale equal to 100 ms or greater. Move the
trigger start point to 200 ms.
4. Power on the unit and observe the soft-start.
5. Turn off the system and connect a programmable DC load to any single output
or to the 5V and 3.3V outputs simultaneously.
6. Power on the unit and observe the soft-start.
7. Turn off the system and disconnect all probes.
Figure B-1 demonstrates the soft-start sequence for the ZVT converter and the
Single-Phase and Multi-Phase converters.
Figure B-2 demonstrates the soft-start sequence with the Single-Phase and
Multi-Phase outputs loaded (5V @ 23A, 3.3V @ 35A).
© 2008 Microchip Technology Inc.
DS70320B-page 93
SMPS AC/DC Reference Design User’s Guide
FIGURE B-1:
Note:
The PFC soft-start (not shown in Figure B-1 and Figure B-2) should not be
observed simultaneously with the secondary side outputs.
FIGURE B-2:
DS70320B-page 94
SOFT-START WITHOUT LOAD
SOFT-START WITH LOAD
© 2008 Microchip Technology Inc.
Test Results
B.2
DYNAMIC LOAD RESPONSE
Dynamic load response is measuring the under/overshoot voltage and settling time of
the output voltage when performing a load step. The SMPS AC/DC Reference Design
has the following load step parameters when the outputs are loaded individually:
• 12V full load step of 30A
• 5V full load step of 23A
• 3.3V full load step of 69A
When the Single-Phase and Multi-Phase converters are loaded simultaneously, the
following are the max load steps:
• 5V full load step of 23A
• 3.3V full load step of 56A
The load response of each unit is tested with the following load steps with a maximum
slew rate of 1A/us:
• 0-15A and 15-0A (for the 12V output)
• 0-35A and 35-0A (for the 3.3V output)
• 0-12A and 12-0A (for the 5V output)
B.2.1
Test Procedure
1.
2.
3.
4.
Ensure that the system is off and that all probes are disconnected.
Connect a programmable DC load to any one of the three outputs.
Connect the oscilloscope probe across the output terminals with the DC load.
Connect the current probe to one of the load cables making sure of the direction
of current flow.
5. Set up the oscilloscope for a single capture and trigger on the current probe on
either edge and set the oscilloscope channel for AC coupling.
6. Perform the load steps and measure the settling time and under/overshoot
voltage.
7. Turn off the system and disconnect all probes.
Figure B-3 through Figure B-8 show the dynamic load response and settling time with
50% load steps.
© 2008 Microchip Technology Inc.
DS70320B-page 95
SMPS AC/DC Reference Design User’s Guide
DS70320B-page 96
FIGURE B-3:
12V LOAD RESPONSE 0-15A
FIGURE B-4:
12V LOAD RESPONSE 15-0A
© 2008 Microchip Technology Inc.
Test Results
FIGURE B-5:
3.3V LOAD RESPONSE 0-35A
FIGURE B-6:
3.3V LOAD RESPONSE 35-0A
© 2008 Microchip Technology Inc.
DS70320B-page 97
SMPS AC/DC Reference Design User’s Guide
DS70320B-page 98
FIGURE B-7:
5V LOAD RESPONSE 0-12A
FIGURE B-8:
5V LOAD RESPONSE 12-0A
© 2008 Microchip Technology Inc.
Test Results
B.3
OUTPUT VOLTAGE RIPPLE
Output voltage ripple is measured across the output capacitors with the shortest probe
ground possible. For production tests, the output voltage ripple is measured at the
terminal blocks. Refer to Figure B-9 for the oscilloscope probe connection location
used to measure the output voltage ripple.
FIGURE B-9:
EXAMPLE OF OSCILLOSCOPE PROBE CONNECTION
Probe connection point
B.3.1
Test Procedure
1. Ensure that the system is OFF and all probes are disconnected.
2. Connect the oscilloscope probe across the output capacitor to be measured as
shown in Figure B-9.
Note:
Oscilloscope probes should have the shortest ground wire possible to
eliminate noise when measuring the ripple voltage.
3. Set up the oscilloscope with a time scale of 2 µs and set the oscilloscope channel
to AC coupling and 50 mV per division.
4. Measure the peak-to-peak voltage.
5. To test the output voltage ripple with a load, connect a programmable DC load to
the output terminal and re-measure the voltage ripple.
6. Turn off the system and disconnect all probes.
Figure B-10 through Figure B-12 show the output voltage ripple of each stage without
load.
© 2008 Microchip Technology Inc.
DS70320B-page 99
SMPS AC/DC Reference Design User’s Guide
DS70320B-page 100
FIGURE B-10:
3.3V OUTPUT VOLTAGE RIPPLE
FIGURE B-11:
5V OUTPUT VOLTAGE RIPPLE
© 2008 Microchip Technology Inc.
Test Results
FIGURE B-12:
B.4
12V OUTPUT VOLTAGE RIPPLE
INPUT CURRENT
The SMPS AC/DC Reference Design implements Power Factor Correction (PFC)
where the current is in phase with the input voltage.
B.4.1
Test Procedure:
1. Ensure that the system is off and that all probes are disconnected.
2. Connect a differential probe across the input terminal (J16). Connect across the
“neutral” and “live” terminals.
WARNING
Do not connect a standard probe across the AC terminal. A differential probe must be
used. Failure to heed this warning may result in bodily harm and damage to the
oscilloscope and/or SMPS AC/DC Reference Design.
3. Connect the current probe around the live or neutral power cable making sure of
the direction of current flow.
4. Set the current probe for 100 mV per Amp and set the oscilloscope channel to a
1:1 ratio.
5. Connect load cables to the 12V output (J1).
6. Verify your connections and turn on the system. You should be able to observe
sinusoidal input voltage and current. If not, connect the current probe to the other
input.
7. Apply a load to the output and verify that the current and the voltage are still
sinusoidal and in phase.
8. Turn off the system and disconnect all probes.
© 2008 Microchip Technology Inc.
DS70320B-page 101
SMPS AC/DC Reference Design User’s Guide
Figure B-13 and Figure B-14 demonstrate the input current and the input voltage at full
load operations at 110 VAC and 220 VAC.
DS70320B-page 102
FIGURE B-13:
INPUT CURRENT AND INPUT VOLTAGE @110 VAC WITH
FULL LOAD
FIGURE B-14:
INPUT CURRENT AND INPUT VOLTAGE @ 220 VAC WITH
FULL LOAD
© 2008 Microchip Technology Inc.
Test Results
B.5
EFFICIENCY
When loading the 12V output (VOUT1), the efficiency is approximately 82 percent. If
loading the 3.3V and 5V outputs (VOUT2 and VOUT3) simultaneously, the efficiency is
approximately 74 percent at 300W output. Efficiency is measured by dividing the output
power by the input power. The power is calculated by multiplying the current (load) by
the output voltage. For example, full load on 12V output will yield 360 Watts (12V * 30A
= 360W).
If using an AC power cable, connect a true RMS multimeter in series with the power
cable to measure the RMS input current, and connect another multimeter in parallel
with the power cable to measure RMS input voltage. To calculate input power, multiply
the input RMS current by the input RMS voltage. To calculate the percent efficiency
divide the output power by the input power and multiply the result by 100.
EXAMPLE B-1:
IOUT1 = 0A @ 12V
IOUT2 = 56A @ 3.3V
IOUT3 = 23A @ 5V
VIN = 110V
IIN = 3.7A
POUT
= (VOUT1 * IOUT1) + (VOUT2 * IOUT2) + (VOUT3 * IOUT3)
= (56A * 3.3V) + (23A * 5V)
= 299.8W
PIN
= VIN * IIN
= 407W
Efficiency (%) = POUT / PIN
= (299.8 / 407) * 100
= 73.7%
B.6
INPUT CURRENT TOTAL HARMONIC DISTORTION (ITHD)
Using a Voltech PM100 or similar power meter, measure the ITHD. ITHD is tested at
the following conditions:
• VIN = 110V @ 60 Hz @ Full Load
• VIN = 220V @ 50 Hz @ Full Load
The current ITHD measurements of the SMPS AC/DC Reference Design at 110 VAC is
approximately 4.8% and the ITHD at 220 VAC is approximately 6%.
B.7
POWER FACTOR
Using a Voltech PM100 or similar power meter, measure the power factor. The power
factor of each reference design is tested at the following:
•
•
•
•
VIN = 110V @ 60 Hz @ Full Load
VIN = 220V @ 50 Hz @ Full Load
VIN = 110V @ 60 Hz @ No Load
VIN = 220V @ 50 Hz @ No Load
The current power factor of the SMPS AC/DC Reference Design at 110V at full load is
.998 and the power factor at 220V at full load is .99.
© 2008 Microchip Technology Inc.
DS70320B-page 103
SMPS AC/DC Reference Design User’s Guide
B.8
TEST RESULTS TABLE
Each SMPS AC/DC Reference Design is extensively tested from no/full load starts to
maximum current steps across the universal input voltage range. In addition, each unit
passes a rigorous 12-hour burn-in test at full load (300W). The following table shows
the complete list of tests performed on the SMPS AC/DC Reference Design before the
unit is shipped.
TABLE B-1:
TEST RESULTS TABLE
Tests
Min
Max
Units
Result
Remarks
Input Current
Max input current at 110 VAC
—
5
A
3.75
Loading VOUT2 @ 56A
and VOUT3 @ 23A
Max input current at 220 VAC
—
1.8
A
1.73
Loading VOUT2 @ 56A
and VOUT3 @ 23A
Maximum in-rush current at VIN @ 220 VAC @ No
Load
—
35
A
27
Output Overshoot at Start-up
VIN = 110 VAC
VOUT1 @ 0A
—
12.48
V
12
VOUT2 @ 0A
—
3.432
V
3.3
VOUT3 @ 0A
—
5.2
V
5
VOUT1 @ 0A
—
12.48
V
12
VOUT2
@ 0A
—
3.432
V
3.3
VOUT3 @ 0A
—
5.2
V
5
0.974
VIN = 220 VAC
Power Factor
VIN = 110 VAC @ No Load @ 60Hz
0.9
—
—
VIN = 220 VAC @ No Load @ 50Hz
0.8
—
—
0.855
= 110 VAC @ Full Load @ 60Hz
0.98
—
—
0.997
VIN = 220 VAC @ Full Load @ 50Hz
0.97
—
—
0.989
VIN
Measured by power
meter
Input current THD @ VTHD @ 2%
VIN = 110 VAC @ Full Load @ 60Hz
—
5
%
4.9
VIN = 220 VAC @ Full Load @ 50Hz
—
7
%
6
Input Power
VOUT1 @ 30 A, VIN @ 110 VAC
—
—
watt
445
VOUT1 @ 0A, VOUT2 @ 56A, VOUT3 @ 23A,
VIN @ 110 VAC
—
—
watt
415
VOUT1 @ 30 A, VIN @ 220 VAC
—
—
watt
431
VOUT1 @ 0A, VOUT2 @ 56A, VOUT3 @ 23A,
VIN @ 220 VAC
—
—
watt
400
DS70320B-page 104
© 2008 Microchip Technology Inc.
Test Results
TABLE B-1:
TEST RESULTS TABLE (CONTINUED)
Tests
Min
Max
Units
Result
Remarks
Set Point Voltage ( @ 110 VAC )
Set point voltage of 12V (VOUT1) @ 15A
11.94
12.06
Volt
11.97
Set point voltage of 3.3V (VOUT2) @ 35A
3.2835
3.3165
Volt
3.31
4.98
5.02
Volt
4.98
Set point voltage of 5V (VOUT3) @ 12A
Measure output
voltage with
multimeter
Line & Load Regulation
VIN = 110 VAC
VOUT1 @ 0A
11.94
12.06
Volt
11.98
VOUT1 @ 30A
11.88
12.12
Volt
11.97
VOUT2 @ 0A
3.2835
3.3165
Volt
3.32
VOUT2 @ 69A
3.2835
3.3165
Volt
3.3
VOUT3 @ 0A
4.98
5.02
Volt
5.01
VOUT3 @ 23A
4.98
5.02
Volt
4.98
11.94
12.06
Volt
11.98
VIN = 220 VAC
VOUT1 @ 0A
VOUT1 @ 30A
11.88
12.12
Volt
11.96
VOUT2 @ 0A
3.2835
3.3165
Volt
3.32
VOUT2 @ 69A
3.2835
3.3165
Volt
3.3
VOUT3 @ 0A
4.98
5.02
Volt
5.01
VOUT3 @ 23A
4.98
5.02
Volt
4.98
Output Voltage Ripple
VIN = 110 VAC
VOUT1 @ 0A
—
—
mV
130
Measured at output
terminals(1)
VOUT1 @ 30A
—
—
mV
160
Measured at output
terminals(1)
VOUT2 @ 0A
—
—
mV
90
Measured at output
terminals(1)
VOUT2 @ 69A
—
—
mV
90
Measured at output
terminals(1)
VOUT3 @ 0A
—
—
mV
160
Measured at output
terminals(1)
VOUT3 @ 23A
—
—
mV
130
Measured at output
terminals(1)
Note 1:
See Section B.3 “Output Voltage Ripple” for details on how to measure output voltage ripple.
Dynamic Load Response Slew Rate = 1A/µs ( Tests @ 110 VAC )
VOUT2 = 0A, VOUT3 = 0A, VOUT1 = 0A-15A
—
—
mV
470
VOUT2 = 0A, VOUT3 = 0A, VOUT1 = 15A-0A
—
—
mV
380
VOUT1 = 0A, VOUT3 = 0A, VOUT2 = 0A-35A
—
—
mV
100
VOUT1 = 0A, VOUT3 = 0A, VOUT2 = 35A-0A
—
—
mV
100
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 0A-12A
—
—
mV
190
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 12A-0A
—
—
mV
210
© 2008 Microchip Technology Inc.
DS70320B-page 105
SMPS AC/DC Reference Design User’s Guide
TABLE B-1:
TEST RESULTS TABLE (CONTINUED)
Tests
Min
Max
Units
Result
—
us
656
Remarks
Settle Time ( @ 110 VAC )
VOUT2 = 0A, VOUT3 = 0A, VOUT1 = 0A-15A
—
VOUT2 = 0A, VOUT3 = 0A, VOUT1 = 15A-0A
—
—
us
1976
VOUT1 = 0A, VOUT3 = 0A, VOUT2 = 0A-35A
—
—
us
120
VOUT1 = 0A, VOUT3 = 0A, VOUT2 = 35A-0A
—
—
us
88
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 0A-12A
—
—
us
528
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 12A-0A
—
—
us
225
Efficiency @ 110 VAC
Efficiency when 12 @30A
—
—
%
81
Efficiency when 3.3V@56A and 5V@23A
—
—
%
72
Set Point Limits
Input Under Voltage Lock-out (no load)
75
—
VAC
76.3
Input Over Voltage Lock-out (no load)
—
270
VAC
269
5V over-current limit
—
27
A
25.4
3.3V over-current limit
—
75
A
74
Full Load Start (Pass or Fail)
VOUT1= 0A, VOUT2 = 56A, VOUT3 = 23A @ 85 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 56A, VOUT3 = 23A @ 110 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 56A, VOUT3 = 23A @ 220 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 56A, VOUT3 = 23A @ 265 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 69A, VOUT3 = 0A @ 85 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 69A, VOUT3 = 0A @ 110 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 69A, VOUT3 = 0A @ 220 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 69A, VOUT3 = 0A @ 265 VAC
—
—
—
Pass
VOUT1= 30A, VOUT2 = 0A, VOUT3 = 0A @ 85 VAC
—
—
—
Pass
VOUT1= 30A, VOUT2 = 0A, VOUT3 = 0A @ 110 VAC
—
—
—
Pass
VOUT1= 30A, VOUT2 = 0A, VOUT3 = 0A @ 220 VAC
—
—
—
Pass
VOUT1= 30A, VOUT2 = 0A, VOUT3 = 0A @ 265 VAC
—
—
—
Pass
—
—
Pass
No Load Start
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 0A @ 85 VAC
—
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 0A @ 110 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 0A @ 220 VAC
—
—
—
Pass
VOUT1= 0A, VOUT2 = 0A, VOUT3 = 0A @ 265 VAC
—
—
—
Pass
DS70320B-page 106
© 2008 Microchip Technology Inc.
Test Results
TABLE B-1:
TEST RESULTS TABLE (CONTINUED)
Tests
Min
Max
Units
Result
Remarks
Pass or Fail (If Fail,
what is max current
step)
Max Current Steps
VOUT1 (0-30A, 30-0A) @ 85 VAC
—
—
—
Pass
VOUT1 (0-30A, 30-0A) @ 110 VAC
—
—
—
Pass
VOUT1 (0-30A, 30-0A) @ 220 VAC
—
—
—
Pass
VOUT1 (0-30A, 30-0A) @ 265 VAC
—
—
—
Pass
Not measured with
time (12-hour burn-in
is done @ 56A)
VOUT2 (0-69A, 69-0A) @ 85 VAC
—
—
—
Pass
VOUT2 (0-69A, 69-0A) @ 110 VAC
—
—
—
Pass
VOUT2 (0-69A, 69-0A) @ 220 VAC
—
—
—
Pass
VOUT2 (0-69A, 69-0A) @ 265 VAC
—
—
—
Pass
VOUT3 (0-23A, 23-0A) @ 85 VAC
—
—
—
Pass
VOUT3 (0-23A, 23-0A) @ 110 VAC
—
—
—
Pass
VOUT3 (0-23A, 23-0A) @ 220 VAC
—
—
—
Pass
VOUT3 (0-23A, 23-0A) @ 265 VAC
—
—
—
Pass
Full Load steps with
VOUT2 while VOUT3
loaded
VOUT2 (0-56A, 56-0A) VOUT3 @ 23A @ 85 VAC
—
—
—
Pass
VOUT2 (0-56A, 56-0A) VOUT3 @ 23A @ 110 VAC
VOUT2 (0-56A, 56-0A) VOUT3 @ 23A @ 220 VAC
—
—
—
Pass
—
—
—
Pass
VOUT2 (0-56A, 56-0A) VOUT3 @ 23A @ 265 VAC
—
—
—
Pass
Full Load steps with
VOUT3 while VOUT2
loaded
VOUT2 @ 56A, VOUT3 (0-23A, 23-0A) @ 85 VAC
—
VOUT2 @ 56A, VOUT3 (0-23A, 23-0A) @ 110 VAC
VOUT2 @ 56A, VOUT3 (0-23A, 23-0A) @ 220 VAC
VOUT2 @ 56A, VOUT3 (0-23A, 23-0A) @ 265 VAC
—
—
Pass
—
—
—
Pass
—
—
—
Pass
—
—
—
Pass
Burn-in Test for 12Hrs continuous @ 300W, VOUT2 @ 56A, VOUT3 @ 23A
VOUT1
11.94
12.06
Volt
Pass
Measured 11.95
VOUT2
3.2835
3.3165
Volt
Pass
Measured 3.3
VOUT3
4.98
5.02
Volt
Pass
Measured 4.98
Pass
Not characterized on
all units
Extended Temperature Tests @ 225W, 50° C
VOUT2 @ 42A, VOUT3 @ 17A
© 2008 Microchip Technology Inc.
DS70320B-page 107
SMPS AC/DC Reference Design User’s Guide
NOTES:
DS70320B-page 108
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Appendix C. References
This section provides the list of references used throughout this document.
1
Herfurth, M., “Active Harmonic Filtering for Line Rectifiers of Higher Output Power”,
Siemens Components XXI (1986), No. 1, pp. 9 - 13.
2
Zhou, C.and Jovanovic, M.M., “Design Trade-offs in Continuous Current-Mode Controlled Boost Power-Factor Correction Circuits”, Proceedings of HFPC '92, May 1992,
pp. 209 - 219.
3
Chen, W, Lee, F. C., Jovanovic, M. M., and Sabate, J. A., “A Comparative Study of a
Class of Full Bridge Zero-Voltage-Switched PWM Converters”, Proceedings of APEC
'95, pp. 893 - 899.
4
Frank, W., Dahlquist, F., Kapels, H., Schmitt, M., and Deboy, G., “Compensation
MOSFETs with Fast Body Diode - Benefits in Performance and Reliability in ZVS
Applications”. Proceedings of IPECSA, the International Power Electronics
Component Systems Applications Conference, San Francisco, CA, USA, March 29 April 1, 2004.
5Snelling,
E.C., “Soft Ferrites. Properties and Applications”, 2nd Edition, Butterworths,
London, UK, 1988.
6
Jongsma, J., “High-Frequency Ferrite Power Transformer and Choke Design - Part 3
Transformer Winding Design”, N. V. Philips, September 1982.
7
Dowell, P. L., “Effects of Eddy Currents in Transformer Windings”, Proceedings of the
IEE, Vol. 113, No. 8, August 1966, pp. 1387 - 1394.
8McLyman, C. W. T., “Transformer and Inductor Design Handbook”, 3rd edition, Marcel
Dekker Inc., New York, USA, 2004.
9
“MOSFETs Move in on Low Voltage Rectification”, MOSPOWER Applications
Handbook, Siliconix, 1984, pp. 5-69 - 5-86.
10
Kutkut, N. H., Divan, D. M., and Gascoigne, R. W., “An Improved Full-Bridge
Zero-Voltage-Switching PWM Converter Using a Two-Inductor Rectifier”, IEEE
Transactions on Industry Applications, Vol. 31, No. 1, January/February 1995,
pp. 119 - 126
11Alivio,
G., Ambrus, J., McDonald, T., and Dowling, R., “Maximizing the Effectiveness
of your SMD Assemblies”, Application Note AN-994, International Rectifier.
12
Wei, J. and Lee, F. C., “An Output Impedance-based Design of Voltage Regulator
Output Capacitors for High Slew-rate Load Current Transients”, Proceedings of IEEE
APEC 2004, pp. 304 - 310.
13
“12 W Input Universal CV Adapter”, Power Integrations Design Idea DI-91, January
2006.
© 2008 Microchip Technology Inc.
DS70320B-page 109
SMPS AC/DC Reference Design User’s Guide
NOTES:
DS70320B-page 110
© 2008 Microchip Technology Inc.
SMPS AC/DC REFERENCE
DESIGN USER’S GUIDE
Index
A
P
Auxiliary Power Supply ............................................ 45
Auxiliary Software .................................................... 64
Fault Source Identification ................................ 65
Input AC Undervoltage Protection .................... 65
Output Sequencing ........................................... 64
Overtemperature Protection ............................. 65
Soft-Start and Soft-Shutdown ........................... 65
PCB Layout
Power Board ..................................................... 75
Signal Board ..................................................... 76
PFC Boost Converter ......................................... 21, 25
Power Factor (PF)...................................................... 9
Power Factor Correction (PFC).................................. 8
B
Reactive Power .......................................................... 9
Reading, Recommended ........................................... 4
Recommended Test Equipment............................... 67
Resources Required for Digital PFC ........................ 21
Resources Required for Digital Phase-Shift ZVT Converter .................................................................... 22
Resources Required for Synchronous
Buck Converters ................................................... 24
Boost PFC Circuit .................................................... 12
Buck Converter ........................................................ 17
Buck PFC Circuit...................................................... 11
Buck/Boost PFC Circuit............................................ 12
C
Common Resources for PFC and
Phase-Shift ZVT Converter .................................. 23
Continuous Output Rating.......................................... 7
Control Scheme
Phase-Shift ZVT ............................................... 56
Control Software ...................................................... 52
ADC Interrupt Service Routine ......................... 53
Fault Loop ......................................................... 53
Idle Loop (Normal Operation) ........................... 53
Initialization Routine.......................................... 53
Peripheral Enable Routine ................................ 53
PFC Boost Converter........................................ 54
Secondary Side (DC_DC)................................. 61
Customer Notification Service.................................... 5
Customer Support ...................................................... 5
D
Documentation
Conventions ........................................................ 3
Layout ................................................................. 1
F
Flyback Converter .................................................... 20
Full-Bridge Converter ............................................... 13
Full-Bridge ZVT Converter ....................................... 30
I
Input and Output Electrical Specifications................ 68
Input and Output Specifications ................................. 7
Internet Address......................................................... 4
R
S
Safety Isolation Information...................................... 68
Schematics
Power Board (Sheet 1 of 8) .............................. 77
Power Board (Sheet 2 of 8) .............................. 78
Power Board (Sheet 3 of 8) .............................. 79
Power Board (Sheet 4 of 8) .............................. 80
Power Board (Sheet 5 of 8) .............................. 81
Power Board (Sheet 6 of 8) .............................. 82
Power Board (Sheet 7 of 8) .............................. 83
Power Board (Sheet 8 of 8) .............................. 84
Signal Board (Sheet 1 of 7)............................... 85
Signal Board (Sheet 2 of 7)............................... 86
Signal Board (Sheet 3 of 7)............................... 87
Signal Board (Sheet 4 of 7)............................... 88
Signal Board (Sheet 5 of 7)............................... 89
Signal Board (Sheet 6 of 7)............................... 90
Signal Board (Sheet 7 of 7)............................... 91
Single-Phase Synchronous Buck Converter ............ 41
Synchronous Buck Converters................................. 23
Synchronous Rectifier Design.................................. 37
System Connections ................................................ 70
Input .................................................................. 70
Output ............................................................... 70
Programming .................................................... 71
System Evaluation and Testing................................ 72
System Power-Up .................................................... 72
M
Microchip Internet Web Site ....................................... 4
Multi-Phase Synchronous Buck Converter .............. 19
O
Output Performance Testing .................................... 73
© 2008 Microchip Technology Inc.
DS70320B-page 111
SMPS AC/DC Reference Design User’s Guide
T
Three-Phase Synchronous Buck Converter............. 43
Time Intervals
t0 < t < t1........................................................... 16
t1 < t < t2........................................................... 16
t2 < t < t3........................................................... 16
t3 < t < t4........................................................... 16
Total Harmonic Distortion (THD) ................................ 9
Typical Power Meter Connections............................ 72
W
Warranty Registration................................................. 4
Working Power ........................................................... 9
WWW Address ........................................................... 4
Z
Zero Current Switching (ZCS) .................................. 14
Zero Voltage Transition (ZVT).................................. 14
DS70320B-page 112
© 2008 Microchip Technology Inc.
Index
NOTES:
© 2008 Microchip Technology Inc.
DS70320B-page 113
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© 2008 Microchip Technology Inc.