### PFC2 - Compensating a PFC Stage

```Compensating a PFC Stage
Agenda
• Introduction
• Deriving a small-signal model
– General method
– Practical example: NCP1605-driven PFC stages
• Compensating the loop
–
–
–
–
Type-2 compensation
Influence of the line and power level
Computing the compensation
Practical example
• Conclusion
Output Voltage Low Frequency Ripple
Vout
Pin(t)
Pin,avg
+
Iin(t)
Vin(t)
 The load power demand is matched in average only
 A low frequency ripple is inherent to the PFC function
PFC Stages are Slow Systems…
 The output ripple must be filtered to avoid current
distortion.
 In practice, the loop frequency is selected in the
range of 20 Hz, which is very low.
 Even if the bandwidth is low, the loop must be
compensated!
Agenda
• Introduction
• Deriving a small-signal model
– General method
– Practical example: NCP1605-driven PFC stages
• Compensating the loop
–
–
–
–
Type-2 compensation
Influence of the line and power level
Computing the compensation
Practical example
• Conclusion
A Simple Representation
• We will consider the PFC stage as a system delivering a
power under an input rms voltage and a control signal
Pout
Vin(rms)
PFC stage
Vcontrol
• Details of the power processing are ignored:
• Operation mode (CrM, CCM, Voltage or Current mode…)
• 100% efficiency, only the average power contribution of the sinusoidal
signals is considered
A Simple Large Signal Model
• Let’s represent the PFC stage as a current source
delivering the power to the bulk capacitor and the load:
ID =
Pin ( avg )
Vout
Bulk
Capacitor
rC
Cbulk
• Pin(avg) depends on Vcontrol (always), on Vin(rms) (in the
absence of feedforward) and sometimes on Vout
• 3 possible sources of perturbations: Vcontrol, Vout and Vin(rms).
NCP1605
• Frequency Clamped Critical Conduction Mode (FCCrM)
• Key features for a master PFC:
• High voltage current source, Soft-SkipTM during standby mode
• “pfcOK” signal, dynamic response enhancer
• Bunch of protections for rugged PFC stages
• Markets: high power AC adapters, LCD TVs
Rbo1
Rout1
Rout2
Vout
STBY control
Rbo2
HV
1
16
2
15
3
14
Vcc
Cbo
CVctrl
FB
L1
Rovp1
Vout
OVP
Rovp2
D1
13
4
CVref
5
12
6
11
7
10
8
9
Rzcd
Ac line
pfcOK
Cin
Ct
EMI
Filter
Vcc
M1
Cbulk
Cosc
pfcOK
Rocp
Rdrv
Icoil
Rcs
Communication signals
NCP1605 – Follower Boost
• Voltage mode operation: the circuit adjusts the power level
by modulating the MOSFET conduction time
• The charge current of the timing capacitor is proportional
to the FB square and hence to (Vout)2:
Ich arg e
⎛ Vout
= It ⋅ ⎜
⎜ Vout ,nom
⎝
⎞
⎟
⎟
⎠
2
where :
 Vout,nom is the Vout regulation voltage
 It is a 370-µA current source
• The on-time is inversely proportional to (Vout)2 allowing the
Follower boost function:
2
ton
Ct ⋅Vton
=
It
⎛ Vout ,nom
⋅ ⎜⎜
⎝ Vout
⎞
⎟⎟
⎠
NCP1605 - Power Expression
200 µA
0.955*Vref
FB
+
Error Amplifier
Vref
+/-20µA
+
• The control signal is VF offset
down and divided by 3 to form
VREGUL used in the PWM section
pfcOK
Vou t low
detect
OVLflag1
Vcon trol
VF
OFF
2R
VF
VREGUL
3V
R
• Hence due to the follower boost function, the power is
inversely dependent on (Vout)2:
Pin( avg )
Ct ⋅ Vin( rms )2 ⎛ Vout ,nom
=
⋅ ⎜⎜
2 ⋅ L ⋅ It
⎝ Vout
⎞ (Vcontrol − VF )
⎟⎟ ⋅
3
⎠
2
NCP1605 - Large Signal Model
• Let’s represent the PFC stage as a current source delivering
the power to the bulk capacitor and the load:
ID =
Pin ( avg )
Replacing Pin,avg by its
expression of slide 10
rC
Vout
2
2⎞ ⎛
⎛ C ⋅V
V
⋅ (Vcontrol − VF ) ⎞
(
)
in
rms
,
t
out
nom
⎟
⎟⋅⎜
ID = ⎜
3
⎜ 6 ⋅ L ⋅ It
⎟ ⎜
⎟
Vout
⎝
⎠ ⎝
⎠
constants
Time varying terms
Cbulk
• 3 sources of perturbations: VCONTROL, Vout and Vin(rms).
Small Signal Model
• A large signal model is nonlinear because ID is formed of the
multiplication and division of Vcontrol, Vin,rms and Vout.
• This model needs to be linearized to assess the AC
contribution of each variable
• The model is perturbed and linearized around a quiescient
operating point (DC point)
Considering Variations Around the DC Value…
• Let’s omit the perturbations of the line magnitude (assumed
constant)
• Let’s consider small variations around the DC values for
∂ID \$
Vout and Vcontrol:
\$i = ∂ID ⋅ v\$
+
⋅v
D
∂Vcontrol
CONTROL
∂Vout
out
• We then obtain:
Vout ,nom + v\$ out
rC
ID + \$i D
where : \$i D =
∂ID
∂ID \$
⋅ v\$ control +
⋅v
∂Vcontrol
∂Vout out
Cbulk
Deriving a Small Signal Model…
• The DC portion can be eliminated
• The partial derivatives are to be computed at the DC point
that is for:
– Vcontrol that is the control signal DC value for the considered working point
– Vout,nom that is the nominal (DC) output voltage
• Replacing the derivations by their expression, we obtain:
v\$ out
rC
I1 =
∂ID \$
⋅ v out
∂Vout
I1 computed for
Vcontrol DC point
I2 =
∂ID
⋅ v\$ control
∂Vcontrol
I2 computed for Vout DC point that is Vout,nom
Cbulk
Contribution of the Vout Perturbations
• Depending on the controller scheme
ID =
Pin,avg
Vout
=
(
f Vin( rms ),Vcontrol
(Vout )
)
• n=0 for NCP1607
• n=1 for NCP1654 (predictive CCM PFC for which
• n=2 for NCP1605 (follower boost – see slide 10)
• At the DC point
Vout = Vout ,nom
where n = 0,1 or 2
n +1
and
Pin,avg ∝
Vcontrol ⋅Vin,rms
Vout
Pin(avg )
(
Vout ,nom
)
2
=
)
1
• Finally:
(
( n + 1) ⋅ f Vin(rms ),Vcontrol
∂ID \$
I1 =
⋅ v out = −
∂Vout
(V )n + 2
out
)
⋅ v\$ out = −
Vout =Vout ,nom
( n + 1) ⋅ Pin(avg )
(Vout,nom )
2
⋅ v\$ out = −
( n + 1) ⋅ v\$
out
2 Resistors…
• Hence, the small signal model can be simplified as follows:
I2 =
∂ID
⋅ v\$ control
∂Vcontrol
v\$ out
rC
n +1
Cbulk
Vout AC contribution
•
Noting that:
n +1
n+2
the model can be further simplified
Finally…
 The small signal model is:
I2 =
⋅
n+2
1 + s ⋅ rC ⋅ Cbulk
⋅C
⎛R
⎞
1 + s ⎜ LOAD bulk ⎟
n+2
⎝
⎠
v\$ out
∂ID
⋅ v\$ control
∂Vcontrol
where : ID =
Z (s ) =
rC
⎜
⎟
⎝ n+2 ⎠
Pin,avg
Vout
Cbulk
 The transfer function is:
v\$ out
v\$
control
R
n+2
⎛ ∂ID
⋅ ⎜⎜
⎝ ∂Vcontrol
⎞
1 + s ⋅ rC ⋅ Cbulk
⋅
⎟⎟
⎠ 1 + s ⎛ RLOAD ⋅ Cbulk ⎞
⎜
⎟
n+2
⎝
⎠
NCP1605 Example
• The large signal model instructed that:
ID =
Pin ( avg )
Vout
2
2⎞ ⎛
⎛ C ⋅V
⋅ (Vcontrol − VF ) ⎞
V
(
)
in
rms
,
t
out
nom
⎟
⎟⋅⎜
=⎜
3
⎜ 6 ⋅ L ⋅ It
⎟ ⎜
⎟
V
out
⎝
⎠ ⎝
⎠
• Hence:
(Vout )n +1
n=2
(
)
2
Ct ⋅ Vin ( rms )
∂ID
=
∂Vcontrol 6 ⋅ L ⋅ It ⋅ Vout ,nom
term
NCP1605 - Small Signal Model
• Finally:
(
)
v\$ out
rC
2
Ct ⋅ Vin ( rms )
I2 =
⋅ v\$ CONTROL
6 ⋅ L ⋅ It ⋅ Vout ,nom
⎜
⎟
⎝ 4 ⎠
Cbulk
• The transfer function is:
v\$ out
v\$ CONTROL
(
RLOAD ⋅ Ct ⋅ Vin ( rms )
=
24 ⋅ L ⋅ It ⋅ Vout ,nom
)
2
⋅
1 + s ⋅ rC ⋅ Cbulk
⋅C
⎛R
⎞
1 + s ⋅ ⎜ LOAD bulk ⎟
4
⎝
⎠
Power Stage Characteristic – Bode Plots
((
))
2
⎛⎛
2⎞
⋅
C
⋅
V
⎜⎜ R
LO
t
in
rm
s
(
)
R
⋅ C ⋅ Vin ( rms ) ⎟
20 ⋅ log ⎜⎜ LOAD t
⎟
20 ⋅ log⎜ 24 ⋅ L ⋅ I ⋅ V 2
tK out⋅ V
,nom3 ⎟
⎜
µ
⋅
L
⋅
1440
⎜
FB
out
⎟
⎝⎜
⎝
⎠
⎞
⎟
⎟
⎟
⎟
⎠
Asymptotic
representation
-20 dB/dec
Gain (dB)
Frequency (Hz)
0°
0°
Phase (°)
-90°
Frequency (Hz)
fp 0 =
2
fz 0 =
1
2π ⋅ rC ⋅ Cbulk
Agenda
• Introduction
• Deriving a small-signal model
– General method
– Practical example: NCP1605-driven PFC stages
• Compensating the loop
–
–
–
–
Type-2 compensation
Influence of the line and power level
Computing the compensation
Practical example
• Conclusion
Compensation Phase Boost
• The zero brought by the bulk capacitor ESR is too high to
bring some phase margin. It is ignored.
• The PFC open loop inherently causes a -360°phase shift:
– Power stage pole
– Error amplifier inversion
– Compensation origin pole
Î -90°
Î -180°
Î -90°
• The compensation must then provide some phase boost
• A type-2 compensation is recommended
Type-2 Compensation
• The NCP1605 embeds a transconductance error amplifier
(OTA)
1
V CONTROL
VOUT
fz1 =
I CONTROL
2π ⋅ R1 ⋅ C1
R1
C1
C2
1
2π ⋅ R1 ⋅ C2
1
f p1 =
2π ⋅ R0 ⋅ C1
fp 2 =
RfbU
OTA
FB
to PWM
comparator
RfbL
pole at the origin
V REF
C2 << C1
– No direct influence of the
RfbU impedance on the compensation
– Only the feedback scale factor interferes
R0 =
Vout ,nom
Vref ⋅ GEA
• Vref is the reference voltage
(generally 2.5 V in ON semi devices)
• GEA is the OTA
(200-µS transconductance gain for
NCP1605, NCP1654 and NCP1631)
Type-2 Characteristic - Example
 fp2 and fz1 set the
phase boost magnitude
and location (frequency)
40.0
0 dB
0
Gc
Gain (dB)
-40.0
-80.0
51
40 dB
-120
 The phase boost
peaks at: ( fPhB = fz1 ⋅ fp2 )
that is 27 Hz
 The phase boost is:
270
⎛ fphB
tan−1 ⎜⎜
⎝ fz
phase boost (60° )
225
45°
⎛
⎞
−1 fphB
⎜
tan
−
⎟⎟
⎜ fp
⎠
⎝
⎞
⎟
⎟
⎠
180
Phase (°)
135
-270 °
90.0
10m
52
100m
1
10
100
1k
frequency in hertz
fz1: compensation zero (6 Hz)
10k
100k
 The origin pole fp1
the phase boost
frequency
fp2: high frequency pole (90 Hz)
fz1 ⋅ fp 2
Phase Boost at the Crossover Frequency
φB =
 The lower fz1 and/or the
higher fp2, the higher the phase
boost (max. value: 90°)
15°
φ’B
φB
-270°
0m
1
10
f
fz1= f’z1 fc
100
i h
fp2
1k
f’p2
10k
⎛
⎞
⎞
−1 fc
⎟
⎜⎜ ⎟⎟ − tan ⎜⎜
⎟
⎝ fz1 ⎠
⎝ fp 2 ⎠
−1 ⎛ fc
tan
100
 Assuming the PFC power
stage pole is well below the
crossover frequency (fc), the
phase boost equates the phase
margin (φm=φB)
 Target a phase boost
between 45° and 75°
Gain Considerations
•
20 dB
A low
frequency
attenuates
more
sharply
the line
ripple
•
•
Lower gain
with a low
frequency
zero
•
f’z1
fz1
fp2
fc
f’p2
2.fline
In the red trace, the
distance between
the zero and the
pole frequencies is
increased
Both characteristics
generate the same
attenuation at the
crossover frequency
The lower the fz1
frequency, the lower
the gain in the low
frequency region
The higher fp2, the
lower the (2.fline)
ripple rejection
Type-2 Compensator - Summary
• The zero should not be placed at a too low frequency (not to
penalize the low-frequency gain)
• The high frequency pole must be placed at a frequency low
enough to attenuate the line ripple
• The phase boost (and phase margin) depends on the zero
and high-frequency pole locations
• The origin pole is set to force the open loop gain to zero at
the targeted crossover frequency
Agenda
• Introduction
• Deriving a small-signal model
– General method
– Practical example: NCP1605-driven PFC stages
• Compensating the loop
–
–
–
–
Type-2 compensation
Influence of the line and power level
Computing the compensation
Practical example
• Conclusion
Compensating for the Full Range?...
• The static gain depends on the load and if there is no
feedforward, on the line magnitude
⎛R
Gstatic ( dB ) = 20 ⋅ log ⎜ LOAD
⎜ n+2
⎝
⎛ ∂ID
⋅ ⎜⎜
⎝ ∂Vcontrol
(
⎛
⎜ RLOAD ⋅ Ct ⋅ Vin ( rms )
⎞⎞
⎟⎟ ⎟⎟ = 20 ⋅ log ⎜
⎜ 24 ⋅ L ⋅ It ⋅ Vout ,nom
⎠⎠
⎜
⎝
)
2
⎞
⎟
⎟
⎟
⎟
⎠
(NCP1605)
• The power stage pole varies as a function of the load:
fp 0 =
n+2
=
2
• What is the worst case when closing the loop?
(NCP1605)
Load Influence on the Open Loop Plots
- 20 dB/dec
20 ⋅ log(α )
Static gain
Gain (dB)
Unchanged Gain and
Phase at the targeted
crossover frequency
Frequency (Hz)
-0°
Phase (°)
-90°
Frequency (Hz)
Asymptotic
representation
fp 0 2 =
fp 0 1
α
fc
fz0 =
1
2π ⋅ rC ⋅ Cbulk
fc and φm are not affected!
Line Influence on the Open Loop Plots
• No feedforward (e.g. NCP1607) and
(Vin(rms)2 = β ⋅Vin(rms)1
)
with β > 1
- 20 dB/dec
40 ⋅ log( β )
Static gain
Gain (dB)
Vin(rms)1
Vin(rms)2
Unchanged Phase
but increased gain
(multiplied by β*β )
Frequency (Hz)
-0°
Phase (°)
-90°
Asymptotic
representation
fp 0 =
n+2
2π ⋅ R LOAD ⋅ C bulk
Frequency (Hz)
fc
fz0
1
=
2π ⋅ rC ⋅ Cbulk
The loop crossover frequency is β 2 increased
– Same crossover frequency at lighter loads
– The zero frequency is set optimally (not at a too low frequency)
• Compensate at high line
– High line is the worst case as in the absence of feedforward, the
2
static gain is proportional to (Vin( rms ) )
( fc )HL
(
(
⎛ V
⎜ in ( rms )
=⎜
⎜⎜ Vin ( rms )
⎝
)
)
2
HL
LL
⎞
⎟
⎟ ⋅ ( fc )LL
⎟⎟
⎠
Where HL stands for Highest Line and LL for Lowest Line
– In universal mains applications, the high-line crossover frequency is
2
9 times higher than the low-line one:
⎛ 265 ⎞
( fc )HL = ⎜
⎟ ⋅ ( fc )LL ≅ 9 ⋅ ( fc )LL
⎝ 90 ⎠
Crossover Frequency Selection
• In the absence of feedforward, ( fc )HL ≤ fline is a good option
f
• With feedforward, ( fc )HL ≤ line
is rather selected for a better
2
attenuation of the low frequency ripple
• Get sure that on the line range, the PFC boost pole remains
lower than the crossover frequency at full load!
fp0 ≤ ( fc )LL
• If not, increase Cbulk
Agenda
• Introduction
• Deriving a small-signal model
– General method
– Practical example: NCP1605-driven PFC stages
• Compensating the loop
–
–
–
–
Type-2 compensation
Influence of the line and power level
Computing the compensation
Practical example
• Conclusion
Compensation Techniques
• Several techniques exist:
 manual placement, “k factor” (Venable)…
+ Systematic
- The PFC boost gain is to be computed at fc
fp 2
- No flexibility in the zero and high pole locations fc = k ⋅ fz1 =
k
 Pole and zero cancellation:
9 Place the compensation zero so that it cancels the power stage pole:
9 Force the pole at the origin to cancel the PFC boost gain when (f = fc)
9 Adjust the phase margin with the high frequency pole
Pole and Zero Cancellation…
K0
-20 dB/dec
Gain (dB)
Frequency (Hz)
Power stage
Open Loop
ESR of the
bulk capacitor
-40 dB/dec
0°
-90°
Phase (°)
-180°
-270°
φm
-360°
fz1 = fp0
Frequency (Hz)
fc
fp 2
2 ⋅ fline
fz 0
 The higher fp2, the larger the phase margin
 The lower fp2, the better the rejection of the low frequency ripple
 φm = 45° if fp2 = fc .
Poles and Zero Placement
• Design the compensation for full load, high line:
• Place the origin pole to cancel
K0, the static gain at fc:
fp 0 =
fc
K0
where :
for
v\$ out
v\$ CONTROL
• Place the zero so that it cancels
the PFC boost pole
( fz1 = fp0 )
= K0 ⋅
1 + s ⋅ rC ⋅ Cbulk
1 + s ⋅ ⎜⎜
n+2
⎝
for
• Place fp2 to obtain the targeted phase margin:
⎞
⎟⎟
⎠
fp 2 =
fc
tan ( 90° − φm )
Example
• A wide mains, 150-W application driven by the NCP1605
• Vout,nom = 390 V
R
⋅ C ⋅ (V ( ) )
v\$
1+ s ⋅ r ⋅ C
=K ⋅
where : K =
⋅C
24 ⋅ L ⋅ I ⋅V
⎛R
⎞
v\$
• (Vin(rms))LL = 90 V
1+ s ⋅ ⎜
⎟
4
⎝
⎠
• (Vin(rms))HL = 265 V
(V
) = 390 ≅ 1 k Ω
R
• L = 150 µH
( ) = (P )
150
• Ct = 4.7 nF
V
390
R =
=
= 780 k Ω (OTA)
V ⋅G
2.5 ⋅ 200 ⋅ 10
• Cbulk = 100 µF
• rC = 500 mΩ (ESR)
⋅ C ⋅ (V ( ) )
R
K
10 ⋅ 4.7 ⋅ 10 ⋅ 265
=
≅ 2.59 µF
C =
=
⋅
f
⋅
R
⋅
⋅
L
⋅
I
⋅
V
⋅
⋅
2
f
R
2
24
2
50
780k ⋅ 24 ⋅ 150 µ ⋅ 370 µ ⋅ 390
π
π
π
⋅
⋅
• fc = 50 Hz
and Φm = 60°
R
⋅C
10 ⋅ 100 ⋅ 10
R =
=
≅ 11.36 k Ω ==> 12 k Ω
n
+
⋅
C
2
(
)
( 2 + 2) ⋅ 2.2 ⋅ 10
@ high line (265 V)
2
out
C
0
bulk
CONTROL
2
t
in rms
t
out ,nom
0
bulk
2
out ,nom
out max
out ,nom
−6
0
ref
EA
2
0(min)
t
in rms
1
c
0
c
0
t
1
fp1 =
tan ( 90° − φm )
2π ⋅ fc ⋅ R1
2
out ,nom
−6
1
C2 =
−9
−6
3
bulk
3
HL
=
tan ( 90° − 60° )
==>
150 nF
1
= 6 Hz
2π ⋅ R1 ⋅ C1
fz1 =
2π ⋅ 50 ⋅ 12 ⋅ 103
1
= 93 mHz
2π ⋅ R0 ⋅ C1
fz1 =
≅ 153 nF
1
= 88 Hz
2π ⋅ R1 ⋅ C2
==>
2.2 µF
Simulation Validation
• The simulation circuit is based on the large signal model:
Vout
Vout
B6
Current
R10
50m
{Ct*Vbulk*Vbulk*Vrms*Vrms}*V(control)/(6*{L}*370u*V(Vout)*V(Vout)*V(Vout))
C5
100u
IC = {Vrms*1.414}
{Vbulk*Vbulk/Pout}
1
L1
1kH
C6
1kF
Vin
Large signal model
of the NCP1605driven PFC stage
6
V4
AC = 1
Generation and
injection of the ac
perturbation
EAout
R4
{Rupper}
7
control
B1
Current
R1
12k
B5
Voltage
V(EAout)
5
R2
100
EAout
FB
{gm}*(2.5-V(FB))
4
C3
2.2u
C1
150nF
R3
{Rlower}
Feedback and
regulation circuit
(including type-2
compensation)
Open Loop Characteristic – Full Load
fc= 52 Hz @ Vin(rms) = 265 V
fc= 7 Hz @ Vin(rms) = 90 V
Vin(rms) = 90 V
0 dB
Gain (dB)
Vin(rms) = 265 V
40 dB
10 mHz
100 mHz
1 Hz
10 Hz
100 Hz
1 kHz
φm = 62°
Phase (°)
10 kHz
6
100 kHz
4
45 °
φm = 87°
0°
Open Loop Characteristic – Mid Load
fc= 52 Hz @ Vin(rms) = 265 V
fc= 8 Hz @ Vin(rms) = 90 V
Vin(rms) = 90 V
0 dB
Gain (dB)
Vin(rms) = 265 V
40 dB
1
2
10 mHz
100 mHz
1 Hz
10 Hz
100 Hz
1 kHz
φm = 58°
Phase (°)
10 kHz
100 kHz
4
3
45 °
φm = 69°
0°
•
•
A 19 V / 7 A loads the PFC stage
The downstream converter swings between 6.3 A and 7.7 A (+/-10%)
with a 2 A/µs slope
Ac line current (5 A/div)
Vin,rms = 90 V
Bulk Voltage (20 VA/div – 380-V offset)
372 V < Vbulk < 396 V
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Ac line current (2 A/div)
Vin,rms = 265 V
Bulk Voltage (20 VA/div – 380-V offset)
375 V < Vbulk < 394 V
The high-line, larger bandwidth reduces the Vbulk deviations and speedsup the output voltage recovery
•
•
A 19 V / 7 A loads the PFC stage
The downstream converter swings between 3.1 A and 3.9 A (+/-10%)
with a 2 A/µs slope
Ac line current (2 A/div)
Vin,rms = 90 V
Bulk Voltage (20 VA/div – 380-V offset)
376 V < Vbulk < 392 V
•
Ac line current (1 A/div)
Vin,rms = 265 V
Bulk Voltage (20 VA/div – 380-V offset)
379 V < Vbulk < 390 V
The circuit still exhibits a first order response
•
•
A 19 V / 7 A loads the PFC stage
The downstream converter swings from 7.0 A to 3.5 A (2 A/µs slope)
Ac line current (2 A/div)
Vin,rms = 90 V
OVP
365 V < Vbulk < 411 V
Bulk Voltage (20 VA/div – 380-V offset)
Ac line current (2 A/div)
365 V < Vbulk < 404 V
Bulk Voltage (20 VA/div – 385-V offset)
Vcontrol (1 V/div)
Vcontrol (2 V/div)
Vin,rms = 230 V
The dynamic response enhancer speeds-up the loop reaction in case of a large undershoot
Implemented in NCP1605 (FCCrM), NCP1654 (CCM) and NCP1631 (Interleaved)
•
The dynamic response enhancer reduces the undershoot at low line
Agenda
• Introduction
• Deriving a small-signal model
– General method
– Practical example: NCP1605-driven PFC stages
• Compensating the loop
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–
–
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Type-2 compensation
Influence of the line and power level
Computing the compensation
Practical example
• Conclusion
Conclusion
• General considerations were illustrated by the case of
NCP1605-driven PFC stages
• A small signal model of PFC boosts can be easily derived
• The proposed method is independent of the operating mode
• A type-2 compensation is recommended
• If no feed-forward is implemented, the loop bandwidth and
phase margin vary as a function of the line magnitude
• The crossover frequency does not vary as a function of the