MIC2125/6 28V Synchronous Buck Controllers Featuring Adaptive ON-Time Control Features General Description • Hyper Speed Control Architecture Enables: - High delta V operation (VIN = 28V and VOUT = 0.6V) - Any Capacitor™ stable • 4.5V to 28V Input Voltage • Adjustable Output Voltage from 0.6V to 24V • 200 kHz to 750 kHz Programmable Switching Frequency The MIC2125 and MIC2126 are constant-frequency synchronous buck controllers featuring a unique adaptive ON-time control architecture. The MIC2125/6 operate over an input voltage range from 4.5V to 28V and can be used to supply load current up to 25A. The output voltage is adjustable down to 0.6V with a guaranteed accuracy of ±1%. The device operates with programmable switching frequency from 200 kHz to 750 kHz. Package Type 2015 Microchip Technology Inc. 16 15 14 13 12 AGND 11 NC 3 10 OVP 4 9 BST VDD 1 PVDD 2 ILIM EP 17 5 6 7 8 SW DL FB MIC2125/6 16-Pin 3 mm x 3 mm QFN (ML) PG Networking/Telecom Equipment Base Stations, Servers Distributed Power Systems Industrial Power Supplies DH • • • • EN Applications The MIC2125/6 offer a full suite of features to ensure protection of the IC during fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, “hiccup” mode short-circuit protection, and thermal shutdown. FREQ • • • HyperLight Load® architecture provides the same high efficiency and ultra-fast transient response as the Hyper Speed Control® architecture under medium to heavy loads. It also maintains high efficiency under light load conditions by transitioning to variable frequency, discontinuous conduction mode operation. VIN • HyperLight Load® (MIC2125) Hyper Speed Control® (MIC2126) Enable Input and Power Good Output Built-in 5V Regulator for Single-Supply Operation Programmable current limit and “hiccup” mode short-circuit protection 7 ms internal soft-start, internal compensation, and thermal shutdown Supports Safe Start-Up into a Prebiased Output –40°C to +125°C Junction Temperature Range Available in 16-pin, 3 mm × 3 mm QFN Package PGND • • • • • DS20005459B-page 1 MIC2125/6 Typical Application Circuit MIC2125/6 3x3 QFN VIN 4.5V TO 28V FREQ PVDD 2.2μF ×3 VIN VDD 220μF BST MIC2125/6 4.7μF AGND 0.1μF DH VOUT 3.3V/20A 0.72μH EN EN SW 90.9kΩ PG PG 470pF 10kΩ 100μF 470μF DL VOUT 56.2kΩ 0.1μF OVP PGND 10kΩ FB 2.26kΩ ILIM 1.2kΩ Functional Block Diagram MIC2125/26 PVDD VIN VDD 4.5V TO 28V VIN LDO R19 4.7μF FIXED TON UVLO 220μF R20 BST VIN 2.2μF ×2 FREQ ESTIMATE VDD MODIFIED TOFF HSD DH Q1 0.1μF 0.72μH 100kΩ SW VOUT 3.3V/20A CONTROL EN EN PVDD LOGIC 470pF TIMER SOFT–START LSD DL 90.9kΩ Q3 1.2kΩ R1 10kΩ 100μF 470μF 0.1μF PGND SOFT START CL DETECTION ILIM R2 2.26kΩ THERMAL SHUTDOWN OVP COMPENSATION VREF 0.6V VDD gm EA FB COMP 49.9kΩ 8% PG PG VREF 0.6V AGND 92% DS20005459B-page 2 2015 Microchip Technology Inc. MIC2125/6 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings † VIN .............................................................................................................................................................. –0.3V to +30V VDD, PVDD .................................................................................................................................................... –0.3V to +6V VSW, VFREQ, VILIM, VEN ....................................................................................................................–0.3V to (VIN +0.3V) VBST to VSW ................................................................................................................................................... –0.3V to 6V VBST ............................................................................................................................................................. –0.3V to 36V VPG................................................................................................................................................. –0.3V to (VDD + 0.3V) VFB ................................................................................................................................................. –0.3V to (VDD + 0.3V) PGND to AGND ........................................................................................................................................... –0.3V to +0.3V ESD Rating(1) ............................................................................................................................................................. 2 kV Operating Ratings ‡ Supply Voltage (VIN) ...................................................................................................................................... 4.5V to 28V VSW, VFREQ, VILIM, VEN ......................................................................................................................................0V to VIN † Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended periods may affect device reliability. ‡ Notice: The device is not guaranteed to function outside its operating ratings. Note 1: Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5 kΩ in series with 100 pF. 2015 Microchip Technology Inc. DS20005459B-page 3 MIC2125/6 TABLE 1-1: ELECTRICAL CHARACTERISTICS Electrical Characteristics: VIN = 12V, VOUT = 1.2V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C. (Note 1). Parameters Min. Typ. Max. Units Conditions Input Voltage Range (VIN) (Note 2) 4.5 — 5.5 V 4.5 — 28 Quiescent Supply Current (MIC2125) — 340 750 µA VFB = 1.5V Quiescent Supply Current (MIC2126) — 1.1 3 mA VFB = 1.5V Shutdown Supply Current — 0.1 5 µA SW unconnected, VEN = 0V VDD Output Voltage 4.8 5.2 5.4 V VIN = 7V to 28V, IDD = 10 mA VDD UVLO Threshold 3.7 4.2 4.5 VDD UVLO Hysteresis — 400 — mV — Load Regulation 0.6 2 3.6 % IDD = 0 to 40 mA 0.597 0.6 0.603 V TJ = 25°C (±0.5%) 0.594 0.6 0.606 — 0.01 0.5 µA VFB = 0.6V EN Logic Level High 1.6 — — V — EN Logic Level Low — — 0.6 EN Hysteresis — 120 — mV — EN Bias Current — 6 30 µA VEN = 12V — 750 — kHz VFREQ = VIN — 375 — Maximum Duty Cycle — 85 — Minimum Duty Cycle — 0 — Minimum On-Time — 100 — Minimum Off-Time 150 220 300 — 7 — ms — Current-Limit Comparator Offset –15 –4 7 mV VFB = 0.6V Current-Limit Source Current 32 36 40 µA VFB = 0.6V Power Supply Input VDD = VIN — VDD Supply VDD rising Reference Feedback Reference Voltage FB Bias Current –40°C ≤ TJ ≤ +125°C (±1%) Enable Control — Oscillator Switching Frequency VFREQ = 50% x VIN % — VFB > 0.6V ns — — Soft-Start Soft-Start Time Short-Circuit Protection and OVP Note 1: 2: Specification for packaged product only. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH. DS20005459B-page 4 2015 Microchip Technology Inc. MIC2125/6 TABLE 1-1: ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Characteristics: VIN = 12V, VOUT = 1.2V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C. (Note 1). Parameters Min. Typ. Max. Units —— 0.62 — V — DH, DL Output Low Voltage — — 0.1 V ISINK = 10 mA DH, DL Output High Voltage VPVDD-0.1 or VBST-0.1 — — DH On-Resistance, High State — 2.5 — DH On-Resistance, Low State — 1.6 — — DL On-Resistance, High State — 1.9 — — DL On-Resistance, Low State — 0.55 — — SW, BST Leakage Current — — 50 µA PG Threshold Voltage 85 89 95 %VOUT PG Hysteresis — 6 — PG Delay Time — 80 — µs Sweep VFB from low to high PG Low Voltage — 60 200 mV VFB < 90% x VNOM, IPG = 1 mA Overtemperature Shutdown — 150 — °C TJ Rising Overtemperature Shutdown Hysteresis — 15 — °C — Overvoltage Protection Threshold Conditions FET Drivers ISOURCE = 10 mA Ω — — Power Good (PG) Sweep VFB from low to high Sweep VFB from high to low Thermal Protection Note 1: 2: Specification for packaged product only. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH. 2015 Microchip Technology Inc. DS20005459B-page 5 MIC2125/6 TEMPERATURE SPECIFICATIONS Parameters Sym. Min. Typ. Max. Units Conditions Junction Operating Temperature TJ –40 — +125 °C Note 1 Temperature Ranges Storage Temperature Range TS –65 — +150 °C — Junction Temperature TJ — — +150 °C — Lead Temperature — — — +260 °C Soldering, 10s JA — 50.8 — °C/W — JC — 25.3 — °C/W — Package Thermal Resistances Thermal Resistance 3 mm x 3 mm QFN-16LD Note 1: The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the maximum allowable power dissipation will cause the device operating junction temperature to exceed the maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability. DS20005459B-page 6 2015 Microchip Technology Inc. MIC2125/6 2.0 Note: TYPICAL PERFORMANCE CURVES The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. FIGURE 2-1: VIN Operating Supply Current vs. Input Voltage (MIC2125). FIGURE 2-4: VIN Shutdown Current vs. Input Voltage (MIC2125). FIGURE 2-2: Feedback Voltage vs. Input Voltage (MIC2125). FIGURE 2-5: Input Voltage. FIGURE 2-3: Output Voltage vs. Input Voltage (MIC2125). FIGURE 2-6: Switching Frequency vs. Temperature (MIC2126). 2015 Microchip Technology Inc. Switching Frequency vs. DS20005459B-page 7 MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. .V FIGURE 2-7: VDD Voltage vs. Input Voltage (MIC2125). FIGURE 2-10: VIN Operating Supply Current vs. Temperature (MIC2125). FIGURE 2-8: Enable Threshold vs. Input Voltage (MIC2125). FIGURE 2-11: Feedback Voltage vs. Temperature (MIC2125). FIGURE 2-9: Output Peak Current Limit vs. Input Voltage (MIC2125). FIGURE 2-12: Load Regulation vs. Temperature (MIC2125). DS20005459B-page 8 2015 Microchip Technology Inc. MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. FIGURE 2-13: VIN Shutdown Current vs. Temperature (MIC2125). FIGURE 2-16: EN Bias Current vs. Temperature (MIC2125). FIGURE 2-14: VDD UVLO Threshold vs. Temperature (MIC2125). FIGURE 2-17: VDD Voltage vs. Temperature (MIC2125). FIGURE 2-15: Enable Threshold vs. Temperature (MIC2125). FIGURE 2-18: Current-Limit Source Current vs. Temperature (MIC2125). 2015 Microchip Technology Inc. DS20005459B-page 9 MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. *Note: For Case Temperature graphs: The temperature measurement was taken at the hottest point on the MIC2125/6 case mounted on a 5 square inch PCBn. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components. FIGURE 2-19: Line Regulation vs. Temperature (MIC2125). FIGURE 2-22: Output Regulation vs. Input Voltage (MIC2125). FIGURE 2-20: Feedback Voltage vs. Output Current (MIC2125). FIGURE 2-23: Case Temperature* vs. Output Current (MIC2125). FIGURE 2-21: Line Regulation vs. Output Current (MIC2125). FIGURE 2-24: Case Temperature* vs. Output Current (MIC2125). DS20005459B-page 10 2015 Microchip Technology Inc. MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. *Note: For Case Temperature graphs: The temperature measurement was taken at the hottest point on the MIC2125/6 case mounted on a 5 square inch PCBn. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components. FIGURE 2-25: Case Temperature* vs. Output Current (MIC2125). FIGURE 2-28: Efficiency (VIN = 18V) vs. Output Current (MIC2125). FIGURE 2-26: Efficiency (VIN = 5V) vs. Output Current (MIC2125). FIGURE 2-29: Efficiency (VIN = 5V) vs. Output Current (MIC2126). FIGURE 2-27: Efficiency (VIN = 12V) vs. Output Current (MIC2125). FIGURE 2-30: Efficiency (VIN = 12V) vs. Output Current (MIC2126). 2015 Microchip Technology Inc. DS20005459B-page 11 MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. IN VIN (10V/div) VSW (10V/div) VIN = 12V VOUT = 1.2V IOUT = 0A VPRE-BIAS = 0.5V VOUT (500mV/div) Time (10ms/div) FIGURE 2-31: Efficiency (VIN = 18V) vs. Output Current (MIC2126). FIGURE 2-34: MIC2125 VIN Start-Up with Prebiased Output. IN VIN = 12V VOUT = 1.2V IOUT = 20A VIN (10V/div) VSW (10V/div) VEN (2V/div) VOUT (1V/div) VOUT (2V/div) IL (20A/div) IL (20A/div) Time (10ms/div) Time (10ms/div) FIGURE 2-32: VIN Soft Turn-On. FIGURE 2-35: VIN (10V/div) VIN = 12V VOUT = 1.2V IOUT = 20A VSW (10V/div) VIN = 12V VOUT = 1.2V IOUT = 20A VEN (2V/div) VOUT (1V/div) VOUT (2V/div) IL (20A/div) IL (20A/div) Time (10ms/div) DS20005459B-page 12 Enable Turn-On/Turn-Off. y IN FIGURE 2-33: VIN = 12V VOUT = 1.2V IOUT = 20A VIN Soft Turn-Off. Time (4ms/div) FIGURE 2-36: Rise Time. Enable Turn-On Delay and 2015 Microchip Technology Inc. MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. VIN = 12V VOUT = 1.2V IOUT = 20A VEN (2V/div) VIN = 12V VOUT = 1.2V IOUT = Short VEN (2V/div) VOUT (500mV/div) VOUT (1V/div) IL (20A/div) IL (10A/div) Time (200μs/div) FIGURE 2-37: Fall Time. Time (4ms/div) Enable Turn-Off Delay and VIN = 12V VOUT = 1.2V IOUT = 20A VEN (1V/div) FIGURE 2-40: Enabled into Short. VIN = 12V VOUT = 1.2V IOUT = Short VIN (2V/div) VOUT (500mV/div) VOUT (1V/div) IL (10A/div) Time (4ms/div) Time (10ms/div) FIGURE 2-38: Enable Thresholds. FIGURE 2-41: Power-Up into Short-Circuit. p VOUT = 1.2V IOUT = 1A VIN = 12V VOUT = 1.2V VIN (2V/div) VOUT (500mV/div) VOUT (500mV/div) IOUT (10A/div) Time (20ms/div) Time (20ms/div) FIGURE 2-39: Rise Time. Enable Turn-On Delay and 2015 Microchip Technology Inc. FIGURE 2-42: Threshold. Output Peak Current-Limit DS20005459B-page 13 MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. VIN = 12V VOUT = 1.2V IOUT = 10A to Short VOUT (500mV/div) VIN = 12V VOUT = 1.2V IOUT = 2A to 12A VOUT (50mV/div) (AC-Coupled) IOUT (10A/div) IL (10A/div) Time (8ms/div) FIGURE 2-43: Time (100μs/div) Short-Circuit. p FIGURE 2-46: Transient Response. y OUT VIN = 12V ILDO = 1.2V VIN = Short to 10A VOUT (20mV/div) (AC-coupled) VIN = 12V VOUT = 1.2V IOUT = 0A VOUT (500mV/div) VSW (5V/div) IL (10A/div) IL (2A/div) Time (8ms/div) Time (8ms/div) FIGURE 2-44: Short-Circuit. p Output Recovery from FIGURE 2-47: MIC2125 Switching Waveform, IOUT = 0A. y OUT vin = 12V VOUT = 1.2V IOUT = 2.5A VOUT (500mV/div) VOUT (20mV/div) (AC-coupled) VIN = 12V VOUT = 1.2V IOUT = 0.1A VSW (5V/div) IL (2A/div) vsw (5V/div) Time (2ms/div) FIGURE 2-45: Output Recovery from Thermal Shutdown. DS20005459B-page 14 Time (4μs/div) FIGURE 2-48: MIC2125 Switching Waveform, IOUT = 0.1A. 2015 Microchip Technology Inc. MIC2125/6 Note: Unless otherwise noted, VIN = 12V, FREQ = 350 kHz. OUT OUT VOUT (20mV/div) (AC-coupled) VIN = 12V VOUT = 1.2V IOUT = 10A VIN = 12V VOUT = 1.2V IOUT = 0.1A IL (2V/div) VSW (10V/div) VSW (5V/div) VDH (10V/div) IL (10A/div) VDL (5V/div) Time (4μs/div) Time (2μs/div) FIGURE 2-49: 10A. Switching Waveform, IOUT = FIGURE 2-52: MIC2125 Switching Waveform, IOUT = 0.1A. IN OUT VIN = 12V VOUT = 1.2V IOUT = 0A VOUT (20mV/div) (AC-Coupled) VIN (5V/div) VOUT (1V/div) VSW (5V/div) VIN = 12V VOUT = 1.2V IOUT = 20A IL (10A/div) VPG (5V/div) Time (2μs/div) FIGURE 2-50: 20A. Time (4ms/div) Switching Waveform, IOUT = FIGURE 2-53: Turn-On. IN OUT VIN = 12V VOUT = 1.2V IOUT = 0A VIN = 12V VOUT = 1.2V IOUT = 0A IL (2A/div) VIN (5V/div) VSW (10V/div) VOUT (1V/div) VDH (10V/div) VDL (5V/div) Power Good at VIN Soft VPG (5V/div) Time (20ms/div) Time (4μs/div) FIGURE 2-51: MIC2125 Switching Waveform, IOUT = 0A. 2015 Microchip Technology Inc. FIGURE 2-54: Turn-Off. Power Good at VIN Soft DS20005459B-page 15 MIC2125/6 3.0 PIN DESCRIPTIONS The descriptions of the pins are listed in Table 3-1. TABLE 3-1: PIN FUNCTION TABLE Pin Number Symbol Description 1 VDD Internal Linear regulator output. Connect a 4.7 μF ceramic capacitor from VDD to AGND for decoupling. In the applications where VIN < +5.5V, VDD should be tied to VIN to by-pass the linear regulator. 2 PVDD 5V supply input for the low-side N-channel MOSFET driver, which can be tied to VDD externally. A 4.7 μF ceramic capacitor from PVDD to PGND is recommended for decoupling. 3 ILIM Current limit setting input. Connect a resistor from SW to ILIM to set the overcurrent threshold for the converter. Low-side gate driver output. The DL driving voltage swings from ground to VDD. 4 DL 5 PGND Power ground. PGND is the return path for the low side gate driver. Connect PGND pin to the source of low-side N-Channel external MOSFET. 6 FREQ Switching frequency adjust input. Connect FREQ to the mid-point of an external resistor divider from VIN to GND to program the switching frequency. Tie to VIN to operate at 750 kHz frequency. 7 DH High-side gate driver output. The DH driving voltage is floating on the switch node voltage (VSW). 8 SW Switch node and current-sense input. Connect the SW pin to the switch node of the buck converter. The SW pin also senses the current by monitoring the voltage across the low-side MOSFET during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to the SW pin using a Kelvin connection. 9 BST Bootstrap Capacitor Input. Connect a ceramic capacitor with a minimum value of 0.1 μF from BST to SW. 10 OVP Output Overvoltage Protection Input. Connect to the mid-point of an external resistive divider from the VOUT to GND to program overvoltage limit. Connect to AGND if the output overvoltage protection is not required. 11 NC 12 AGND 13 FB Feedback input. Input to the transconductance amplifier of the control loop. The FB pin is regulated to 0.6V. A resistor divider connecting the feedback to the output is used to set the desired output voltage. 14 PG Open-drain Power good output. Pull-up with an external pull-up resistor to VDD or to an external power rail. 15 EN Enable input. A logic signal to enable or disable the buck converter operation. Logic-high enables the device; logic-low shuts down the regulator. In disable mode, the VDD supply current for the device is minimized to 0.1 µA typically. Do not pull-up EN pin to VDD/PVDD. 16 VIN Supply voltage input. The VIN operating voltage range is from 4.5V to 28V. A 1 μF ceramic capacitor from VIN to AGND is required for decoupling. 17 EP Exposed Pad. Connect the exposed pad to the AGND copper plane to improve the thermal performance. DS20005459B-page 16 No connect. Analog Ground. Connect AGND to the exposed pad. 2015 Microchip Technology Inc. MIC2125/6 4.0 FUNCTIONAL DESCRIPTION The MIC2125 and MIC2126 are adaptive on-time synchronous buck controllers built for high input voltage to low output voltage applications. They are designed to operate over a wide input voltage range from 4.5V to 28V and their output is adjustable with an external resistive divider. An adaptive ON-time control scheme is employed to obtain a constant switching frequency and to simplify the control compensation. Overcurrent protection is implemented when sensing low-side MOSFET’s RDS(ON). The device features internal soft-start, enable, UVLO, and thermal shutdown. 4.1 Theory of Operation The MIC2125/6 Functional Block Diagram appears on page two. The output voltage is sensed by the MIC2125/6 feedback pin (FB), and is compared to a 0.6V reference voltage (VREF) at the low gain transconductance error amplifier (gm). Figure 4-1 shows the MIC2125/6 control loop timing during steady-state operation. When the feedback voltage decreases and the amplifier output is below 0.6V, the comparator triggers and generates an ON-time period. The ON-time period is predetermined by the fixed tON estimator circuitry value from Equation 4-1: EQUATION 4-2: t S – t OFF MIN 220ns D MAX = ----------------------------------- = 1 – --------------tS tS Where: tS 1/fSW It is not recommended to use MIC2125/6 with an OFF-time close to tOFF(MIN) during steady-state operation. The adaptive ON-time control scheme results in a constant switching frequency in the MIC2125/6. The actual ON-time and resulting switching frequency varies with the different rising and falling times of the external MOSFETs. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications. EQUATION 4-1: V OUT t ON ESTIMATED = ----------------------V IN f SW Where: VOUT Output Voltage VIN Power Stage Input Voltage fSW Switching Frequency At the end of the ON-time, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time depends upon the feedback voltage. When the feedback voltage decreases and the output of the gm amplifier is below 0.6V, the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(min), which is about 220 ns, the MIC2125/6 control logic applies the tOFF(min) instead. tOFF(min) is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET. The maximum duty cycle is obtained from the 220 ns tOFF(MIN): 2015 Microchip Technology Inc. FIGURE 4-1: Timing MIC2125/6 Control Loop Figure 4-2 shows the operation of the MIC2125/6 during load transient. The output voltage drops due to a sudden increase in load, which results in the VFB falling below VREF. This causes the comparator to trigger an ON-time period. At the end of the ON-time, a minimum OFF-time tOFF(min) is generated to charge CBST if the feedback voltage is still below VREF. The next ON-time is triggered immediately after the tOFF(min) due to the low feedback voltage. This operation results in higher switching frequency during load transients. The switching frequency returns to the nominal set frequency once the output stabilizes at new load current level. The output recovery time is fast and the output voltage deviation is small in MIC2125/6 converter due to the varying duty cycle and switching frequency. DS20005459B-page 17 MIC2125/6 IL CROSSES 0 AND VFB > 0.6. DISCONTINUOUS CONDUCTION MODE STARTS. IL VFB > 0.8. WAKE UP FROM DISCONTINUOUS CONDUCTION MODE. 0 VFB VREF ZC VHSD FIGURE 4-2: Response MIC2125/6 Load Transient Unlike true current-mode control, the MIC2125/6 uses the output voltage ripple to trigger an ON-time period. In order to meet the stability requirements, the MIC2125/6 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gm amplifier. The recommended feedback voltage ripple is 20 mV ~ 100 mV over the full input voltage range. If a low-ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the gm amplifier. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the ESR of the output capacitor is very low. For these applications, ripple injection is required to ensure proper operation. Refer to the Ripple Injection section under Application Information for details about the ripple injection technique. 4.2 Discontinuous Conduction Mode (MIC2125 Only) ESTIMATED ON TIME VLSD FIGURE 4-3: MIC2125 Control Loop Timing (Discontinuous Conduction Mode) The typical no load supply current during discontinuous conduction mode is only about 340 μA, allowing the MIC2125 to achieve high efficiency at light load operation. 4.3 Soft-Start Soft-start reduces the power supply inrush current at startup by controlling the output voltage rise time. The MIC2125/6 implements an internal digital soft-start by ramping up the reference voltage VREF from 0 to 100% in about 7 ms. Once the soft-start is completed, the related circuitry is disabled to reduce the current consumption. The MIC2125 operates in discontinuous conduction mode at light load. The MIC2125 has a zero crossing comparator (ZC detection) that monitors the inductor current by sensing the voltage drop across the low-side MOSFET during its ON-time. If the VFB > 0.6V and the inductor current goes slightly negative, the MIC2125 turns off both the high-side and low-side MOSFETs. During this period, the efficiency is optimized by shutting down all the non-essential circuits and the load current is supplied by the output capacitor. The control circuitry wakes up when the feedback voltage falls below VREF and triggers a tON pulse. Figure 4-3 shows the control loop timing in discontinuous conduction mode. DS20005459B-page 18 2015 Microchip Technology Inc. MIC2125/6 4.4 Current Limit The MIC2125/6 uses the low-side MOSFET RDS(ON) to sense the inductor current. Because MOSFET RDS(ON) varies from 30% to 40% with temperature, it is recommended to add a 50% margin to ICL in the previous equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect the SW pin directly to the drain of the low-side MOSFET to accurately sense the MOSFET’s RDS(ON). 4.5 Negative Current Limit (MIC2126 Only) The MIC2126 implements negative current limit by sensing the SW voltage when the low-side FET is off. If the SW node voltage exceeds 12 mV typical, the device turns off the low-side FET until the next ON-time event is triggered. The negative current limit value is given by Equation 4-4. EQUATION 4-4: FIGURE 4-4: Circuit MIC2125/6 Current-Limiting In each switching cycle of the MIC2125/6 converter, the inductor current is sensed by monitoring the voltage across the low-side MOSFET during the OFF period. An internal current source of 36 µA generates a voltage across the external resistor RCL. The ILIM pin voltage V(ILIM) is the sum of the voltage across the low side MOSFET and the voltage across the resistor (VCL). The sensed voltage V(ILIM) is compared with the power ground (PGND) after a blanking time of 150 ns. If the absolute value of the voltage drop across the low side MOSFET is greater than VCL, the current limit event is triggered. Eight consecutive current limit events triggers hiccup mode. The hiccup sequence, including the soft-start, reduces the stress on the switching FETs and protects the load and supply from severe short conditions. The current limit can be programmed by using Equation 4-3. EQUATION 4-3: I CLIM + PP 0.5 R DS ON – V OFFSET R CL = --------------------------------------------------------------------------------------------------------I CL Where: INLIM Negative Current Limit RDS(ON) On-Resistance of Low-Side Power MOSFET 4.6 Desired Current Limit ∆PP Inductor Current Peak-to-Peak RDS(ON) On-Resistance of Low-Side Power MOSFET VOFFSET Current-Limit Comparator Offset (Typical Value is –4 mV per Table 1-1) ICL Current-Limit Source Current (Typical Value is 36 µA, per Table 1-1) 2015 Microchip Technology Inc. MOSFET Gate Drive The MIC2125/6 high-side drive circuit is designed to switch an N-Channel MOSFET. Figure 4-1 shows a bootstrap circuit, consisting of a PMOS switch and CBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. If the bias current of the high-side driver is less than 10 mA, a 0.1 μF capacitor is sufficient to hold the gate voltage within minimal droop, (i.e., ∆BST = 10 mA × 3.33 μs/0.1 μF = 333 mV). A small resistor, RG in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. 4.7 Where: ICLIM 12mV I NLIM = -------------------R DS ON Overvoltage Protection The MIC2125/6 includes the OVP feature to protect the load from overshoots due to input transients and output short to a high voltage. When the overvoltage condition is triggered, the converter turns off immediately to allow the output voltage to discharge. The MIC2125/6 power should be recycled to enable it again. DS20005459B-page 19 MIC2125/6 5.0 APPLICATION INFORMATION 5.2 5.1 Setting the Switching Frequency Voltage rating, on-resistance, and total gate charge are important parameters for MOSFET selection. The MIC2125/6 are adjustable-frequency, synchronous buck controllers featuring a unique adaptive ON–time control architecture. The switching frequency can be adjusted between 200 kHz and 750 kHz by changing the resistor divider network consisting of R19 and R20. MIC2125/26 AGND The voltage rating for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VIN. A safety factor of 30% should be added to the VIN(MAX) while selecting the voltage rating of the MOSFETs to account for voltage spikes due to circuit parasitic elements. The power dissipated in the MOSFETs is the sum of conduction losses (PCONDUCTION) and switching losses (PAC). VDD/PVDD VDD 5V 4.7μF MOSFET Selection BST EQUATION 5-2: P SW = P CONDUCTION + P AC VIN VIN SW CS EQUATION 5-3: R19 2.2μF x3 FREQ R20 FB 2 P CONDUCTION = I SW RMS R DS ON PGND Where: FIGURE 5-1: Adjustment. Switching Frequency RDS(ON) On-Resistance of the MOSFET ISW(RMS) RMS current of the MOSFET Equation 5-1 gives the estimated switching frequency. The total high-side MOSFET switching loss is: EQUATION 5-1: R20 f SW ADJ = f O -------------------------R19 + R20 Where: fO Switching Frequency when R19 is 100 kΩ and R20 is open. fO is typically 750 kHz. For more precise setting, it is recommended to use Figure 5-2. EQUATION 5-4: P AC = 0.5 V IN I LOAD t R + t F f SW Where: tR/tF Switching Transition Times ILOAD Load Current fSW Switching Frequency Turn-on and turn-off approximated by: transition times can be EQUATION 5-5: Q SW HS R HSD PULL – UP + R HS GATE t R = ----------------------------------------------------------------------------------------------------------V DD – V TH FIGURE 5-2: R20 DS20005459B-page 20 Switching Frequency vs. 2015 Microchip Technology Inc. MIC2125/6 EQUATION 5-6: Q SW HS R HSD PULL – UP + R HS GATE t F = ----------------------------------------------------------------------------------------------------------V TH The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. EQUATION 5-9: Where: RHSD(PULL-UP) I L PK = I OUT MAX + 0.5 I L PP High-Side Gate Driver Pull-Up Resistance RHSD(PULL-DOWN) High-Side Gate Driver Pull-Down Resistance RHS(GATE) High-Side MOSFET Gate Resistance QSW(HS) Switching Gate Charge of the High-Side MOSFET VTH Gate Threshold Voltage The saturation current rating is given by: EQUATION 5-10: R CL I CL – V OFFSET I L SAT = --------------------------------------------------------R DS ON Where: RCL The high-side MOSFET switching losses increase with the switching frequency and the input voltage. The low-side MOSFET switching losses are negligible and can be ignored for these calculations. 5.3 Inductor Selection Inductance value, saturation, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Larger peak-to-peak ripple current increases the power dissipation in the inductor and MOSFETs. Larger output ripple current also requires more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple current requires a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss, and cost is to set the inductor ripple current to be equal to 40% of the maximum output current. The inductance value is calculated by Equation 5-7. EQUATION 5-7: V OUT V IN MAX – V OUT L = ------------------------------------------------------------------------------------V IN MAX f SW 0.4 I OUT MAX Where: fSW Switching Frequency 0.4 Ratio of AC Ripple Current to DC Output Current VIN(MAX) Maximum Power Stage Input Voltage The peak-to-peak inductor current ripple is: Current-Limit Resistor ICL Current-Limit Source Current VOFFSET Current-Limit Comparator Offset RDS(ON) On-Resistance of Low-Side Power MOSFET The RMS inductor current is used to calculate the I2R losses in the inductor. EQUATION 5-11: 2 I L RMS = I OUT MAX 2 I L PP + --------------------12 Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high-frequency operation of the MIC2125/6 requires the use of ferrite materials. Lower cost iron powder cores may be used, but the increase in core loss reduces the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized, although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be significant. Core loss information is usually available from the magnetics vendor. The amount of copper loss in the inductor is calculated by Equation 5-12: EQUATION 5-12: 2 EQUATION 5-8: P INDUCTOR CU = I L RMS R WINDING V OUT V IN MAX – V OUT I L PP = -------------------------------------------------------------------V IN MAX f SW L 2015 Microchip Technology Inc. DS20005459B-page 21 MIC2125/6 5.4 Output Capacitor Selection The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Voltage and RMS current capability are two other important factors for selecting the output capacitor. Recommended capacitor types are ceramic, tantalum, low-ESR aluminum electrolytic, OS-CON, and POSCAP. The output capacitor’s ESR is usually the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR is calculated by Equation 5-13. EQUATION 5-13: ESR C V OUT PP --------------------------OUT I L PP The power dissipated in the output capacitor is: EQUATION 5-16: 2 P DISS COUT = I COUT RMS ESR COUT 5.5 Input Capacitor Selection The input capacitor reduces peak current drawn from the power supply and reduces noise and voltage ripple on the input. The input voltage ripple depends on the input capacitance and ESR. The input capacitance and ESR values are calculated by using Equation 5-17 and Equation 5-18. EQUATION 5-17: Where: ∆VOUT(PP) Peak-to-Peak Output Voltage Ripple ∆IL(PP) Peak-to-Peak Inductor Current Ripple The required output capacitance is calculated in Equation 5-14. EQUATION 5-14: I L PP C OUT = --------------------------------------------------V OUT PP f SW 8 I OUT D 1 – D C IN = ---------------------------------------------- V IN C f SW Where: IOUT ƞ Power Conversion Efficiency ∆VIN(C) Input Ripple Due to Capacitance Value EQUATION 5-18: V IN ESR ESR CIN = ------------------------I L PK Where: COUT Output Capacitance Value fSW Switching Frequency As described in the Theory of Operation subsection of the Functional Description, the MIC2125/26 requires at least 20 mV peak-to-peak ripple at the FB pin to ensure that the gm amplifier and the comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Refer to the Ripple Injection subsection for details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 5-15. EQUATION 5-15: IC OUT RMS DS20005459B-page 22 Load Current Where: ∆VIN(ESR) Input Ripple Due to Capacitor ESR Value IL(PK) Peak Inductor Current The input capacitor should be qualified for ripple current rating and voltage rating. The RMS value of the input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: EQUATION 5-19: I CIN RMS I OUT MAX D 1 – D The power dissipated in the input capacitor is: EQUATION 5-20: 2 I L PP = -----------------12 P DISS CIN = I CIN RMS ESR CIN 2015 Microchip Technology Inc. MIC2125/6 5.6 Output Voltage Setting 5.7 The MIC2125/26 requires two resistors to set the output voltage, as shown in Figure 5-3. Output Overvoltage Limit Setting The output overvoltage limit should be typically 20% higher than the nominal output voltage. Set the OVP limit by connecting a resistor divider from the output to ground as shown in Figure 5-4. R1 FB gm AMP R1 OVP R2 R2 VREF VREF FIGURE 5-3: Configuration. Voltage-Divider The output voltage is determined by Equation 5-21: FIGURE 5-4: Configuration. OVP Voltage-Divider Choose R2 in the range of 10 kΩ to 49.9 kΩ and calculate R1 using Equation 5-23. EQUATION 5-21: R1 V OUT = V FB 1 + ------- R2 EQUATION 5-23: V OVP R1 = R2 ------------- – 1 0.6 Where: VFB 0.6V A typical value of R1 can be in the range of 3 kΩ and 15 kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using Equation 5-22. EQUATION 5-22: V FB R1 R2 = ----------------------------V OUT – V FB 5.8 Ripple Injection The VFB ripple required for proper operation of the MIC2125/6 gm amplifier and comparator is 20 mV to 100 mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For low output voltages, such as a 1V, the output voltage ripple is only 10 mV to 20 mV, and the feedback voltage ripple is less than 20 mV. If the feedback voltage ripple is so small that the gm amplifier and comparator cannot sense it, then the MIC2125/6 loses control and the output voltage is not regulated. In order to have sufficient VFB ripple, a ripple injection method should be applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: • Enough ripple at the feedback voltage due to the large ESR of the output capacitors (Figure 5-5). The converter is stable without any ripple injection. 2015 Microchip Technology Inc. DS20005459B-page 23 MIC2125/6 SW L MIC2125/26 FB FIGURE 5-5: R1 COUT R2 ESR R2 V FB PP = -------------------- ESR C I L PP OUT R1 + R2 Where: Peak-to-Peak Value of the Inductor Current Ripple • Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feed-forward capacitor, Cff in this situation, as shown in Figure 5-7. The typical Cff value is between 1 nF and 100 nF. MIC2125/26 FB L R1 R2 Cff R1 Cff R2 COUT ESR Invisible Ripple at FB. The process of sizing the ripple injection resistor and capacitors is as follows. EQUATION 5-24: SW MIC2125/26 FB RINJ FIGURE 5-7: Enough Ripple at FB. The feedback voltage ripple is: ∆IL(PP) L CINJ SW COUT ESR • Select CINJ as 100 nF, which can be considered as short for a wide range of the frequencies. • Select Cff to feed all output ripples into the feedback pin. Typical choice of Cff is 0.47 nF to 47 nF, if R1 and R2 are in the kΩ range. The Cff value can be calculated using Equation 5-26: EQUATION 5-26: t S V IN D 1 – D 1 C ff » ------ ------------------------------------------------------------------------------ V IN D 1 – D – V FB PP RP Where: VIN Power Stage Input Voltage D Duty Cycle tS 1/fSW RP (R1//R2//RINJ) ∆VFB(PP) Feedback Ripple • Select RINJ according to Equation 5-27. EQUATION 5-27: FIGURE 5-6: Inadequate Ripple at FB. With the feed-forward capacitor, the feedback voltage ripple is very close to the output voltage ripple. V IN D 1 – D 1 R INJ = ------- -------------------------------------------- C ff V FB PP f SW EQUATION 5-25: V FB PP ESR I L PP • Virtually no ripple at the FB pin voltage due to the very low ESR of the output capacitors. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor RINJ and a capacitor CINJ, as shown in Figure 5-7. DS20005459B-page 24 2015 Microchip Technology Inc. MIC2125/6 6.0 PCB LAYOUT GUIDELINES PCB layout is critical to achieve reliable, stable and efficient performance. The following guidelines should be followed to ensure proper operation of the MIC2125/26 converter. 6.1 IC • The ceramic bypass capacitors which are connected to the VDD and PVDD pins must be located right at the IC. Use wide traces to connect to the VDD, PVDD and AGND, PGND pins respectively. • The signal ground pin (AGND) must be connected directly to the ground planes. • Place the IC close to the point-of-load (POL). • Signal and power grounds should be kept separate and connected at only one location. 6.2 Input Capacitor • Place the input ceramic capacitors as close as possible to the MOSFETs. • Place several vias to the ground plane close to the input capacitor ground terminal. 6.3 Inductor • Keep the inductor connection to the switch node (SW) short. • Do not route any digital lines underneath or close to the inductor. • Keep the switch node (SW) away from the feedback (FB) pin. • The SW pin should be connected directly to the drain of the low-side MOSFET to accurately sense the voltage across the low-side MOSFET. 6.4 Output Capacitor • Use a copper plane to connect the output capacitor ground terminal to the input capacitor ground terminal. • The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. 6.5 MOSFETs • MOSFET gate drive traces must be short. The ground plane should be the connection between the MOSFET source and PGND. • Choose a low-side MOSFET with a high CGS/CGD ratio and a low internal gate resistance to minimize the effect of dv/dt inducted turn-on. • Use a 4.5V VGS rated MOSFET. Its higher gate threshold voltage is more immune to glitches than a 2.5V or 3.3V rated MOSFET. For more information about the Evaluation board layout, please contact Microchip sales. 2015 Microchip Technology Inc. DS20005459B-page 25 MIC2125/6 7.0 PACKAGING INFORMATION 16-Lead QFN 3 mm x 3 mm Package Outline and Recommended Land Pattern Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging DS20005459B-page 26 2015 Microchip Technology Inc. MIC2125/6 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging 2015 Microchip Technology Inc. DS20005459B-page 27 MIC2125/6 DS20005459B-page 28 2015 Microchip Technology Inc. MIC2125/6 APPENDIX A: REVISION HISTORY Revision A (November 2015) • Original Conversion of this Document. Revision B (December 2015) • Corrected the erroneous listing of the MIC2126 example with a 64LD package. Replaced with correct 16LD package information. 2015 Microchip Technology Inc. DS20005459B-page 29 MIC2125/6 NOTES: DS20005459B-page 30 2015 Microchip Technology Inc. MIC2125/6 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office. PART NO. XX X Examples: a) MIC2125YML: Device Temperature Package Device: MIC2125: 28V, Synchronous Buck Controller featuring Adaptive On-Time Control with HyperLight Load 28V, Synchronous Buck Controller featuring Adaptive On-Time Control with Hyper Speed Control MIC2126: Temperature: Y = Package: ML = –40°C to +125°C 16-Pin 3 mm x 3 mm QFN 2015 Microchip Technology Inc. 28V, Synchronous Buck Controller featuring Adaptive On-Time Control with HyperLight Load, –40°C to +125°C junction temperature range, 16LD QFN b) MIC2126YML: 28V, Synchronous Buck Controller featuring Adaptive On-Time Control with Hyper Speed Control, –40°C to +125°C junction temperature range, 16LD QFN DS2005459B-page 31 MIC2125/6 NOTES: DS2005459B-page 32 2015 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights unless otherwise stated. Trademarks The Microchip name and logo, the Microchip logo, dsPIC, FlashFlex, flexPWR, JukeBlox, KEELOQ, KEELOQ logo, Kleer, LANCheck, MediaLB, MOST, MOST logo, MPLAB, OptoLyzer, PIC, PICSTART, PIC32 logo, RightTouch, SpyNIC, SST, SST Logo, SuperFlash and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. The Embedded Control Solutions Company and mTouch are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, BodyCom, chipKIT, chipKIT logo, CodeGuard, dsPICDEM, dsPICDEM.net, ECAN, In-Circuit Serial Programming, ICSP, Inter-Chip Connectivity, KleerNet, KleerNet logo, MiWi, motorBench, MPASM, MPF, MPLAB Certified logo, MPLIB, MPLINK, MultiTRAK, NetDetach, Omniscient Code Generation, PICDEM, PICDEM.net, PICkit, PICtail, RightTouch logo, REAL ICE, SQI, Serial Quad I/O, Total Endurance, TSHARC, USBCheck, VariSense, ViewSpan, WiperLock, Wireless DNA, and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. Silicon Storage Technology is a registered trademark of Microchip Technology Inc. in other countries. GestIC is a registered trademark of Microchip Technology Germany II GmbH & Co. KG, a subsidiary of Microchip Technology Inc., in other countries. All other trademarks mentioned herein are property of their respective companies. © 2015, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. ISBN: 978-1-5224-0039-4 QUALITYMANAGEMENTSYSTEM CERTIFIEDBYDNV == ISO/TS16949== 2015 Microchip Technology Inc. Microchip received ISO/TS-16949:2009 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. 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