INTERSIL ISL6232CAZA-T

ISL6232
®
Data Sheet
April 18, 2005
High Efficiency System Power Supply
Controller for Notebook Computers
The ISL6232 is a high efficiency, quad output controller
optimized for converting battery, wall adapter or network DC
input voltage into system supply voltages required for
portable applications. The ISL6232 includes two PWM
controllers generating 0.8V to 5.5V, or fixed 5V and 3.3V
outputs. It also features 5V and 3.3V always linear regulators
with up to 100mA output current.
ISL6232 uses constant frequency current mode PWM
control with out of phase operation for reducing the input
ripple current and the ESR requirement of the input
capacitors. Over 95% efficiency is achieved through
synchronous rectification and dual PWM/Skip mode
architecture. High light load efficiency with skip mode
extends the battery life in system standby or shutdown
mode. The 5V and 3.3V always linear regulators take their
inputs from battery or ac adapter; and, to further improve
efficiency, their outputs are switched to the 5V or 3.3V
outputs from switching regulators when 5V or 3.3V is
available. Ultrasonic pulse skipping mode maintains
switching frequency above 25kHz to eliminate the audio
noise for high light load efficiency, and fixed frequency PWM
operation mode reduces the RF interference in sensitive
applications. External loop compensation is used to optimize
the transient response with optimized external components.
An accurate current sensing resistor in series with an output
inductor, or DC resistance of the inductor is used to sense
the output current of the current ramp signal, and
overcurrent protection. A peak current detecting scheme is
used for overcurrent protection and to prevent the inductor
from saturation.
The ISL6232 has internal soft-start to control the inrush
current. The soft-stop feature avoids negative output voltage
for undervoltage protection, overcurrent protection, and
shutdown by discharging output through an internal switch,
and by damping the inductor current. The ISL6232 also
features overvoltage protection, power-up sequences, power
good output, and thermal shutdown. It has quiescent power
dissipation as low as 3.5mW.
FN9116.0
Features
• Supply Voltage Range: 5.5V to 25V
• 3.3V and 5V Fixed or Adjustable Outputs from 0.8V to
5.5V
• 5V, 3.3V/100mA Always Linear Regulators
• Out of Phase Operation Reduces the ESR Requirement of
the Input Capacitors
• ±1.5% Output Voltage Accuracy over Temperature
• Fixed 300kHz Current Mode Control Architecture
• Accurate Current Sensing or DCR Current Sensing
• Internal Soft-Start and Soft-Stop Output Discharge
• Selectable Power-up Sequence
• Selectable Forced PWM, Pulse Skipping, and Ultrasonic
Pulse Skipping Mode (25kHz min)
• Peak Overcurrent Limit Prevents Inductor Saturation
• Overvoltage Protection, Undervoltage Shutdown
• Power Good Output
• Thermal shutdown
• 5µA Shutdown Current
• Integrated Bootstrap Schottky Diodes
• 3.5mW Quiescent Power Dissipation
• Pb-Free Available (RoHS Compliant)
Applications
• Notebook, Sub-notebook, and Tablet Computers
• 2-4 cell Li-Ion Battery-Powered Devices
• Dual Output Supplies for DSP, Memory, Logic and
Microprocessor
• Telecom Systems, Network servers, and Storage
Ordering Information
PART
NUMBER*
TEMP.
RANGE (°C)
ISL6232CAZA
(Note 1)
-10° to 100°
PACKAGE
28 Ld QSOP
(Pb-free)
PKG.
DWG. #
M28.15
NOTE: Intersil Pb-free products employ special Pb-free material
sets; molding compounds/die attach materials and 100% matte tin
plate termination finish, which are RoHS compliant and compatible
with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
*Add “-T” for Tape and Reel.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-352-6832 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2005. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6232
Pinout
ISL6232 (QSOP)
TOP VIEW
2
OUT3
1
28
BOOT3
CS3
2
27
PHASE3
EN3
3
26
UGATE3
PGOOD
4
25
LGATE3
COMP3
5
24
LDO3
FB3
6
23
PGND
SHDN#
7
22
VIN
SKIP#
8
21
LDO5
REF
9
20
LGATE5
GND
10
19
VCC
FB5
11
18
UGATE5
COMP5
12
17
PHASE5
EN5
13
16
BOOT5
CS5
14
15
OUT5
FN9116.0
April 18, 2005
ISL6232
Absolute Maximum Ratings
Thermal Information
SHDN#, VIN to GND . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
BOOT3, BOOT5 to GND . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
VCC, LDO3, LDO5, CS3, CS5, OUT3, OUT5, COMP3, COMP5,
FB3, FB5, SKIP#, FREQ, PGOOD, EN3, EN5, REF
to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
BOOT3 to PHASE3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
BOOT5 to PHASE5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
UGATE3 to PHASE3 . . . . . . . . . . . . . . . . . . -0.3V to (BOOT3+0.3V)
UGATE5 to PHASE5 . . . . . . . . . . . . . . . . . . -0.3V to (BOOT5+0.3V)
LGATE3 to GND . . . . . . . . . . . . . . . . . . . . . . -0.3V to (LDO5+0.3V)
LGATE5 to PGND . . . . . . . . . . . . . . . . . . . . . -0.3V to (LDO5+0.3V)
PHASE3 to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-1V to 28V
PHASE5 to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -1V to 28V
PGND to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 1
Thermal Resistance
θJA (°C/W)
QSOP Package (Note 1) . . . . . . . . . . . . . . . . . . . . .
77
Junction Temperature Range. . . . . . . . . . . . . . . . . .-55°C to +150°C
Operating Temperature Range . . . . . . . . . . . . . . . -10°C to +100°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
VIN = 12V, SHDN# = EN3 = EN5 = VCC, SKIP# = FB3 = FB5 = 0V, ILDO5 = 0mA, ILDO3 = 0mA, CREF = 0.22µF,
CLDO3 = CLDO5 = 4.7µF, TA = -10°C to +100°C, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
25
V
SMPS CONTROLLER
VIN Voltage Range
(Note 3)
3.3V Fixed Output Voltage
VIN = 5.5V to 25V, FB3 = 0V, SKIP# = VCC
3.250
3.300
3.350
V
5.0V Fixed Output Voltage
VIN = 5.5V to 25V, FB5 = 0V, SKIP# = VCC
5.023
5.100
5.177
V
FB3/FB5 at Programmable Mode
VIN = 5.5V to 25V, SKIP# = VCC
0.788
0.800
0.812
V
0.1
0.2
0.3
V
5.5
V
FB3/FB5 Programmable Mode Threshold
Voltage
OUT3/OUT5 Voltage Range at Programmable
Mode
(Note 3)
Line Regulation
VIN = 5.5V to 25V @ OUT3
VIN = 5.5V to 25V @ OUT5
5.5
0.8
0.005
%/V
SKIP# = VCC, IOUT = 0A to 5A
-0.1
SKIP# = 0V, IOUT = 0A to 5A
-0.5
SKIP# = REF, IOUT = 0A to 5A
-0.5
OUT3/OUT5 Input Leakage Current
LDO5 = OUT = 5.5V, EN3 = EN5 = 0V
0.1
1
µA
CS3/CS5 Input Leakage Current
LDO5 = CS = 5.5V, EN3 = EN5 = 0V
0.1
1
µA
FB3/FB5 Input Bias Current
FB = 0.75V
0.01
0.1
µA
COMP3/COMP5 Trans-Conductance
COMP = 2.5V
50
100
150
µS
Positive Current Limit Threshold
CS-OUT
64
80
96
mV
Pulse Skipping Current Threshold
CS-OUT
3
13
26
mV
Zero Crossing Current Threshold
CS-OUT
6
mV
Negative Current Limit Threshold
CS-OUT
-20
mV
Operating Frequency
VIN = 5.5V to 25V
Load Regulation
Maximum Duty Cycle
255
300
%
345
94
OUT3/OUT5 Soft-Start Period
EN = VCC
3
1.0
kHz
%
1.2
1.4
ms
FN9116.0
April 18, 2005
ISL6232
Electrical Specifications
VIN = 12V, SHDN# = EN3 = EN5 = VCC, SKIP# = FB3 = FB5 = 0V, ILDO5 = 0mA, ILDO3 = 0mA, CREF = 0.22µF,
CLDO3 = CLDO5 = 4.7µF, TA = -10°C to +100°C, unless otherwise noted. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
UGATE/LGATE Gate Driver Sink Current
UGATE and LGATE are forced to 2.5V
2
A
UGATE/LGATE Gate Driver Source Current
UGATE and LGATE are forced to 2.5V
1
A
UGATE Gate Driver Pull-Up Resistance
Force BOOT-PHASE to 5V
2
4
Ω
UGATE Gate Driver Pull-Down Resistance
Force BOOT-PHASE to 5V
1
2.5
Ω
LGATE Gate Driver Pull-Up Resistance
LDO5 = 5V
2
4
Ω
LGATE Gate Driver Pull-Down Resistance
LDO5 = 5V
1
2.5
Ω
20
40
Ω
0.3
0.4
V
5
10
µA
150
200
µA
1
7
µA
5.0
5.1
V
OUT3/OUT5 On-Resistance at Discharge Mode
OUT3/OUT5 Low-Side Switch Turn-On
Threshold at Discharge Mode
0.2
LINEAR REGULATOR AND REFERENCE
VIN Shutdown Current
VIN = 5.5V to 25V, SHDN# = 0V
VIN Standby Current
VIN = 5.5V to 25V, EN3 = EN5 = 0
VIN Operating Supply Current
LDO5 switched to OUT5, 5V SMPS enabled and
LDO3 switched to OUT3, 3.3V SMPS enabled.
LDO5 Output Voltage
VIN = 5.5V to 25V, EN3 = EN5 = 0, ILDO5 = 0 to 100mA
4.9
LDO5 Maximum Output Current
VIN = 5.5V to 25V, EN3 = EN5 = 0
100
LDO5 Current Limit
LDO5 pulled to GND
170
300
mA
LDO5 Undervoltage Lockout Threshold
Rising Edge
4.3
4.5
V
Falling Edge
LDO5 Switch-Over Threshold
mA
4.0
4.2
V
4.63
4.78
4.93
V
2
3
Ω
3.28
3.345
V
LDO5 Switch-Over Resistance
OUT5 to LDO5
LDO3 Output Voltage
VIN = 5.5V to 25V, EN3 = EN5 = 0, ILDO3 = 0 to 100mA
LDO3 Maximum Output Current
VIN = 5.5V to 25V, EN3 = EN5 = 0
LDO3 Current Limit
LDO3 pulled to GND
170
300
mA
LDO3 Switch-Over Threshold
Rising Edge
3.00
3.10
V
Falling Edge
3.215
100
2.85
mA
2.95
V
LDO3 Switch-Over Resistance
OUT3 to LDO3
2.5
3.8
Ω
Quiescent Power Consumption
VIN = 5.5V to 25V, FB3 = FB5 = SKIP# = 0V, Both
SMPSs are enabled
3.5
5
mW
REF Output Voltage
No Load
2.00
2.03
V
REF Load Regulation
0µA<IREF<100µA
25
mV
117
%
1
µs
1.97
FAULT DETECTION
Output Overvoltage Trip Threshold
OUT is above the target voltage at no load
Rising
Output Overvoltage Fault Propagation Delay
FB = 1.0V
Output Undervoltage Trip Threshold
OUT is below the target voltage at no load
70
75
78
%
Output Undervoltage Latch Blanking Time
FB = 0.5V
15
20
25
ms
PGOOD Trip Threshold
Rising (After soft-start cycle complete)
91
97
%
Falling
109
83
113
88
%
PGOOD Propagation Delay
FB = 0.8V
10
20
µs
PGOOD Low Level Voltage
ISINK = 5mA
0.1
0.2
V
4
FN9116.0
April 18, 2005
ISL6232
Electrical Specifications
VIN = 12V, SHDN# = EN3 = EN5 = VCC, SKIP# = FB3 = FB5 = 0V, ILDO5 = 0mA, ILDO3 = 0mA, CREF = 0.22µF,
CLDO3 = CLDO5 = 4.7µF, TA = -10°C to +100°C, unless otherwise noted. (Continued)
PARAMETER
TEST CONDITIONS
PGOOD Leakage Current
Forced to VCC
MIN
TYP
-1
Thermal Shutdown Threshold Hysteresis
MAX
UNITS
1
µA
150
25
°C
INPUTS
EN3/EN5 Input Voltage
Low
Delay start threshold voltage
High
SKIP# Input Voltage
0.8
REF0.05
REF
High
SHDN# Input Voltage
0.8
REF0.05
REF
REF+
0.2
V
2.4
Low
Input Leakage Current
V
2.4
Low
Ultrasonic skip mode threshold voltage
REF+
0.2
0.7
High
2.4
EN3/EN5/SKIP#/SHDN#
-1
1
V
µA
NOTES:
2. Specifications to -10°C are guaranteed by design and not production tested.
3. Guaranteed by design.
5
FN9116.0
April 18, 2005
ISL6232
Circuit in Figure 29, no load on LDO5, LDO3, OUT3 OUT5, and REF, VIN = 12V,
EN3 = EN5 = VCC, SHDN# = VIN, TA = 25°C, unless otherwise noted.
100
100
90
90
80 7V-SKIP
80
EFFICIENCY (%)
EFFICIENCY (%)
Typical Operating Performance
70
60
12V-SKIP
25V-SKIP
50
40
7V-PWM
12V-PWM
25V-PWM
30
60
25V-ULTRA SKIP
50
40
30
20
10
10
0.010
0.100
1.000
3.3V OUTPUT LOAD (A)
0
0.001
10.000
FIGURE 1. EFFICIENCY OF 3.3V OUTOUT vs LOAD
(7V, 12V, 25V-PWM, SKIP)
350
350
300
300
250
200
7V-PWM
150
100
7V-ULTRA SKIP
7V-SKIP
0.100
1.000
3.3V OUTPUT LOAD (A)
10.000
250
200
12V-PWM
150
100
12V-ULTRA SKIP
12V-SKIP
50
50
0
0.001
0.010
0.100
1.000
10.000
3.3V OUTPUT LOAD (A)
0
0.001
100.000
FIGURE 3. FREQUENCY OF 3.3V OUTPUT vs LOAD
(7V-PWM, SKIP, ULTRA SKIP)
0.010
0.100
1.000
3.3V OUTPUT LOAD (A)
10.000
FIGURE 4. FREQUENCY OF 3.3V OUTPUT vs LOAD
(12V-PWM, SKIP, ULTRA SKIP)
350
3.34
300
3.32
OUTPUT VOLTAGE (V)
12V-ULTRA SKIP
250
25V-PWM
200
150
100
0.010
FIGURE 2. EFFICIENCY OF 3.3V OUTPUT vs LOAD
(7V, 12V, 25V-ULTRA SKIP)
FREQUENCY (kHz)
FREQUENCY (kHz)
7V-ULTRA SKIP
12V-ULTRA SKIP
20
0
0.001
FREQUENCY (kHz)
70
25V-ULTRA SKIP
25V-SKIP
3.28
12V-PWM
12V-SKIP
3.26
3.24
3.22
50
0
0.001
3.30
0.010
0.100
1.000
3.3V OUTPUT LOAD (A)
FIGURE 5. FREQUENCY OF 3.3V OUTPUT vs LOAD
(25V-PWM, SKIP, ULTRA SKIP)
6
10.000
3.20
0.001
0.010
0.100
1.000
3.3V OUTPUT LOAD (A)
10.000
FIGURE 6. OUTPUT VOLTAGE REGULATION OF 3.3V vs
LOAD (12V-PWM, SKIP, ULTRA SKIP)
FN9116.0
April 18, 2005
ISL6232
Typical Operating Performance
Circuit in Figure 29, no load on LDO5, LDO3, OUT3 OUT5, and REF, VIN = 12V,
EN3 = EN5 = VCC, SHDN# = VIN, TA = 25°C, unless otherwise noted. (Continued)
100
100
90
12V-ULTRA SKIP
80
70
80
EFFICIENCY (%)
OUTPUT RIPPLE (mV)
90
12V-SKIP
60
50
40
30
20
50
7V-PWM
12V-PWM
25V-PWM
40
30
20
12V-PWM
10
10
0
0.001
0.010
0.100
1.000
3.3V OUTPUT LOAD (A)
0
0.001
10.000
FIGURE 7. OUTPUT RIPPLE OF 3.3V vs LOAD
(12V-PWM, SKIP, ULTRA SKIP)
0.010
0.100
1.000
5V OUTPUT LOAD (A)
10.000
FIGURE 8. EFFICIENCY OF 5V OUTPUT vs LOAD
(7V, 12V, 25V-PWM, SKIP)
100
350
90
300
80
7V-Ultra SKIP
12V-Ultra SKIP
25V-Ultra SKIP
70
60
FREQUENCY (kHz)
EFFICIENCY (%)
7V-SKIP
12V-SKIP
70 25V-SKIP
60
50
40
30
250
7V-PWM
200
150
100
7V-ULTRA SKIP
7V-SKIP
20
50
10
0
0.001
0.010
0.100
1.000
5V OUTPUT LOAD (A)
FIGURE 9. EFFICIENCY OF 5V OUTPUT vs LOAD
(7V, 12V, 25V-UPTRA SKIP)
350
350
300
300
250
12V-PWM
200
150
100
12V-ULTRA SKIP
12V-SKIP
50
0
0.001
0.010
0.100
1.000
5V OUTPUT LOAD (A)
10.000
FIGURE 10. FREQUENCY OF 5V OUTPUT vs LOAD
(7V-PWM, SKIP, ULTRA SKIP)
FREQUENCY (kHz)
FREQUENCY (kHz)
0
0.001
10.000
250
25V-PWM
200
150
100
25V-ULTRA SKIP
25V-SKIP
50
0.010
0.100
1.000
5V OUTPUT LOAD (A)
FIGURE 11. FREQUENCY OF 5V OUTPUT vs LOAD
(12V-PWM, SKIP, ULTRA SKIP)
7
10.000
0
0.001
0.010
0.100
1.000
5V OUTPUT LOAD (A)
10.000
FIGURE 12. FREQUENCY OF 5V OUTPUT vs LOAD
(25V-PWM, SKIP, ULTRA SKIP)
FN9116.0
April 18, 2005
ISL6232
Typical Operating Performance
Circuit in Figure 29, no load on LDO5, LDO3, OUT3 OUT5, and REF, VIN = 12V,
EN3 = EN5 = VCC, SHDN# = VIN, TA = 25°C, unless otherwise noted. (Continued)
60
12V-PWM
5.15
OUTPUT RIPPLE (mV)
OUTPUT VOLTAGE (V)
5.20
5.10
12V-SKIP
12V-ULTRA SKIP
5.05
12V-ULTRA SKIP
50
12V-SKIP
40
30
20
12V-PWM
10
5.00
0.001
0.010
0.100
1.000
0
0.001
10.000
0.010
5V OUTPUT LOAD (A)
50
700
45
600
40
35
30
25
6
8
10
12
14
16
18
20
22
24
400
300
200
100
26
6
8
10
12
14
16
18
20
22
24
26
INPUT VOLTAGE (V)
FIGURE 15. PWM NO-LOAD BATTERY CURRENT vs INPUT
VOLTAGE
FIGURE 16. IDLE NO-LOAD BATTERY CURRENT vs INPUT
VOLTAGE
160
5
158
4.5
BATTERY CURRENT (µA)
BATTERY CURRENT (µA)
10.000
500
INPUT VOLTAGE (V)
156
154
152
150
148
146
144
142
140
1.000
FIGURE 14. OUTPUT RIPPLE OF 5V vs LOAD
(12V-PWM, SKIP, ULTRA SKIP)
BATTERY CURRENT (µA)
BATTERY CURRENT (mA)
FIGURE 13. OUTPUT VOLTAGE REGULATION OF 5V vs LOAD
(12V-PWM, SKIP, ULTRA SKIP)
20
0.100
5V OUTPUT LOAD (A)
4
3.5
3
2.5
2
1.5
1
0.5
6
8
10
12
14
16
18
20
22
24
26
INPUT VOLTAGE (V)
FIGURE 17. STANDBY INPUT CURRENT vs INPUT VOLTAGE
8
0
6
8
10
12
14
16
18
20
22
24
26
INPUT VOLTAGE (V)
FIGURE 18. SHUTDOWN INPUT CURRENT vs INPUT
VOLTAGE
FN9116.0
April 18, 2005
ISL6232
Typical Operating Performance
Circuit in Figure 29, no load on LDO5, LDO3, OUT3 OUT5, and REF, VIN = 12V,
EN3 = EN5 = VCC, SHDN# = VIN, TA = 25°C, unless otherwise noted. (Continued)
5.2
LDO5 OUTPUT VOLTAGE (V)
LDO3 OUTPUT VOLTAGE (V)
3.5
3.4
EN3=0
3.3
3.2
3.1
EN3=VCC
3
2.9
0
10
20
30
40
50
60
70
80
90
100
5.15
5.1
EN5=VCC
5.05
5
4.95
EN5=0
4.9
4.85
4.8
LDO3 OUTPUT CURRENT (mA)
0
10
20
30
40
50
60
70
80
90
100
LDO3 OUTPUT CURRENT (mA)
FIGURE 19. LDO3 REGULATION OUTPUT VOLTAGE vs
OUTPUT CURRENT
FIGURE 20. LDO5 REGULATION OUTPUT VOLTAGE vs
OUTPUT CURRENT
2
VREF (V)
1.99
1.98
VIN
(5V/div)
1.97
LDO3
(1V/div)
1.96
LDO5
(2V/div)
REF
1.95
-10
0
10
20
30
40 50 60
IREF (µA)
70
80
90
100
FIGURE 21. REFERENCE VOLTAGE vs OUTPUT CURRENT
(1V/div)
FIGURE 22. REF, LDO3, AND LDO5 POWER UP
EN
EN5
(2V/div)
(2V/div)
VOUT3
VOUT3
(2V/div)
(2V/div)
VOUT5
VOUT5
(2V/div)
(2V/div)
FIGURE 23. DELAYED START WAVEFORMS (EN5=REF)
9
FIGURE 24. DELAYED START WAVEFORMS (EN3=REF)
FN9116.0
April 18, 2005
ISL6232
Typical Operating Performance
Circuit in Figure 29, no load on LDO5, LDO3, OUT3 OUT5, and REF, VIN = 12V,
EN3 = EN5 = VCC, SHDN# = VIN, TA = 25°C, unless otherwise noted. (Continued)
VOUT5
(100mV/div)
IL3
(2A/div)
IL5
(5A/div)
IL3
(2A/div)
VOUT3
LGATE3
(2V/div)
(5V/div)
VOUT5
(5V/div)
FIGURE 25. SOFT-START WAVEFORMS
FIGURE 26. 5V PWM-MODE LOAD TRANSIENT RESPONSE
VOUT3
(100mV/div)
IL3
(2A/div)
LGATE3
(5V/div)
FIGURE 27. 3.3V PWM-MODE LOAD TRANSIENT RESPONSE
Functional Pin Descriptions
BOOT3
It powers the upper MOSFET driver for OUT3. Connect a
0.1µF ceramic capacitor to PHASE3.
BOOT5
It powers the upper MOSFET driver for OUT5. Connect a
0.1µF ceramic capacitor to PHASE5.
UGATE3
High side N-MOSFET gate drive output for OUT3. Swing
between PHASE3 and BOOT3.
UGATE5
High side N-MOSFET gate drive output for OUT5. Swing
between PHASE5 and BOOT5.
LGATE3, LGATE5
Low-side N-MOSFET gate drive outputs for OUT3 and
OUT5, respectively. Swing between 0V and LDO5.
10
LDO3
3.3V internal LDO output. It can provide a total of 100mA. If
OUT3 is greater than the LDO3 switch-over threshold, the
LDO3 regulator shuts down and LDO3 pin connects to OUT3
through a 2.5Ω switch. Bypass a 4.7µF ceramic capacitor to
ground.
FB3, FB5
Output feedback inputs for OUT3 and OUT5. Connect to
ground for fixed 3.3V and 5V outputs. Connect to output
through a voltage divider for adjustable outputs.
CS3, CS5
Inductor current sensing positive inputs for OUT3 and OUT5.
The current sensing signal is compared with the 80mV
internal set threshold to perform overcurrent protection. It
has negative 20mV current limit for ultrasonic skipping mode
operation. It is also used as current ramp for current mode
control.
FN9116.0
April 18, 2005
ISL6232
PGOOD
PHASE3, PHASE5
Open drain output. Active high after soft-start cycle delay
when both outputs are above 90% of the regulated voltage;
Pull low immediately when either output is below 90% of the
regulated output.
Phase connection pins for OUT3 and OUT5, respectively.
Connect to joint points of the high side MOSFET source,
output inductor, and low side MOSFET drain.
GND
Connect to the 5V output. It is used to sense the output
voltage and connect to the negative terminal of the sensing
resistor. If OUT5 is greater than the LDO5 switch-over
threshold, the LDO5 internal linear regulator shuts down and
LDO5 connects to OUT5 through a 2Ω switch.
Analog ground.
VIN
This pin is the input of the internal 5V and 3.3V LDO
regulators. Connect VIN to the battery or AC adapter output.
LDO5
5V internal LDO output. LDO5 is the gate driver supply for
the external MOSFETs. It can provide a total of 100mA,
including MOSFET gate drive requirements and external
loads. If OUT5 is greater than the LDO5 switch-over
threshold, the LDO5 regulator shuts down and LDO5 pin
connects to OUT5 through a 2Ω switch. Bypass a 4.7µF
ceramic capacitor to ground.
OUT5
OUT3
Connect to the 3.3V output. It is used to sense the output
voltage and connected to the negative terminal of the
sensing resistor. If OUT3 is greater than the LDO3 switchover threshold, the LDO3 internal linear regulator shuts
down and LDO3 connects to OUT3 through a 2.5Ω switch.
PGND
Power ground.
VCC
SHDN#
VCC is derived from LDO5. This pin is used to power the
internal analog integrated circuit only. The only connection to
this pin is a 0.1µF ceramic capacitor to ground.
Shut down control input. Connect to ground, for shuting
down all internal circuitry. Connect to VIN for automatic start
up.
EN3
SKIP#
3.3V output enable input. Connect to high for enabling 3.3V
output. Connect to low for disabling 3.3V output. When it is
connected to REF, the 3.3V output starts after 5V output
reaches regulation. Drive EN3 below the clear fault level to
reset the fault latches.
Mode selection input. Connect to ground for pulse skip
operation. Connect to VCC for forced PWM operation.
Connect to REF for ultrasonic pulse skipping operation. For
debugging purposes, SKIP# can be pulled 1V above VCC to
disable the latch-off features of overcurrent, undervoltage,
and overvoltage protections.
EN5
5V output enable input. Connect to high for enabling 5V
output. Connect to low for disabling to 5V output. When it is
connected to REF, the 5V output starts after 3.3V output
reaches regulation. Drive EN5 below the clear fault level to
reset the fault latches.
11
REF
Reference output. Bypass a 0.22µF ceramic capacitor to
ground. REF can source up to 100uA for external loads.
COMP3, COMP5
External loop compensation for OUT3 and OUT5,
respectively. Connect a resistor in series with a capacitor to
ground.
FN9116.0
April 18, 2005
ISL6232
Typical Operating Performance
Typical Application Circuits
The typical application circuits shown in Figure 28 and
Figure 29 generate 5V/5A and 3.3V/5A for system power
supplies in a notebook computer.
VIN: 5.5V to 25V
C13
0.1µF
5V ALWAYS ON
VCC
VIN
LDO5
C6
4.7µF
4.7
ISL6232
C10
10µF
OUT3
3.3V/5A
C11
220µF
220
12mW
4.0V
Q3
IRF7807V
R2: 8mΩ,1%
,1% L2: 4.7µF
C9
0.1µF
Q4
IRF7811AV
BOOT3
BOOT5
UGATE3
UGATE5
PHASE3
PHASE5
LGATE3
LGATE5
CS3
OUT3
GND
C3
R3
270pF 390kΩ
FB3
COMP3
SKIP#
Q2
IRF7811AV
OUT5
5V/5A
C2
180µF
12mW
6.3V
OUT5
FB5
R5
C5
390kΩ 270pF
VCC
COMP5
REF
R4
R
100kΩ
100k
C7
0.22µF
0.22
EN3
SHDN#
OFF
R1: 8mΩ,1%
CS5
EN5
ON
C4
0.1µF
0.1
Q1
IRF7807V
L1: 6.8 µH
PGND
PGOOD
VCC
C1
10µF
10
LDO3
VCC
3.3V ALWAYS ON
C12
4.7µF
4.7
FIGURE 28. ISL6232 TYPICAL APPLICATION CIRCUIT WITH ACCURATE CURRENT SENSING
12
FN9116.0
April 18, 2005
ISL6232
VIN: 5.5V to 22V
C13
0.1µF
5V ALWAYS ON
VIN
VCC
LDO5
C14
1µF
1
ISL6232
C10
10µF
10
BOOT3
Q3
IRF7807V
OUT3
3.3V/5A
C11
220
220µF
12mΩ
Ω
4.0V
UGATE3
C9
0.1µF
Rdc218mΩ
L2:4.7µH
L2: 4.7ÿH
Rdc2:12mΩ
C6
4.7
4.7µF
C1
10µF
10
BOOT5
Q1
IRF7807V
UGATE5
PHASE3
PHASE5
LGATE3
LGATE5
C4
0.1µF
0.
L1:6.8µH Rdc1:12mΩ
L1:6.8ÿH
Rdc1:22mΩ
C15: 0.47µF R9: 1.0kΩ
R7:1.5kΩ
R7:
Q4
IRF7811AV
C8: 0.47µF
Q2
IRF7811AV
PGND
R4: 2.0kΩ
CS3
OUT5
5V/5A
C2
180µF
12m Ω
6.3V
R8: 3kΩ
CS5
OUT3
OUT5
GND
FB5
C3
R3
270pF 300kΩ
Ω
FB3
R5
C5
390kΩ
Ω 270pF
VCC
COMP5
COMP3
R6
100kΩ
REF
SKIP#
PGOOD
REF
EN3
EN5
ON
SHDN#
OFF
LDO3
C7
0.22
0.22µF
VCC
3.3V ALWAYS ON
C12
4.7µF
4.7
FIGURE 29. ISL6232 TYPICAL APPLICATION CIRCUIT WITH DCR CURRENT SENSING
Theory of Operation
The ISL6232 is a high-efficiency quad output controller
optimized for converting battery, wall adapter, or network DC
input voltage into system supply voltages required for
portable applications where high efficiency and low
quiescent supply current are required. The ISL6232
includes two PWM controllers that are fixed at 5V and 3.3V
respectively, or they can be programmed from 0.8V to 5.5V.
Figure 30 shows its functional block diagram. ISL6232 uses
a constant-frequency, 300kHz, peak current-mode PWM
control scheme with 180o out-of-phase operation for
reducing the input ripple current and also ESR requirement
of the input capacitors. Light-load efficiency is improved by
the variable-frequency pulse-skipping operation that reduces
switching losses and gate-charge losses. In order to
eliminate the audio noise at extremely light load condition,
the ultrasonic pulse skipping mode is selectable by tying
13
SKIP# pin to REF so that a minimum 25kHz switching
frequency can be maintained.
Each switching-mode step-down circuit includes two
external N-MOSFETs and an LC output filter. The output
voltage is the average AC voltage at PHASE node, which is
regulated by changing the duty cycle of the external NMOSFETs. The gate-drive signal to the high side MOSFET
must exceed VIN voltage and is provided by a 0.1µF boost
capacitor, which is connected between BOOT and PHASE.
FN9116.0
April 18, 2005
ISL6232
VCC
LDO5
BOOT5
BOOT3
UGATE5
UGATE3
PHASE5
PHASE3
LGATE5
5V SYNCH
COMP5
PWM BUCK
PWM BUCK
CONTROLLER
CONTROLLER
COMP3
CLK5
CLK3
CS3
ENA5
ENA3
CS5
FB5
LGATE3
3.3V SYNCH
DECODER
DECODER
PGOOD5
6pF
OUT5
PGOOD3
6pF
4.5V
OUT3
3.0V
+
+-
+
-
-
FB3
LDO5
LDO3
5V LDO
3.3V LDO
VIN
PGOOD
OSC
SHDN#
SHUTDOWN
REF
REF
STANDBY
EN3
POWER-UP
EN5
SQUENCE
FIGURE 30. FUNCTIONAL DIAGRAM
Each buck controller includes a feedback resistor divider
network, a multiplexer for programmable mode, a transconductance error amplifier, a PWM comparator, high-side
gate and low-side gate drivers, and control logic circuit.
Figure 31 shows the synchronous buck PWM controller
block diagram.
The external loop compensator is used to optimize the
transient response with optimized external components. An
accurate current sensing resistor in series with output
inductor or the DC resistance of the inductor is used to
sense the output current for current ramp signal and
overcurrent protection. Moreover, it contains fault-protection
14
circuitry that can monitor the undervoltage and overvoltage
conditions of the OUT3 and OUT5 buck controller output
voltages. A power-on sequence is implemented to control
the power-up timing of OUT3 and OUT5. The power good
signal, PGOOD, is toggled to logic high once both OUT3 and
OUT5 reach 90% of the regulation points and soft start
period is finished.
The ISL6232 also includes 5V and 3.3V linear regulators, 2V
reference, and automatic switch-over circuits. All the blocks
inside ISL6232 are not directly powered by VIN voltage.
Instead, the VIN voltage is stepped down to 5V by the 5V
LDO5 regulator to supply both internal circuitry and the gate
FN9116.0
April 18, 2005
ISL6232
COMP
SKIP#
BOOT
+
+
0.8V
+
COMP
EAMP
UGATE
CLK
Soft
Start
Slope
Comp
PWM/SKIP
LOGIC
CONTROLLER
PHASE
LDO5
LGATE
+
DC
OFFSET
OV
UV
FB
++
CSA
x16
ÿ20
ÿ
+
CS
OUT
0.2V
+
OCP
1.0V
OUT
0.2V
+
SKIP
+
PGOOD
0.72V
0.06V
+
Zero
Cross
+
OV
0.9V
+
+
20ms
Blanking
NOCP
+
UV
0.1V
NCSA
x8
0.6V
FIGURE 31. SYNCHRONOUS BUCK PWM CONTROLLER BLOCK DIAGRAM
drivers. The low side drivers are directly powered from LDO5
and the high side drivers are indirectly powered from LDO5
through the internal Schottky diode and external bootstrap
capacitor. Only after soft-start is finished and when OUT5 is
above 4.75V, an automatic switch-over circuit turns off the
internal LDO5 regulator and powers the device from OUT5.
This prevents the LDO5 and LDO3 from a voltage dip during
the switch-over. It switches back to LDO5 when OUT5 is
disabled for EN5 = 0. Similary, only after soft-start is finished
and when OUT3 is above 3.0V, it turns off the 3.3V LDO3
regulator and powers the device from OUT3. It switches
back to LDO3 just before OUT3 is disabled.
ISL6232 has internal soft-start to control the inrush current.
This soft-stop feature avoids negative output voltage for
undervoltage protection and overcurrent protection so that
the part can be shut down by first discharging OUT3 or
OUT5 through an internal 20Ω switch and damping the
15
inductor current. Finally, thermal shutdown is included in
ISL6232 to protect the part from over-heating.
PWM Controller
The two-buck controllers are nearly identical. The only
difference is the fixed output voltage, 3.3V versus 5V. Both
buck controllers use a peak current-mode PWM control
scheme. For peak current mode control, the system can be
unstable when the duty cycle is higher than 0.5. A slope
compensation signal is used to stabilize the system. A PWM
comparator compares the integrated voltage feedback signal
(COMP) with the sum of the amplified current-sense signal
and the slope-compensation ramp. At each rising edge of
the internal clock, the high side MOSFET turns on until the
PWM comparator trips. During this on-time, current ramps
up through the inductor, sourcing current to the output and
storing energy in the inductor. The current-mode feedback
system regulates the peak inductor current as a function of
FN9116.0
April 18, 2005
ISL6232
the output voltage error signal. To preserve loop stability, a
slope-compensation ramp is summed into the main PWM
comparator. During the off cycle, the external high side
MOSFET turns off and the external low side MOSFET turns
on. The inductor releases the stored energy as its current
ramps down while still providing current to the output. The
output capacitor stores the charge when the inductor current
exceeds the load current and releases the charge when the
inductor current is lower, smoothing the voltage across the
load. During an overcurrent or short-circuit condition, it
immediately turns off the high side MOSFET and turns on
the low side MOSFET. This peak current limit prevents the
inductor from saturation. If the overcurrent still exists at the
rising edge of the next clock, the high side MOSFET will stay
off and the low side MOSFET remains on to let the inductor
current ramp down.
When SKIP# = GND, the efficiency is automatically
optimized throughout the entire load current range. Skip
mode significantly improves light-load efficiency by reducing
the effective frequency, which reduces switching losses. The
automatic transition to skip mode is determined by the
current’s zero-cross comparator, which detects inductor
current zero crossing and turns off the low side MOSFET.
The boundary is set by the following equation:
V
(1 – D)
OUT
I OUT = -----------------------------------2Lf s
The PWM controller keeps the peak inductor current about
15% of the overcurrent limit in an active cycle, thus allowing
subsequent cycles to be skipped as long as the COMP pin
voltage is low enough. The switching waveform at light load
behaves noisy and is asynchronous due to pulse skipping.
Skip mode transits smoothly to fixed-frequency PWM
operation as load current increases.
When SKIP# = REF, the ultrasonic mode is enabled so that
the minimum switching frequency can be maintained higher
than 25kHz. This ultrasonic pulse-skipping mode eliminates
the audio noise that can occur in skip mode at very light load
condition. Ultrasonic pulse skipping occurs if no switching
has taken place within the last 30µs. The low side MOSFET
turns on to induce a negative inductor current. Then, the
high side MOSFET turns on when the inductor current
reaches the negative current limit, or when the PWM
comparator output has toggled to high before the next clock
cycle. The negative current limit is determined by the
following equation:
NLIM
TABLE 1. OPERATION MODE TABLE
SKIP#
MODE
LOAD
CONDITION
GND
Skip
Light
GND
PWM
Heavy
REF
Ultrasonic
Skip
Light
REF
PWM
Heavy
Constant frequency PWM
VCC
PWM
Light
Constant frequency PWM
VCC
PWM
Heavy
Constant frequency PWM
(EQ. 1)
where D = duty cycle, fs = switching frequency, L = inductor
value, IOUT = output loading current, VOUT = output voltage.
I
When SKIP# = VCC, the controller always operates in forced
PWM mode for the lowest noise and zero-cross detection is
bypassed. The inductor current becomes negative at light
load condition because the PWM loop tries to maintain a
duty cycle set by VOUT/VIN, leading to poor efficiency at light
loads. During forced PWM operation, each clock rising edge
sets the main PWM latch that turns on the high side switch
for a period determined by the duty cycle. As the high side
MOSFET turns off, the synchronous rectifier latch sets and
the low side MOSFET turns on. The low side MOSFET stays
on until the beginning of the next clock cycle. Table 1 shows
the operation mode.
V NLIM
= ------------------R CS
(EQ. 2)
where VNLIM is the negative current limit threshold and RCS
is current sense resistance.
16
DESCRIPTION
Pulse skipping, DCM. Turn
off UGATE when the
inductor current reaches the
skip current threshold.
Constant frequency PWM
Pulse skipping, DCM. Turn
on LGATE if there is no
switching after 30us. Turn it
off once it reaches negative
current limit or PWM
comparator's output has
toggled to high before the
next clock cycle.
UGATE and LGATE Drivers
A 0.1µF capacitor connected between BOOT and PHASE,
as well as the internal Schottky diode connected from LDO5
to BOOT, generate the gate drive for the high side MOSFET.
When the low side MOSFET turns on, PHASE goes to
PGND. LDO5 charges the bootstrap capacitor through the
Schottky diode. When the low side MOSFET turns off and
the high side MOSFET turns on, PHASE voltage goes to
VIN. The Schottky diode prevents the capacitor from
discharging into LDO5. The LGATE synchronous rectifier
drivers are powered by LDO5.
Both UGATE and LGATE gate drivers sink 2A peak current
out of gate terminal, ensuring adequate gate drive for highcurrent applications. The internal pull-down transistors that
drive LGATE low have a 1Ω typical on-resistance. These low
on-resistance pull-down transistors can prevent LGATE from
being pulled up during the fast rise time of the PHASE nodes
due to capacitive coupling from the drain to the gate of the
low side MOSFETs. In the case of high-current applications,
some combinations of both high side and low side MOSFETs
can still cause sufficient gate-drain coupling, which leads to
shoot-through currents and poor efficiency. To get around
FN9116.0
April 18, 2005
ISL6232
this situation, a small resistor (a few ohms) in series with the
BOOT pin can be added to increase the turn-on time of the
high side MOSFETs at the cost of efficiency.
Dead-time control circuitry is also implemented to monitor
the UGATE and LGATE voltages so that one of the external
MOSFETs can be prevented from turning on before the other
one completely turns off. This method can allow operation
without shoot-through with a wide selection range of external
MOSFETs, minimizing delays and maintaining efficiency. To
achieve this, the trace from UGATE and LGATE to the
MOSFET gates must be low resistance and low inductance.
Otherwise, the control circuitry will regard the MOSFET gate
as in the off-state when there is still some charge left on the
gate.
CURRENT SENSE INPUTS, CS AND OUT
An internal current-sense amplifier produces a current signal
proportional to the voltage generated by the sense
resistance and the inductor current (RCS*IL). The amplified
current-sense signal and the internal slope-compensation
signal sum together at the comparator inverting input. The
PWM comparator turns off the high side MOSFET when this
summed voltage exceeds the COMP voltage of the error
amplifier.
The ISL6232 has a positive current limit threshold of 80mV
with a ±20% tolerance. Whenever the voltage difference
between CS and OUT exceeds 80mV, the high side
MOSFET turns off and the low side MOSFET turns on. This
lowers the duty cycle and causes the output voltage to drop
until the current limit is no longer exceeded.
The external low-value sense resistor, RCS, should be
picked for 65mV/IPEAK, where IPEAK is the required peak
inductor current to support the full load current. Also, the
other components must be chosen to sustain continuous
current of 95mV/RCS. It is useful to wire the current-sense
inputs with a twisted pair, which can reduce the possible
noise picked up at CS and OUT as well as avoid unstable
switching.
A negative current limit threshold, typical of 20mV, is
implemented to prevent excessive reverse inductor currents
when OUT dumps charges. This negative current limit is
used to determine when the low side MOSFET should turn
off at ultrasonic pulse skipping mode.
Mode Transition Between DCM and CCM
The automatic transition to skip mode is determined by the
current zero-cross comparator, which detects the inductor
current's zero crossing and turns off the low side MOSFET.
The threshold between pulse skipping pulse frequency
modulation (PFM) and non-skipping PWM can not
completely coincide with the boundary between continuous
current mode (CCM) and discontinuous current mode
17
(DCM). In CCM mode, the boundary is set by the following
equation,
V OUT ( 1 – D )
I OUT = ---------------------------------2Lf s
(EQ. 3)
where D = duty cycle, fs = switching frequency, L = inductor
value, IOUT = output loading current, VOUT = output
voltage.
However, the boundary is set by the following formula in
DCM condition.
V SKIP
I OUT = ---------------2R CS
(EQ. 4)
where VSKIP is the current limit threshold at skip mode. The
above two boundary values can not be completely matched
due to the tolerance of the pulse skipping current limit
threshold, inductance, frequency, and line input voltage. The
ISL6232 is designed in such a way that it operates in a
mixed mode between DCM mode CCM mode during the
mode transition, which may have one longer pulse and is
followed by one shorter pulse. But this does not affect the
output ripple voltage. This is a normal operation and it is not
the loop stability issue. The inductor current is regulated in
the CCM mode to meet the load current requirement since
the inductor current is fixed in the DCM mode during the
mixed mode operation.
POWER GOOD (PGOOD)
PGOOD is kept low during soft-start. When both OUT3 and
OUT5 voltages reach 90% of the regulation points, PGOOD
toggles to high after the end of soft-start period. When either
output turns off or is 10% below its regulation point , or a
fault occurs in either output, PGOOD goes low. PGOOD is
set to low during shutdown, standby, and soft-start.
DISCHARGE MODE
When the output is disabled by toggling EN3 or EN5 from
high to low or latched off due to the undervoltage or
overcurrent fault, it is discharged through an internal 20Ω
switch from PHASE to PGND until the output drops to 0.3V.
After the output drops below 0.3V, LGATE is forced to high to
discharge the output to ground. LDO5, VCC, and REF are
active at this mode.
POWER-ON RESET, DIGITAL SOFT-START, AND UVLO
When VIN rises above approximately 3.8V, power-on reset
occurs. After internal reference voltages and bias currents
are ready, both LDO3 and LDO5 are enabled. After LDO5
reaches undervoltage lockout (UVLO) voltage, 4.3V, the
buck controller is enabled if either EN3 or EN5 is tied to
VCC. Then, the internal digital soft-start circuitry begins to
charge-up the output capacitor of the buck controller
gradually in 44 steps within 1.2ms (typ), so that the VIN inrush current can be reduced. Each buck controller includes
FN9116.0
April 18, 2005
ISL6232
its own internal digital soft-start circuit. In shutdown or
standby mode, the soft-start output is reset to zero.
Fault Protection
Undervoltage Protection
When the output undervoltage is detected at below 75%
(typ) of the regulation output for 20ms blanking time, it enters
the discharge mode by discharging the output through the
internal 20Ω switch connected from PHASE to PGND. When
the output voltage drops below 0.3V, the external low side
MOSFET is latched on to discharge the output to ground.
When either output is in UVP, both outputs are latched off
through soft-discharge. The latches can be reset by toggling
VIN, SHDN#, or EN.
Overvoltage Protection
When either output voltage is above 113% (typ) of the
regulation point, both outputs are latched off by turning on
the low side MOSFET and turning off the high side MOSFET.
Discharging the output capacitors through the inductor and
low-side MOSFET causes negative output voltage. For loads
that cannot tolerate a negative voltage, place a 1A power
Schottky diode across the output to act as a reverse-polarity
clamp. If the overvoltage is due to a short in the high side
MOSFET, the battery fuse will be blown and isolated from
the output.
Thermal Protection
Thermal-overload protection limits total power dissipation in
the device. When the junction temperature exceeds 150°C,
a thermal sensor forces most of the internal circuitry into
shutdown mode, thus allowing the device to cool down. The
thermal sensor turns the device on again after the junction
temperature drops by 25°C, causing a pulsed output during
continuous overload conditions. The digital soft-start
sequence begins after the thermal shutdown condition is
removed.
Power-Up Sequence
EN3 and EN5 control the power-up sequencing of buck
controllers. Setting EN above 2.4V enables the outputs, and
setting EN below 0.8V disables the outputs. Connecting EN3
or EN5 to REF forces the respective output off until the other
output reaches 90% of the regulation point and soft-start
cycle has ended. One of the buck controllers can remain on
even though the other buck controller turns off. Table 2
shows the power sequence selection.
TABLE 2. POWER-UP SEQUENCE TABLE
EN3
EN5
LDO3
Low
x
x
OFF
OFF
OFF
OFF
High
Low
Low
ON
ON
OFF
OFF
Overcurrent Protection
High
Low
High
ON
ON
ON
OFF
The output current is continuously monitored through either
an accurate sensing resistor or the DCR of the inductor.
When the inductor peak current reaches the overcurrent limit
threshold, it immediately turn off the high side MOSFET and
turn on the low side MOSFET. This peak current limiting
prevents inductor saturation. If the overcurrent or short
circuit condition is detected for more than 20mS (typical), the
high side MOSFET is latched off and the output is
discharged through the internal 20Ω switch connected from
PHASE to PGND. When the output voltage drops below
0.3V, the low side MOSFET is latched on to discharge the
output to ground. When either output is latched off due to
overcurrent, the other output is also latched off through softdischarge.
High
High
Low
ON
ON
OFF
ON
High
High
High
ON
ON
ON
ON
High
High
REF
ON
ON
ON after
3.3V up
ON
High
REF
High
ON
ON
ON
ON after
5V up
18
LDO5 5V BUCK
3.3V
BUCK
SHDN#
SHUTDOWN MODE
When SHDN# is set below 0.8V, the part is completely shut
down with a 5µA (typ) shutdown VIN current. When SHDN#
is set above 2.4V, both LDO outputs and REF are active.
This is prerequisite for enabling buck controllers. For
automatic shutdown and startup, SHDN# can be tied to VIN.
Table 3 is the summary of various operation modes.
FN9116.0
April 18, 2005
ISL6232
TABLE 3. SUMMARY FOR VARIOUS OPERATION MODES
MODE
Shutdown
CONDITION
COMMENT
SHDN# = Low.
All circuitry off.
Standby
SHDN# = High.
EN3 = EN5 = Low.
LDO5, LDO3, and 2V
reference active. LGATE
stays high.
Soft-Start
LDO5>UVLO EN3
or/and EN5 enabled.
Output voltage ramps up
in 1.2ms.
Normal
Operation
SHDN# = High. EN3 and All circuitry is running.
EN5 enabled.
Discharge
Either output is still high Discharging the output
in standby mode.
through an internal 20Ω
switch from PHASE to
PGND. One output may
still operate while the
other is in discharge
mode. LDO5 active.
Undervoltage Either output is below
Protection 75% of nominal after a
20ms blanking time and
output enabled.
Overvoltage Either output voltage is
Protection 13% higher than the
nominal.
Lower side MOSFET is
latched on after discharge
mode terminates. LDO5 is
active. Reset by toggling
EN3, EN5, SHDN#, VIN
POR.
Low side MOSFET is
forced high and high side
MOSFET is forced low.
LINEAR REGULATORS AND 2V REFERENCE
In ISL6232, there are two internal regulators available, which
are LDO5 (5V) and LDO3 (3.3V). Once LDO5 is higher than
4.3V, it provides power for buck controllers, 2V reference,
and all the other blocks powered by VCC. The maximum
guaranteed output current that both LDO5 and LDO3
regulators can supply is 100mA. The real maximum current
drawn from the LDOs is determined by the maximum power
dissipation allowed in the package. A short-circuit or
overcurrent limit protection, 170mA (typ), is implemented for
both LDO5 and LDO3. Bypass LDO5 and LDO3 with a 4.7µF
ceramic capacitor.
When OUT5 is larger than the LDO5 switch-over threshold
(4.78V) and after soft-start is finished, LDO5 is shorted to
OUT5 through an internal 2Ω switch and the LDO5 regulator
is disabled to reduce the power dissipation. Similarly, when
OUT3 is larger than the LDO3 switch-over threshold (3.0V)
and after soft-start is finished, LDO3 is shorted to OUT3
through an internal 2.5Ω switch and LDO3 is turned off. All
the internal blocks (powered by VCC) get the power from the
high-efficiency switching power supply instead of the linear
regulator.
The reference voltage, REF, is 2V with a ±1.5% accuracy.
REF provides the reference voltage, 0.8V, for buck
controllers. REF is bypassed to GND with a 0.22µF
capacitor.
19
Application Information
This section describes how to select the external
components including the inductor, input and output
capacitors, switching MOSFETs, current sensing resistors
and loop compensator design.
The inductor selection has to accommodate trade-offs
between cost, size and efficiency. For example, the lower the
inductance, the smaller the inductor size, but ripple current is
higher; this results in higher ac losses in the magnetic core
and the windings, which decrease the system efficiency. On
the other hand, the higher inductance results in lower ripple
current and smaller output filter capacitors, but higher DCR
(dc resistance of the inductor) loss and slower transient
response. Practical inductor design is based on the inductor
ripple current being ±(15-20)% of the maximum operating dc
current at maximum input voltage. The required inductance
can be calculated from:
V IN – VOUT VOUT
L = ---------------------------------- ----------------VIN f s
∆I L
(EQ. 5)
where VIN is input voltage, VOUT is the output voltage, ∆IL is
the inductor ripple current and fs is the switching frequency.
The practical inductor ripple current is chosen at 30% of the
output current: ∆I L = 30% ⋅ I OUT
For VIN = 12V, VOUT = 5V, IOUT = 5A, and fs = 300kHz,
12 – 5
5
L = ----------------- -------------------------------------3- = 6.5µH
0.3 × 5 12 × 300 × 10
(EQ. 6)
Ferrite core inductors are often the best choice since they
are optimized at 300kHz to 600kHz operation with low core
loss. The inductor must be large enough not to saturate at
the overcurrent limit IOC
95 mV
I OC = ----------------R
CS
(EQ. 7)
One important factor is that the smaller the inductance, the
faster the transient response. One of the parameters limiting
the converters response to load transient is the time required
to change the inductor current. Given a sufficiently fast
control loop design, the ISL6232 can provide either
approximately 5% or 95% duty cycle in response to a load
transient. The response time is the time required to slew the
inductor current from an initial current value to the transient
current level. During this interval the difference between the
inductor current and the transient current level must be
supplied by the output capacitor. Minimizing the response
time can minimize the output capacitance required. The
response time to a transient is different for the application of
load and the removal of load.
FN9116.0
April 18, 2005
ISL6232
The following equations give the approximate response time
interval for application and removal of a step transient load:
LI
LI
STEP
STEP
T
≈ ----------------------------------, T
≈ ---------------------rise V – V
fall V
IN
OUT
OUT
(EQ. 8)
Where ISTEP is the transient load current step, Trise and Tfall
are the response time to the application and the removal of
load, respectively. The worst-case response time can be
either at the application or removal of load. Be sure to check
both of these equations at the minimum and maximum
output levels for the worst-case response time.
Determining the Overcurrent Limit
The minimum current-limit threshold must be great enough
to support the maximum load current when the current limit
is at the minimum tolerance value. ISL6232 uses peak
current detection. The peak inductor current occurs at
IOUT,MAX plus half of the ripple current; therefore,
∆I L
I LIMIT > I OUTMAX + -------2
(EQ. 9)
The minimum current-limit threshold voltage is 65mV. For
accurate current sense-resistor with 8mΩ, the current limit
ILIMIT is 8.1A, which is higher than 5.75A, calculated from
the above equation. So, the circuit can easily deliver fullrated 5A using 65mV current limit threshold.
For DCR of inductor current sensing (Refer to Figure 29), if
the voltage drop across the DCR of the inductor is higher
than 65mV, then a resistor divider across the inductor has to
be used so that the output voltage across the capacitor
reaches current limit threshold (65mV minimum) at the
maximum DCR. The inductor time constant has to match
with the RC current sensing network for good current
sensing accuracy, that is,
L
R R8
17
-------------≤ ---------------------- C
R dc1
R 7 + R 8 16
(EQ. 10)
This requirement is not so stringent because it is used for
overcurrent protection and not for the adaptive output
voltage positioning applications. Besides, DCR of the
inductor is also a function of the temperature. A good
general rule for copper is to allow 3.9% additional resistance
for each 10°C of temperature rise. Since there is 1MΩ input
impedance from CS to ground, to achieve good current
sensing accuracy, R7, and R8 have to meet the following
inequality:
V
OUT ----------------------------≤ 2mV
RX
1 + ------------------R // R
8
7
(EQ. 11)
Where Rx is the input impedance from CS to ground.
Given Rdc1 = 15mΩ at 85°C, L = 6.8µH, we choose
R7 = 1.5kΩ, R8 = 3kΩ, and C16 = 0.47µF.
20
Check the current limit ILIMIT as the following equation:
R8
--------------------R
×I
= 65mV
R 7 + R 8 dc1 LIMIT
(EQ. 12)
We have ILIMIT = 6.5A. Therefore, the circuit can easily
deliver the fully rated 5A current.
Output Capacitor Selection
The output filter capacitor must have low enough equivalent
series resistance (ESR) to meet output ripple and loadtransient requirements. The ISL6232 uses peak current
mode control, which does not require high enough ESR to
satisfy stability requirements. The output capacitance must
also be high enough to absorb the inductor energy while
transitioning from full-load to no-load conditions without
tripping the overvoltage fault latch. In applications where the
output is subject to large load transients, the output capacitor
size depends on how much ESR is needed to prevent the
output from dipping too low under a load transient, ignoring
the sag due to finite capacitance.
The ESR of the output capacitors has to meet the following
equation:
V DIP
ESR < ---------------I STEP
(EQ. 13)
where VDIP is the maximum tolerable transient voltage drop
or rise. In system power applications, the ESR of the output
capacitors usually determines the steady-state output
voltage ripple, which is practically designed below 1% of the
output voltage. Thus, we have
V pp
ESR ≤ ---------∆I L
(EQ. 14)
where Vpp is the peak-to-peak output voltage ripple. The
actual capacitance value required relates to the physical size
needed to achieve low ESR, as well as to the chemistry of
the capacitor technology and loop bandwidth.
Since the voltage dip or spike due to loop transient response
is usually smaller than that of voltage dip or spike due to
ESR during the load step transient, the capacitor is usually
selected by ESR and voltage rating rather than by
capacitance value. The commonly used output capacitors
are POSCAP from Sanyo and SPCAP from Panasonic due
to smaller size, low ESR and reasonable price.
Most power supplies requires an overall voltage accuracy of
±5%, including steady-state tolerance, steady-state output
ripple, line regulation and step load transient tolerance. The
ISL6232 has ±1.5% accuracy for the band gap, ±0.5% for
steady-state output ripple and line regulation. This allows
±3% tolerance due to the step load transient. For 5V output,
the required ESR is given by
3% × 5V
ESR ≤ ----------------------- = 50mΩ
3A
(Assume 3A step load)
(EQ. 15)
FN9116.0
April 18, 2005
ISL6232
Input Capacitor Selection
The input capacitors must meet the input ripple current
(IRMS) requirement imposed by the switching current. The
ISL6232 dual switching regulators operate at the same
switching frequency with out of phase. This interleaves the
current pulses drawn by the two regulators and have no
overlap time at normal operation. The input RMS current is
much smaller when compared with both regulators operating
in phase or operating at different switching frequencies. The
input RMS current varies with load and the input voltage.
The maximum input capacitor RMS current for a single buck
regulator is given by:
V OUT ( V IN – V OUT )
I rms = I OUT --------------------------------------------------------V
(EQ. 16)
IN
when VIN = 2VOUT (D = 50%), Irms has maximum current of
IOUT/2. The ESR of the input capacitor is important for
determining capacitor power dissipation. All the power (I2rms
x ESR) heats up the capacitor and reduces efficiency. Nontantalum chemistries (ceramic, polymer such as POSCAP, or
SPCAP) are preferred due to their low ESR and resilience to
power-up surge currents. Choose input capacitors that
exhibit less than +10°C temperature rise at the RMS input
current for optimal circuit longevity.
MOSFET Selection
The synchronous buck regulator has the input voltage from
either AC adapter output or battery output. The maximum
AC adapter output voltage does not exceed 24V while the
maximum battery voltage does not exceed 17V for a 4 series
Li-Ion battery cell battery pack. Therefore, a 30V logic
MOSFET should be used.
The high side MOSFET must be able to dissipate the
conduction losses plus the switching losses. The input
voltage of the synchronous regulator is equal to the AC
adapter output voltage or battery voltage. The maximum
efficiency is achieved by selecting a high side MOSFET that
has the conduction losses equal to the switching losses.
Ensure that the ISL6232 LGATE gate driver can supply
sufficient gate current to prevent it from conduction,
otherwise, cross-conduction problems may occur.
Conduction is due to the injected current into the drain-togate parasitic capacitor (Miller capacitor Cgd) caused by the
voltage rising rate at phase node during the moment of the
high-side MOSFET turn-on. Reasonably slowing turn-on
speed of the high-side MOSFET by connecting a resistor
between the BOOT pin and gate drive supply source, and
high sink current capability of the low-side MOSFET gate
driver, helps reduce the possibility of cross-conduction.
For the high-side MOSFET, the worst-case conduction
losses occur at the minimum input voltage:
V OUT 2
P Q1, Conduction = ---------------- I OUT R DSON
V IN
21
(EQ. 17)
The optimum efficiency occurs when the switching losses
equal the conduction losses. However, it is difficult to
calculate the switching losses in the high-side MOSFET
since it must allow for difficult-to-quantify factors that
influence the turn-on and turn-off times. These factors
include the MOSFET internal gate resistance, gate charge,
threshold voltage, stray inductance, and the pull-up and pulldown resistance of the gate driver. The following switching
loss calculation provides a rough estimate.
(EQ. 18)
Q gd
Q gd
1
1
P Q1, Switching = --- V IN I LV f s ------------------------- + --- V IN I LP f s ----------------- + Q rr V IN f s
I g, source 2
I g, sin k
2
where Qgd: drain-to-gate charge, Qrr: total reverse recovery
charge of the body-diode in low side MOSFET, ILV: inductor
valley current, ILP:is Inductor peak current, Ig,sink and
Ig,source are the peak gate-drive source/sink current of Q1.
To achieve low switching losses requires low drain-to-gate
charge Qgd. Generally, the lower the drain-to-gate charge,
the higher the on-resistance. Therefore, there is a trade-off
between the on-resistance and drain-to-gate charge. Good
MOSFET selection is based on the Figure of Merit (FOM),
which is the product of the total gate charge and onresistance. Usually, the smaller the value of FOM, the higher
the efficiency for the same application.
For the low-side MOSFET, the worst-case power dissipation
occurs at minimum output voltage and maximum input
voltage:
V OUT 2

P Q2 =  1 – --------------- I
R
V IN  OUT DSON

(EQ. 19)
Choose a low-side MOSFET that has the lowest possible
on-resistance with a moderate-sized package, like SO-8,
and one that is reasonably priced. The switching losses are
not an issue for the low side MOSFET because it operates at
zero-voltage-switching.
Choose an Schottky diode, in parallel with the low side
MOSFET Q2, with a forward voltage drop low enough to
prevent the low-side MOSFET Q2 body-diode from turning
on during the dead time. This also reduces the power loss in
the high-side MOSFET associated with the reverse recovery
of the low-side MOSFET Q2 body diode. As a general rule,
select a diode with a DC current rating equal to one-third of
the load current. One option is to choose a combined
MOSFET with the Schottky diode in a single package. The
integrated packages may work better in practice because
there is less stray inductance due to short connection. This
Schottky diode is optional and may be removed if efficiency
loss can be tolerated.
Loop Compensation Design
ISL6232 uses constant frequency peak current mode control
architecture to achieve fast loop transient response. An
accurate current sensing resistor in series with the output
FN9116.0
April 18, 2005
ISL6232
inductor, or DCR of the output inductor, is used for peak
current control signal and overcurrent protection. The
inductor is not considered as a state variable since its peak
current is constant, and the system becomes single order
system. It is much easier to design a type II compensator to
stabilize the loop than to implement voltage mode control.
Peak current mode control has inherent input voltage feedforward function to achieve good line regulation. Figure 32
shows the small signal model of the synchronous buck
regulator.
^
iL
^
^
Vin
^
1:D
I Ld
Vind^
+
RT
Rc
Ro
K
C
1
1
Where ω esr = --------------- ,Q p ≈ R o ------o- ,ω o = --------------Rc Co
L
LC o
Transfer function F2(S) from control to inductor current is
given by:
(EQ. 24)
T i ( S ) = R T F m F 2 ( S )H e ( S )
The voltage loop gain with open current loop is:
Tv (S)
He(S)
v^comp
(EQ. 23)
Current loop gain Ti(S) is expressed as the following
equation:
Ti(S)
Fm
+
S
1 + -----------ω esr
vˆ o
F 1 ( S ) = ------ = V in --------------------------------------2
dˆ
S
S
------- + --------------- + 1
2 ω Q
o
p
ωo
1
where ω z = -------------Ro Co .
Co
d^
Transfer function F1(S) from control to output voltage is:
S
1 + -----ˆI
ωz
V in
o
F 2 ( S ) = ---- = --------------------- --------------------------------------Ro + RL 2
dˆ
S
S
------- + --------------- + 1
2 ω Q
o p
ωo
vo^
L
+
i in
Power Stage Transfer Functions
T v ( S ) = KFm F 1 ( S )A v ( S )
(EQ. 25)
-Av(S)
FIGURE 32. SMALL SIGNAL MODEL OF SYNCHRONOUS
BUCK REGULATOR
PWM COMPARATOR GAIN FM:
The PWM comparator gain Fm for peak current mode
control is given by:
1
dˆ
F m = ----------------- = -------------------------------ˆv
S
(
+
e S n )T s
comp
(EQ. 20)
Where Se is the slew rate of the slope compensation and Sn
is given by
V in – V o
S n = R t --------------------L
(EQ. 21)
where RT is trans-resistance, and is the product of the
current sensing resistance and gain of the current amplifier
in current loop.
The Voltage loop gain with current loop closed is given by:
Tv ( S )
L v ( S ) = ----------------------1 + Ti ( S )
(EQ. 26)
V FB
K = ----------- , V
FB is the feedback voltage of the voltage
Where
Vo
error amplifier. If Ti(S)>>1, then the above equation can be
simplified as follows:
S
1 + -----------ω esr A v ( S )
V FB R o + R L
1
L v ( S ) = ----------- --------------------- ---------------------- ---------------- , ω p ≈ --------------Vo
S H (S)
RT
Ro Co
1 + ------- e
ωp
(EQ. 27)
From the above equation, it is shown that the system is a
single order system, which has a single pole located at ω p
before the half switching frequency. Therefore, a simple type
II compensator can be easily used to stabilize the system.
CURRENT SAMPLING TRANSFER FUNCTION He(S):
In current loop, the current signal is sampled every switching
cycle. It has the following transfer function:
2
(EQ. 22)
S
S
H e ( S ) = ------- + --------------- + 1
2 ω Q
n n
ωn
where Qn and ωn are given by
22
2
Q n = – ---, = ω n = πf s
π
FN9116.0
April 18, 2005
ISL6232
Example: Vin = 12V, Vo = 5V, Io = 5A, fs = 300kHz,
Co = 180µF/12mΩ, L = 6.8µH, gm = 100µS, RT = 0.128
(Rcs = 8mΩ, Ac = 16), VFB = 0.8V, Se = 1.5×105V/s,
Sn = 1.318×105V/s, fc = 45kHz, then compensator
resistance R1 = 400kΩ.
Vo
R2
C3
V FB
V REF
-
V COMP
Put the compensator zero at 1.5kHz (~1.5x CoRo), and put
the compensator pole at esr zero which is 49kHz. The
compensator capacitors are:
GM
+
R1
C2
C1 = 270pF, C2 = 10pF (There is approximately 8pF
parasitic capacitance from VCOMP to GND; Therefore, C2
optional).
C1
Figure 34 shows the simulated voltage loop gain. It is shown
that it has 30kHz loop bandwidth with 85° phase margin and
20dB gain margin.
FIGURE 33. TYPE II COMPENSATOR
Figure 33 shows the type II compensator and its transfer
function is expressed as follows:
40
(EQ. 28)
GAIN (dB)
S
S 
 1 + ------------ 1 + -------------

ω cz2
gm
ω cz1 
vˆ comp
- = --------------------- --------------------------------------------------------S) = ---------------C1 + C2
S
vˆ FB
S  1 + ----------

ω cp
60
where
VLOOP
20
0
-20
C1 + C2
1
1 -, ω = ---------------------ω cz1 = --------------- , ω cz2 = -------------R1 C1
R 2 C 3 cp R 1 C 1 C 2
-40
-60
100
1·103
Compensator design goal:
1·104
1·105
1·106
FREQUENCY (Hz)
High DC gain
1
110
1
- f
Loop bandwidth fc:  --4- to ----10 s
85
Phase margin: 40°
60
PHASE(o)
Gain margin: >10dB
The compensator design procedure is as follows:
1
--------------Put compensator zero ω cz1 = ( 1to3 ) R
C
o o
Put one compensator pole at zero frequency to achieve high
DC gain, and put another compensator pole at either esr
zero frequency or half switching frequency, whichever is
lower. ωCZ2 is an internal zero due to 8pF and 600kΩ.
The loop gain Tv(S) at cross over frequency of fc has unity
gain. Therefore, the compensator resistance R1 is
determined by
2πf c V o C o R T
R 1 = ----------------------------------g m V FB
(EQ. 29)
where gm is the trans-conductance of the voltage error
amplifier. Compensator capacitor C1 is then given by
1
1
C 1 = ----------------- ,C 2 = ------------------------R 1 ω cz
2πR 1 f esr
(EQ. 30)
23
VLOOP
35
10
-15
-40
100
1·103
1·104
1·105
1·106
FIGURE 34. SIMULATED LOOP GAIN
12V Auxiliary Supply
A flyback transformer, or coupled inductor can be substituted
for the inductor in 5V or 3.3V supply to generate an 12V
auxiliary output as shown in Figure 35, which can be used to
drive N-channel MOSFETs. The ISL6232 is particularly well
suited for such applications because it can be configured in
ultrasonic or forced PWM mode to ensure good load
regulation when the main supplies are in light load
FN9116.0
April 18, 2005
ISL6232
conditions. An additional post-regulation circuit can be used
to improve load regulation if necessary.
The power requirements of the auxiliary supply must be
considered in the design of the main output. The flyback
transformer must be designed to deliver the required current
in both the primary and the secondary outputs with the
proper turns ratio and inductance. The overcurrent limit
threshold may also be adjusted accordingly. Power from the
main and secondary outputs is combined to get an
equivalent current referred to the main output, which is given
by the following equation.
(EQ. 31)
P main + P auxiliary
I total = ------------------------------------------------V OUT
where Pmain and Pauxiliary are the main power and auxiliary
power, respectively.
For the circuit in Figure 35, the turns ratio N of the flyback is
determined by
(EQ. 32)
V SEC + V F – V OUT
N = --------------------------------------------------V OUT + V RECT
where VSEC is the minimum required rectified secondary
voltage, VF is the forward voltage drop across the secondary
rectifier, and VRECT is the on-state voltage drop across the
synchronous rectifier MOSFET. The secondary rectifier in
the flyback must withstand flyback voltages, which is given
by the following formula
(EQ. 33)
V REV = V SEC + N • ( V IN – V OUT )
The secondary rectifier's reverse breakdown voltage rating
must also accommodate any ringings due to leakage
inductance. This voltage ringings can be minimized by
adding a snubber circuit across the secondary rectifier. Its
current rating should be at least twice the DC load current on
the auxiliary output. The optional linear post regulator must
be selected to deliver the required load current, and it should
be configured to run close to dropout to minimize power
dissipation.
R
PHASE5
1:N
Q2
• Use a star ground connection on the power plane to
minimize the cross-talk between OUT3 and OUT5.
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
• Keep the power traces and load connections short. This
practice is essential for high efficiency. Using thick copper
PC boards (2oz vs. 1oz) can enhance full-load efficiency
by 1% or more. Correctly routing PC board traces must be
approached in terms of fractions of centimeters, where a
single milliohm of excess trace resistance causes a
measurable efficiency loss.
• When trade-offs in trace lengths must be made, it is
preferable to allow the inductor-charging path to be made
longer than the discharge path. For example, it is better to
allow some extra distance between the input capacitors
and the high-side MOSFET than to allow distance
between the inductor and the synchronous rectifier or
between the inductor and the output filter capacitor,
because the synchronous rectifier conduction time is
usually longer than that of high-side MOSFET.
• Ensure that the OUT connection to the output capacitors is
short and direct. This reduces the voltage spike or dip due
to the trace resistance between OUT and output
capacitors.
• Keep the FB traces as short as possible for good radiated
immunity design.
Q1
1:
N
• Isolate the power components from the sensitive analog
components. Use a separate power plane ground and
signal power ground if possible.
12V/20mA
Optional
UGATE5
Use the following guidelines for good PC board layout:
12V
ISL6232
D1
Careful PC board layout is critical to achieve minimal
switching losses and clean, stable operation. This is
especially true when multiple converters are on the same PC
circuit board, where one circuit can affect the other due to
the noise coupling through the power ground. The switching
power stages require particular attention. Mount all of the
power components on the top-side of the board with their
ground terminals flush against one another, if possible.
• Route high-speed switching nodes (BOOT, UGATE,
PHASE, and LGATE) away from sensitive analog areas
(REF, COMP, FB, and CS). Use PGND3 and PGND5 as
an EMI shield to keep radiated switching noise away from
the ICs feedback divider and analog bypass capacitors.
LDO
BOOT5
PCB Layout Guidelines
D2
4.7µF
4.7
OUT5
T: L1- 6.8 ÿµH
H
N = 2.2
LGATE5
T: Delta Electronics
STQ125-6822
FIGURE 35. FLYBACK SECONDARY OUTPUT
24
FN9116.0
April 18, 2005
ISL6232
Quarter Size Outline Plastic Packages
(QSOP)
M28.15
28 LEAD QUARTER SIZE OUTLINE
PLASTIC PACKAGE
INCHES
N
INDEX
AREA
H
0.25(0.010) M
GAUGE
PLANE
-B1
2
3
0.25
0.010
SEATING PLANE
-A-
B M
E
h x 45o
A
D
L
α
e
A2
A1
B
C
0.10(0.004)
0.17(0.007) M
C A M
MIN
MAX
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
A2
-
0.061
-
1.54
-
B
0.008
0.012
0.20
0.30
9
C
0.007
0.010
0.18
0.25
-
D
0.386
0.394
9.81
10.00
3
E
0.150
0.157
3.81
3.98
4
e
-C-
B S
0.025 BSC
0.635 BSC
-
H
0.228
0.244
5.80
6.19
-
h
0.0099
0.0196
0.26
0.49
5
L
0.016
0.050
0.41
N
α
NOTES:
MILLIMETERS
SYMBOL
28
0o
1.27
28
8o
0o
6
7
8o
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication Number 95.
Rev. 0 2/95
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch)
per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B”
dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
25
FN9116.0
April 18, 2005