MIC28512 70V, 2A Synchronous Buck Regulator Features General Description • 4.6V to 70V Operating Input Voltage Supply • Up to 2A Output Current • Integrated High-Side and Low-Side N-Channel MOSFETs • HyperLight Load (MIC28512-1) and Hyper Speed Control (MIC28512-2) Architecture • Enable Input and Power Good (PGOOD) Output • Programmable Current-Limit and Foldback “Hiccup” Mode Short-Circuit Protection • Built-In 5V Regulator for Single-Supply Operation • Adjustable 200 kHz to 680 kHz Switching Frequency • Fixed 5 ms Soft-Start • Internal Compensation and Thermal Shutdown • Thermally-Enhanced 24-Pin 3 mm x 4 mm FCQFN Package • –40°C to +125°C Junction Temperature Range The MIC28512 is a synchronous step-down switching regulator with internal power switches capable of providing up to 2A output current from a wide input supply range from 4.6V to 70V. The output voltage is adjustable down to 0.8V with a guaranteed accuracy of ±1%. A constant switching frequency can be programmed from 200 kHz to 680 kHz. The Hyper Speed Control® and HyperLight Load® architectures of the MIC28512 allow for high VIN (low VOUT) operation and ultra-fast transient response while reducing the required output capacitance and providing very good light-load efficiency. The MIC28512 offers a full suite of features to ensure protection under fault conditions. These include undervoltage lockout to ensure proper operation under power sag conditions, internal soft-start to reduce inrush current, foldback current-limit, “hiccup” mode short-circuit protection and thermal shutdown. Applications 2016 Microchip Technology Inc. Package Type FREQ PGND PGND SW MIC28512 24-Pin 3 mm x 4 mm FCQFN (FL) 24 23 22 21 1 20 PVDD PGND 2 19 VDD DH 3 18 ILIM PVIN 4 17 VIN LX 5 16 EN BST 6 27 (SW) 7 FB 8 13 AGND 9 10 11 12 SW PGOOD 14 PGND PVIN 15 PGND PVIN 26 (PGND) DL 25 (PVIN) Industrial Power Supplies Distributed Supply Regulation Base Station Power Supplies Wall Transformer Regulation High-Voltage Single-Board Systems PVIN • • • • • DS20005524A-page 1 MIC28512 Typical Application Circuit MIC28512 3x4 FQFN VDD PVDD BST EN VIN 5.5V – 70V VOUT 5V (0A TO 2A) ILIM PVIN SW MIC28512 VIN FREQ LX FB AGND PGND Functional Block Diagram VIN CIN DBST CVDD VIN VDD PVDD 17 19 PVIN 20 4, 7, 8, 9, 25 BST LINEAR REGULATOR 6 UVLO RBST 3 DH THERMAL SHUTDOWN ON HSD OFF M1 16 R3 EN R4 FREQ CBST L 12, SW 21, 27 FIXED TON ESTIMATION 24 CONTROL LOGIC 5 ZCD VOUT LX SOFT-START LSD 3.3V 15 POWER GOOD COMPARATOR gm PGOOD X90% COMPENSATION RPGOOD CURRENT LIMIT DETECTION COUT 1 DL PVDD RLIM M2 PGND 10, 11, 22, 23, 26 RINJ 2 VREF 0.8V PGND CINJ 18 ILIM R1 13 AGND 14 CFF FB R2 DS20005524A-page 2 2016 Microchip Technology Inc. MIC28512 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings † PVIN, VIN to PGND..................................................................................................................................... –0.3V to +76V VDD, PVDD to PGND..................................................................................................................................... –0.3V to +6V VBST to VSW, VLX ......................................................................................................................................... –0.3V to +6V VBST to PGND.....................................................................................................................................–0.3V to (VIN + 6V) VSW, VLX to PGND ...........................................................................................................................–0.3V to (VIN + 0.3V) VFB, VPG, VFREQ, VILIM, VEN to AGND........................................................................................... –0.3V to (VDD + 0.3V) PGND to AGND ........................................................................................................................................ –0.3V to +0.3V ESD Rating(1) (HBM) .............................................................................................................................................. 1.5 kV ESD Rating(1) (MM) ..................................................................................................................................................150V Operating Ratings ‡ Supply Voltage (PVIN, VIN)......................................................................................................................... +4.6V to +70V Enable Input (VEN) ..............................................................................................................................................0V to VIN VSW, VFREQ, VILIM, VEN ......................................................................................................................................0V to VIN † Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended periods may affect device reliability. ‡ Notice: The device is not guaranteed to function outside its operating ratings. Note 1: Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5 kΩ in series with 100 pF. 2016 Microchip Technology Inc. DS20005524A-page 3 MIC28512 TABLE 1-1: ELECTRICAL CHARACTERISTICS Electrical Characteristics: VIN = 12V, TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C. (Note 1). Parameters Min. Typ. Max. Units Conditions Input Voltage Range (PVIN, VIN) 4.6 — 70 V Quiescent Supply Current — 0.4 0.75 mA — 0.7 1.5 — 0.1 10 µA SW unconnected, VEN = 0V VDD Output Voltage 4.8 5.2 5.4 V VIN = 7V to 60V, IVDD = 10 mA VDD UVLO Threshold 3.8 4.2 4.6 V VDD rising VDD UVLO Hysteresis — 400 — mV — Load Regulation at 40 mA 0.6 2 4.0 % — 0.792 0.8 0.808 V +25°C (±1.0%) 0.784 0.8 0.816 — 5 500 nA VFB = 0.8V EN Logic Level High 1.8 — — V — EN Logic Level Low — — 0.6 EN Hysteresis — 200 — mV — EN Bias Current — 23 40 µA VEN = 12V 450 680 800 kHz VFREQ = VIN — 340 — Maximum Duty Cycle — 85 — Minimum Duty Cycle — 0 — 110 200 270 ns — High-Side NMOS On-Resistance — 77 — mΩ — Low-Side NMOS On-Resistance — 43 — Current-Limit Threshold –30 –14 0 Short-Circuit Threshold –24 –7 8 Current-Limit Source Current 50 70 90 Short-Circuit Source Current 25 36 43 Power Supply Input Shutdown Supply Current — VFB = 1.5V (MIC28512-1) VFB = 1.5V (MIC28512-2) VDD Supply Reference Feedback Reference Voltage FB Bias Current –40°C ≤ TJ ≤ +125°C (±2%) Enable Control — Oscillator Switching Frequency Minimum Off-Time VFREQ = 50% x VIN % — VFB > 0.8V Internal MOSFET — Short-Circuit Protection Note 1: mV VFB = 0.79V VFB = 0V µA VFB = 0.79V VFB = 0V Specification for packaged product only. DS20005524A-page 4 2016 Microchip Technology Inc. MIC28512 TABLE 1-1: ELECTRICAL CHARACTERISTICS (CONTINUED) Electrical Characteristics: VIN = 12V, TA = 25°C, unless noted. Bold values indicate –40°C ≤ TJ ≤ +125°C. (Note 1). Parameters Min. Typ. Max. Units Conditions — — 50 µA PGOOD Threshold Voltage 85 90 95 %VOUT PGOOD Hysteresis — 6 — PGOOD Delay Time — 100 — µs Sweep VFB from low to high PGOOD Low Voltage — 70 200 mV VFB < 90% x VNOM, IPGOOD = 1 mA Overtemperature Shutdown — 160 — °C TJ Rising Overtemperature Shutdown Hysteresis — 15 — °C — — 5 — ms — Leakage SW, BST Leakage Current — Power Good (PGOOD) Sweep VFB from low to high Sweep VFB from low to high Thermal Protection Soft-Start Soft-Start Time Note 1: Specification for packaged product only. 2016 Microchip Technology Inc. DS20005524A-page 5 MIC28512 TEMPERATURE SPECIFICATIONS Parameters Sym. Min. Typ. Max. Units Conditions Junction Operating Temperature TJ –40 — +150 °C — Storage Temperature Range TS –65 — +150 °C — Junction Temperature TJ — — +150 °C — Lead Temperature — — — +300 °C Soldering, 10s JA — 30 — °C/W Temperature Ranges Package Thermal Resistances Thermal Resistance 3 mm x 4 mm FCQFN-24LD DS20005524A-page 6 — 2016 Microchip Technology Inc. MIC28512 2.0 TYPICAL PERFORMANCE CURVES The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: 4.9 VOUT = 5V IOUT = 0A FSW = 300kHz 4.0 Rising 4.7 4.5 VIN THRESHOLD (V) SUPPLY CURRENT (mA) 5.0 3.0 2.0 1.0 4.3 4.1 Falling 3.9 3.7 VIN=12V IOUT= 0A 3.5 0.0 3.3 5 10 15 20 25 30 35 40 45 50 55 60 65 70 INPUT VOLTAGE (V) 6.0 30.0 5.0 20.0 15.0 10.0 0 FIGURE 2-4: Temperature. 35.0 25.0 -25 0.0 5.0 FIGURE 2-2: Input Voltage. 18.0 31.0 44.0 57.0 INPUT VOLTAGE (V) 75 100 125 VIN ULVO Threshold vs. VIN= 12V VOUT = 5.0V FSW = 300kHz 4.0 3.0 2.0 0.0 -50 70.0 VIN Shutdown Current vs. -25 FIGURE 2-5: vs. Temperature. 1.50 0 25 50 75 TEMPERATURE (°C) 100 125 Output Peak Current Limit 0.812 VIN= 12V VOUT = 5.0V IOUT = 0A 1.20 Falling 0.90 0.60 Hyst FEEDBACK VOLTAGE (V) Rising ENABLE THRESHOLD (V) 50 1.0 5.0 0.30 25 TEMPERATURE (°C) CURRENT LIMIT (A) SHUTDOWN CURRENT (µA) FIGURE 2-1: VIN Operating Supply Current vs. Input Voltage (MIC28512-1). -50 0.808 0.804 0.800 0.796 0.00 5 10 15 20 25 30 35 40 45 50 55 60 65 70 INPUT VOLTAGE (V) FIGURE 2-3: Voltage. Enable Threshold vs. Input 2016 Microchip Technology Inc. 0.792 -50 FIGURE 2-6: Temperature -25 0 25 50 75 100 TEMPERATURE (°C) 125 Feedback Voltage vs. DS20005524A-page 7 MIC28512 . 5.2 5.3 5.1 5.0 4.9 VOUT = 5V IOUT = 1A FSW= 300kHz 4.8 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 5.2 5.1 5.0 4.9 VIN= 12V VOUT = 5V 4.7 4.8 0.0 5 10 15 20 25 30 35 40 45 50 55 60 65 70 INPUT VOLTAGE (V) FIGURE 2-7: Voltage. Output Voltage vs. Input FIGURE 2-10: Current. Output Voltage vs. Output 5.0V 3.3V 90 EFFICIENCY (%) Rising 1.4 1.3 1.2 1.1 Falling 1.0 0.9 0.8 80 70 60 50 40 30 0.7 VIN= 12V VDD= 5V -25 FIGURE 2-8: Temperature. 0 25 50 75 TEMPERATURE (°C) 100 20 10 0.01 125 Enable Threshold vs. 600 100 550 90 500 80 450 400 350 300 250 100 0.0 FIGURE 2-9: Output Current. DS20005524A-page 8 0.1 1 OUTPUT CURRENT (A) 10 5.0V 3.3V 70 60 50 40 30 200 150 FSW = 300kHz FIGURE 2-11: Efficiency (VIN = 12V) vs. Output Current MIC28512-1. EFFICIENCY (%) ENABLE THRESHOLD (V) 1.5 0.6 SWITCHING FREQUENCY (kHz) 2.0 100 1.6 0.5 -50 0.5 1.0 1.5 OUTPUT CURRENT (A) VIN= 12V VOUT = 5V 0.5 1.0 1.5 OUTPUT CURRENT (A) 20 2.0 Switching Frequency vs. 10 0.01 FSW = 300kHz 0.1 1 OUTPUT CURRENT (A) 10 FIGURE 2-12: Efficiency (VIN = 24V) vs. Output Current MIC28512-1. 2016 Microchip Technology Inc. MIC28512 5.2 EFFICIENCY (%) 90 5.0V 3.3V 80 70 60 50 40 OUTPUT VOLTAGE (V) 100 30 10 0.01 4.9 VIN= 12V = 5V VOUT 4.8 0.0 0.1 1 10 OUTPUT CURRENT (A) FIGURE 2-13: Efficiency (VIN = 48V) vs. Output Current MIC28512-1. 20.0 2.0 100 18.0 5.0V 3.3V 90 16.0 14.0 12.0 10.0 8.0 6.0 4.0 70 60 50 40 FSW= 300kHz 20 10 0.01 5 10 15 20 25 30 35 40 45 50 55 60 65 70 INPUT VOLTAGE (V) FIGURE 2-14: VIN Operating Supply Current vs. Input Voltage MIC28512-2. 80 30 VOUT = 5V IOUT = 0A FSW = 300kHz 2.0 0.0 0.5 1.0 1.5 OUTPUT CURRENT (A) FIGURE 2-16: Output Voltage vs. Output Current MIC28512-2. EFFICIENCY (%) SUPPLY CURRENT (mA) 5.0 FSW = 300kHz 20 0.1 1 OUTPUT CURRENT (A) 0.812 100 VIN = 12V VOUT = 5.0V IOUT = 0A 0.808 0.804 0.800 0.796 90 5.0V 3.3V 80 70 60 50 40 30 FSW = 300kHz 20 0.792 -50 10 FIGURE 2-17: Efficiency (VIN = 12V) vs. Output Current MIC28512-2. EFFICIENCY (%) FEEDBACK VOLTAGE (V) 5.1 -25 0 25 50 75 100 125 TEMPERATURE (°C) FIGURE 2-15: Feedback Voltage vs. Temperature MIC28512-2 2016 Microchip Technology Inc. 10 0.01 0.1 1 OUTPUT CURRENT (A) 10 FIGURE 2-18: Efficiency (VIN = 24V) vs. Output Current MIC28512-2. DS20005524A-page 9 MIC28512 100 EFFICIENCY (%) 5.0V 3.3V 80 70 60 50 40 30 FSW = 300kHz 20 IC POWER DISSIPATION (W) 2.4 90 0.1 1 OUTPUT CURRENT (A) 0.4 0 0.5 1 1.5 OUTPUT CURRENT (A) 2 IC Power Dissipation vs. 2.5 VIN =12V fSW = 300kHz 0.8 OUTPUT CURRENT (A) IC POWER DISSIPATION (W) 0.8 FIGURE 2-22: Output Current. 1.0 0.6 0.4 5.0V 3.3V 0.2 0 0.5 1 1.5 OUTPUT CURRENT (A) FIGURE 2-20: Output Current. 2 2.0 5.0V 3.3V 1.5 Vin =24V =12V fSW = 300kHz Tjmax =125°C Θja = 30°C/W 40°C/W 1.0 0.5 0.0 25 40 55 70 85 100 AMBIENT TEMPERATURE (°C) IC Power Dissipation vs. FIGURE 2-23: 1.2 12V Input Thermal Derating. 2.5 OUTPUT CURRENT (A) IC POWER DISSIPATION (W) 5.0V 3.3V 1.2 10 FIGURE 2-19: Efficiency (VIN = 48V) vs. Output Current MIC28512-2. 0.0 1.6 0.0 10 0.01 Vin VIN ==24V 48V fSW == 300kHz 300kHz SW 2.0 VIN =24V Vin =24V fSW = 300kHz 1.0 0.8 5.0V 3.3V 0.6 0.4 0.2 0.0 0 0.5 1 1.5 2 2.0 1.5 1.0 DS20005524A-page 10 IC Power Dissipation vs. Vin =24V fSW = 300kHz Tjmax =125°C Θja = 30°C/W 0.5 0.0 25 OUTPUT CURRENT (A) FIGURE 2-21: Output Current. 5.0V 3.3V FIGURE 2-24: (MIC28512-1). 40 55 70 85 100 AMBIENT TEMPERATURE (°C) 24V Input Thermal Derating 2016 Microchip Technology Inc. MIC28512 OUTPUT CURRENT (A) 2.5 2.0 1.5 1.0 VIN (10V/div) 5.0V 3.3V VSW (10V/div) VIN =48V fSW = 300kHz TJmax =125°C ΘJA = 30°C/W VOUT (5V/div) VIN = 12V VOUT = 5V IOUT = 2A 0.5 0.0 25 40 55 70 85 100 AMBIENT TEMPERATURE (°C) FIGURE 2-25: (MIC28512-1). 48V Input Thermal Derating ILX (2A/div) Time (2ms/div) FIGURE 2-28: (MIC28512-2). VIN Shutdown SWITCHING FREQUENCY (KHz) 600 590 580 VEN (2V/div) 570 560 550 540 VOUT (2V/div) 530 520 510 VIN = 12V VOUT = 5V IOUT = 2A IL (1A/div) 500 7.0 16.0 25.0 34.0 43.0 52.0 61.0 70.0 INPUT VOLTAGE (V) Time (2μs/div) FIGURE 2-26: Switching Frequency vs. Input Voltage (MIC28512-1). VIN (10V/div) FIGURE 2-29: Enable Turn-On Delay and Rise Time (MIC28512-2). VEN (2V/div) VSW (10V/div) VOUT (2V/div) VOUT (5V/div) VIN = 12V VOUT = 5V IOUT = 2A LX (2A/div) Time (4ms/div) FIGURE 2-27: MIC28512-2. VIN Soft Start Turn-On 2016 Microchip Technology Inc. IL (1A/div) VIN = 12V VOUT = 5V IOUT = 2A Time (2ms/div) FIGURE 2-30: Enable Turn-Off Delay and Rise Time (MIC28512-2). DS20005524A-page 11 MIC28512 VIN (2V/div) VSW (10V/div) VIN = 12V VOUT = 5V VEN (1V/div) VIN = 12V VOUT = 5V IOUT = 2A VOUT (2V/div) ILX (1A/div) ILX (2A/div) Time (400ms/div) Time (1ms/div) FIGURE 2-31: VIN Start-Up with 1V Pre-Biased Output (MIC28512-2). FIGURE 2-34: Enable into Short Circuit. IDC = 10mA ILDO = 1mA VIN = 3.6V VIN (10V/div) VSW (010V/div) VOUT (2V/div) VIN = 12V VOUT = 5V IOUT = 2A VOUT (2V/div) VIN = 12V VOUT = 5V IL (1A/div) ILX (2A/div) Time (4μs/div) Time (400ms/div) FIGURE 2-32: VIN Start-Up with 2V Pre-Biased Output (MIC28512-2). FIGURE 2-35: Inductor Peak Current (Output Short Applied) (MIC28512-2). VIN = 12V VOUT = 5V VIN = 12V VOUT = 5V VIN (5V/div) VOUT (2V/div) ILX (1A/div) IL (2A/div) Time (20μs/div) Time (200ms/div) FIGURE 2-33: (MIC28512-2). DS20005524A-page 12 Start-Up Into Short Circuit FIGURE 2-36: Short Circuit (MIC28512-2). 2016 Microchip Technology Inc. MIC28512 VOUT (2V/div) VIN = 12V VOUT = 5V VOUT (2V/div) VIN = 12V VOUT = 5V IL (1A/div) IL (1A/div) Time (1ms/div) Time (4μs/div) FIGURE 2-37: Inductor Peak Current (Output Short Applied) (MIC28512-1). FIGURE 2-40: Output Recovery from Short Circuit to No Load. VIN = 12V VOUT = 5V VIN = 12V VOUT = 5V VOUT (2V/div) VOUT (2V/div) IL (1A/div) IL (1A/div) Time (20μs/div) FIGURE 2-38: Time (100μs/div) Short Circuit (MIC28512-1). VIN = 12V VOUT = 5V VOUT (2V/div) IL (1A/div) Short Circuit. (MIC28512-1) VIN = 12V VOUT = 5V VIN (5V/div) IL (1A/div) Time (1ms/div) FIGURE 2-39: Circuit to 2A. FIGURE 2-41: Output Recovery from Short 2016 Microchip Technology Inc. Time (100μs/div) FIGURE 2-42: Input Voltage Applied with Output Shorted (MIC28512-1). DS20005524A-page 13 MIC28512 VIN (10V/div) VIN (5V/div) VOUT (2V/div) VIN = 12V VOUT = 5V IOUT = 2A IL (1A/div) Time (1ms/div) FIGURE 2-43: (MIC28512-1). VIN Soft Start Turn-On VIN (5V/div) IL (1A/div) FIGURE 2-46: (MIC28512-1). VIN = 12V VOUT = 5V IOUT = 2A IL (1A/div) VPREBIAS = 2V IOUT = 0A VOUT = 5V VIN = 12V Time (1ms/div) VIN Shutdown FIGURE 2-47: VIN Start-up w Pre-Biased Output (MIC28512-1). VIN (5V/div) IOUT = 0A VOUT = 5V VIN = 12V VOUT (2V/div) IL (1A/div) VPREBIAS = 2V IOUT = 2A VOUT = 5V VIN = 12V VOUT (2V/div) IL (1A/div) Time (1ms/div) DS20005524A-page 14 Soft Start Power Good IL (1A/div) Time (200μs/div) FIGURE 2-45: Time (1ms/div) VOUT (2V/div) VOUT (2V/div) FIGURE 2-44: (MIC28512-1). IOUT = 2A VOUT = 5V VIN = 12V VOUT (2V/div) VPGOOD (2V/div) Soft Start (MIC28512-1). Time (1ms/div) FIGURE 2-48: VIN Start-up w Pre-Biased Output (MIC28512-1). 2016 Microchip Technology Inc. MIC28512 VOUT AC-Coupled (50mV/div) IOUT = 0A VIN = 12V VOUT = 5V VOUT (AC-Coupled) (20mV/div) VSW (5V/div) VSW (5V/div) Time (1μs/div) Time (400μs/div) FIGURE 2-49: Waveform. MIC58512-1 Switching VOUT AC-Coupled (20mV/div) IOUT = 0A VIN = 12V VOUT = 5V VSW (5V/div) FIGURE 2-52: Waveform. IOUT = 0A to 2A VIN = 12V VOUT = 5V IIL (1A/div) Time (4ms/div) MIC58512-2 Switching VOUT (AC-Coupled) (20mV/div) IOUT = 2A VIN = 12V VOUT = 5V FIGURE 2-53: Response. MIC28512-1 Transient IOUT = 0A to 2A VIN = 12V VOUT = 5V VOUT (AC-Coupled) (100mV/div) IIL (1A/div) VSW (5V/div) Time (40μs/div) Time (1μs/div) FIGURE 2-51: Waveform.. MIC58512-2 Switching VOUT (AC-Coupled) (100mV/div) Time (1μs/div) FIGURE 2-50: Waveform. IOUT = 2A VIN = 12V VOUT = 5V MIC58512-1 Switching 2016 Microchip Technology Inc. FIGURE 2-54: Response. MIC28512-2 Transient DS20005524A-page 15 MIC28512 IOUT = 0A to 0.6A VIN = 12V VOUT = 5V IOUT = 0.6A to 1.2A VIN = 12V VOUT = 5V VOUT (AC-Coupled) (50mV/div) VOUT (AC-Coupled) (50mV/div) IIL (500mA/div) IIL (1A/div) Time (1ms/div) FIGURE 2-55: Response. MIC28512-1 Transient Time (40μs/div) FIGURE 2-58: Response. IOUT = 1.2A to 2A VIN = 12V VOUT = 5V IOUT = 0A to 0.6A VIN = 12V VOUT = 5V VOUT (AC-Coupled) (50mV/div) VOUT (AC-Coupled) (100mV/div) IIL (500mA/div) IIL (1A/div) Time (40μs/div) Time (40μs/div) FIGURE 2-56: Response. MIC28512-2 Transient IOUT = 0.6A to 1.2A VIN = 12V VOUT = 5V VOUT (AC-Coupled) (50mV/div) IIL (0.5A/div) FIGURE 2-59: Response. VOUT (AC-Coupled) (100mV/div) DS20005524A-page 16 MIC28512-1 Transient IOUT = 1.2A to 2A VIN = 12V VOUT = 5V IIL (1A/div) Time (40μs/div) Time (40μs/div) FIGURE 2-57: Response. MIC28512-2 Transient MIC28512-1 Transient FIGURE 2-60: Response. MIC28512-2 Transient 2016 Microchip Technology Inc. MIC28512 3.0 PIN DESCRIPTIONS The descriptions of the pins are listed in Table 3-1. TABLE 3-1: PIN FUNCTION TABLE Pin Number Symbol Description 1 DL Low-Side Gate Drive. Internal low-side power MOSFET gate connection. This pin must be left unconnected or floating. 2 PGND PGND is the return path for the low-side driver circuit. Connect to the source of low-side MOSFET’s (PGND, pins 10, 11 22, 23, and 26) through a low-impedance path. 3 DH 4, 7, 8, 9, 25 (25 is ePad) PVIN Power Input Voltage. The PVIN pins supply power to the internal power switch. Connect all PVIN pins together and bypass locally with ceramic capacitors. The positive terminal of the input capacitor should be placed as close as possible to the PVIN pins, the negative terminal of the input capacitor should be placed as close as possible to the PGND pins 10,11, 22, 23, and 26. 5 LX The LX pin is the return path for the high-side driver circuit. Connect the negative terminal of the bootstrap capacitor directly to this pin. Also connect this pin to the SW pins 12, 21, and 27, with a low-impedance path. The controller monitors voltages on this and PGND for zero current detection. 6 BST Bootstrap Pin. This pin provides bootstrap supply for the high-side gate driver circuit. Connect a 0.1 µF capacitor and an optional resistor in series from the LX (pin 5) to the BST. 10, 11, 22, 23, 26 (26 is ePad) PGND Power Ground. These pins are connected to the source of the low-side MOSFET. They are the return path for the step-down regulator power stage and should be tied together. The negative terminal of the input decoupling capacitor should be placed as close as possible to these pins. 12, 21, 27 (27 is ePad) SW Switch Node. The SW pins are the internal power switch outputs. These pins should be tied together and connected to the output inductor. 13 AGND Analog Ground. The analog ground for VDD and the control circuitry. The analog ground return path should be separate from the power ground (PGND) return path. 14 FB Feedback Input. The FB pin sets the regulated output voltage relative to the internal reference. This pin is connected to a resistor divider from the regulated output such that the FB pin is at 0.8V when the output is at the desired voltage. 15 PGOOD The power good output is an open drain output requiring an external pull-up resistor to external bias. This pin is a high impedance open circuit when the voltage at FB pin is higher than 90% of the feedback reference voltage (typically 0.8V). 16 EN Enable Input. The EN pin enables the regulator. When the pin is pulled below the threshold, the regulator will shut down to an ultra-low current state. A precise threshold voltage allows the pin to operate as an accurate UVLO. Do not tie EN to VDD 17 VIN Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to 70V. A ceramic capacitor from VIN to AGND is required for decoupling. The decoupling capacitor should be placed as close as possible to the supply pin. 18 ILIM Current Limit Setting. Connect a resistor from this pin to the SW pin node to allow for accurate current limit sensing programming of the internal low-side power MOSFET. 19 VDD Internal +5V Linear Regulator: VDD is the internal supply bus for the IC. Connect to an external 1 µF bypass capacitor. When VIN is <5.5V, this regulator operates in drop-out mode. Connect VDD to VIN. 20 PVDD A 5V supply input for the low-side N-channel MOSFET driver circuit, which can be tied to VDD externally. A 1 μF ceramic capacitor from PVDD to PGND is recommended for decoupling. 24 FREQ Switching Frequency Adjust pin. Connect this pin to VIN to operate at 680 kHz. Place a resistor divider network from VIN to the FREQ pin to program the switching frequency. High-Side Gate Drive. Internal high-side power MOSFET gate connection. This pin must be left unconnected or floating. 2016 Microchip Technology Inc. DS20005524A-page 17 MIC28512 FUNCTIONAL DESCRIPTION The MIC28512 is an adaptive on-time synchronous buck regulator with integrated high-side and low-side MOSFETs suitable for high-input voltage to low-output voltage conversion applications. It is designed to operate over a wide input voltage range (4.6V to 70V) which is suitable for automotive and industrial applications. The output is adjustable with an external resistive divider. An adaptive on-time control scheme is employed to produce a constant switching frequency in continuous-conduction mode and reduced switching frequency in discontinuous-operation mode, improving light load efficiency. Overcurrent protection is implemented by sensing low-side MOSFET’s RDS(ON). The device features internal soft-start, enable, UVLO, and thermal shutdown. 4.1 Theory of Operation As illustrated in the Functional Diagram, the output voltage of the MIC28512 is sensed by the feedback (FB) pin via the voltage dividers R1 and R2, and compared to a 0.8V reference voltage VREF at the error comparator through a low-gain transconductance (gM) amplifier. If the feedback voltage decreases and the amplifier output is below 0.8V, the error comparator will trigger the control logic and generate an ON-time period. The ON-time period length is predetermined by the “Fixed tON Estimator” circuitry: EQUATION 4-1: t ON ESTIMATED V OUT = ---------------------V IN f SW EQUATION 4-2: D MAX = 1 – t OFF MIN f SW It is not recommended to use MIC28512 with an OFF-time close to tOFF(MIN) during steady-state operation. The adaptive ON-time control scheme results in a constant switching frequency in the MIC28512. The actual ON-time and resulting switching frequency will vary with the different rising and falling times of the external MOSFETs. Also, the minimum tON results in a lower switching frequency in high VIN to VOUT applications. During load transients, the switching frequency is changed due to the varying OFF-time. Figure 4-1 shows the allowable range of the output voltage versus the input voltage. The minimum output voltage is 0.8V which is limited by the reference voltage. The maximum output voltage is 24V which is limited by the internal circuitry. 30.0 OUTPUIT VOLTAGE (V) 4.0 25.0 Fsw = 600kHz 20.0 Fsw = 400kHz Fsw = 200kHz 15.0 Allowable Range 10.0 5.0 0.8V min. Where: VOUT Output Voltage VIN Power Stage Input Voltage fSW Switching Frequency 0.0 5 duty DS20005524A-page 18 cycle is obtained 31 44 57 70 INPUT VOLTAGE (V) FIGURE 4-1: Allowable Output Voltage Range vs. Input Voltage. At the end of the ON-time period, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF-time period length depends upon the feedback voltage in most cases. When the feedback voltage decreases and the output of the gM amplifier is below 0.8V, then the ON-time period is triggered and the OFF-time period ends. If the OFF-time period determined by the feedback voltage is less than the minimum OFF-time tOFF(MIN), which is about 200 ns (typical), the MIC28512 control logic will apply the tOFF(MIN) instead. The tOFF(MIN) is required to maintain enough energy in the boost capacitor (CBST) to drive the high-side MOSFET. The maximum Equation 4-2. 18 To illustrate the control loop operation, both the steady-state and load transient scenarios will be analyzed. Figure 4-2 shows the MIC28512 control loop timing during steady-state operation. During steady-state, the gM amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the ON-time period. The ON-time is predetermined by the tON estimator. The termination of the OFF-time is controlled by the feedback voltage. At the valley of the feedback voltage ripple, which occurs when VFB falls below VREF, the OFF period ends and the next ON-time period is triggered through the control logic circuitry. from 2016 Microchip Technology Inc. MIC28512 current ripple if the ESR of the output capacitor is large enough. The MIC28512 control loop has the advantage of eliminating the need for slope compensation. IL IOUT ¨IL(PP) VOUT ¨VOUT(PP) = ESRC × ¨IL(PP) OUT VFB ¨VFB(PP) = ¨VOUT(PP) × VREF HSD R2 R1 + R2 TRIGGER ON-TIME IF VFB IS BELOW VREF ESTIMATED ON-TIME FIGURE 4-2: Timing. MIC28511 Control Loop Figure 4-3 shows the operation of the MIC28512 during a load transient. The output voltage drops due to the sudden load increase, which causes the VFB to be less than VREF. This will cause the error comparator to trigger an ON-time period. At the end of the ON-time period, a minimum OFF-time tOFF(MIN) is generated to charge CBST because the feedback voltage is still below VREF. Then, the next ON-time period is triggered due to the low feedback voltage. Therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. With the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in MIC28512 converter. FULL LOAD IOUT If a low-ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the gM amplifier and the error comparator. Also, if the ESR of the output capacitor is very low, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple. In these cases, ripple injection is required to ensure proper operation. Please refer to “Ripple Injection” subsection in Application Information for more details about the ripple injection technique. 4.2 Discontinuous Mode (MIC28512-1 Only) In continuous mode, the inductor current is always greater than zero; however, at light loads the MIC28512-1 is able to force the inductor current to operate in discontinuous mode. Discontinuous mode occurs when the inductor current falls to zero, as indicated by trace (IL) shown in Figure 4-4. During this period, the efficiency is optimized by shutting down all the non-essential circuits and minimizing the supply current. The MIC28512-1 wakes up and turns on the high-side MOSFET when the feedback voltage VFB drops below 0.8V. The MIC28512-1 has a zero crossing comparator that monitors the inductor current by sensing the voltage drop across the low-side MOSFET during its ON-time. If the VFB > 0.8V and the inductor current goes slightly negative, then the MIC28521-1 automatically powers down most of the IC circuitry and goes into a low-power mode. NO LOAD VOUT VFB VREF HSD TOFF(MIN) FIGURE 4-3: Response. In order to meet the stability requirements, the MIC28512 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the gM amplifier and the error comparator. The recommended feedback voltage ripple is 20 mV ~ 100 mV. MIC28512 Load Transient Once the MIC28512-1 goes into discontinuous mode, both DH and DL are low, which turns off the high-side and low-side MOSFETs. The load current is supplied by the output capacitors and VOUT drops. If the drop of VOUT causes VFB to go below VREF, then all the circuits will wake up into normal continuous mode. First, the bias currents of most circuits reduced during the discontinuous mode are restored, and then a tON pulse is triggered before the drivers are turned on to avoid any possible glitches. Finally, the high-side driver is turned on. Figure 4-4 shows the control loop timing in discontinuous mode. Unlike true current-mode control, the MIC28512 uses the output voltage ripple to trigger an ON-time period. The output voltage ripple is proportional to the inductor 2016 Microchip Technology Inc. DS20005524A-page 19 MIC28512 4.5 IL IL CROSSES 0 AND VFB > 0.8. DISCONTINUOUS MODE STARTS VFB < 0.8. WAKEUP FROM DISCONTINUOUS MODE. 0 VFB VREF ZC Current Limit The MIC28512 uses the RDS(ON) of the internal low-side power MOSFET to sense overcurrent conditions. In each switching cycle, the inductor current is sensed by monitoring the low-side MOSFET during its ON period. The sensed voltage, V(ILIM), is compared with the power ground (PGND) after a blanking time of 150 ns. The voltage drop of the resistor RILIM is compared with the low-side MOSFET voltage drop to set the overcurrent trip level. The small capacitor connected from the ILIM pin to PGND can be added to filter the switching node ringing, allowing a better short limit measurement. The time constant created by RILIM and the filter capacitor should be much less than the minimum off time. The overcurrent limit can be programmed by using Equation 4-3: VHSD EQUATION 4-3: VLSD ESTIMATED ON-TIME I CLIM – 0.5 I L PP R DS ON + V CL R ILIM = ---------------------------------------------------------------------------------------------------I CL Where: FIGURE 4-4: MIC28512-1 Control Loop Timing (Discontinuous Mode). During discontinuous mode, the bias current of most circuits are reduced. As a result, the total power supply current during discontinuous mode is only about 450 μA, allowing the MIC28512-1 to achieve high efficiency in light load applications. 4.3 VDD Regulator ICLIM Desired Current Limit RDS(ON) On-Resistance of Low-Side Power MOSFET (40 mΩ typical) VCL Current-limit threshold. 14 mV (typical absolute value). See the Electrical Characteristics table. ICL Current-limit source current. 70µA (typical). See the Electrical Characteristics table. ∆IL(PP) Inductor Current Peak-to-Peak Use Equation 4-4 to calculate the inductor ripple current The MIC28511 provides a 5V regulated VDD to bias internal circuitry for VIN ranging from 5.5V to 70V. When VIN is less than 5.5V, VDD should be tied to VIN pins to bypass the internal linear regulator. The peak-to-peak inductor current ripple is calculated with Equation 4-4. 4.4 EQUATION 4-4: Soft-Start Soft-start reduces the power supply inrush current at startup by controlling the output voltage rise time while the output capacitor charges. The MIC28511 implements an internal digital soft-start by ramping up the 0.8V reference voltage (VREF) from 0 to 100% in about 5 ms with 9.7 mV steps. This controls the output voltage rate of rise at turn on, minimizing inrush current and eliminating output voltage overshoot. Once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. DS20005524A-page 20 V OUT V IN MAX – V OUT I L PP = ------------------------------------------------------------------V IN MAX f SW L The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended to use the RDS(ON) at maximum junction temperature with a 20% margin to calculate RILIM in Equation 4-3. In case of hard short, the current-limit threshold is folded down to allow an indefinite hard short on the output without any destructive effect. It is mandatory to make sure that the inductor current used to charge the output capacitor during soft-start is under the folded short limit; otherwise the supply will go into hiccup mode and may not be finishing the soft-start successfully. 2016 Microchip Technology Inc. MIC28512 4.6 Power Good (PG) The power good (PG) pin is an open-drain output that indicates logic-high when the output is nominally 90% of its steady state voltage. 4.7 MOSFET Gate Drive The Functional Block Diagram shows a bootstrap circuit, consisting of DBST, CBST, and RBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode DBST is reverse-biased and CBST floats high while continuing to bias the high-side gate driver. The bias current of the high-side driver is less than 10 mA, so a 0.1 μF to 1 μF capacitor is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. ∆BST = 10 mA x 1.25 μs/0.1 μF = 125 mV. When the low-side MOSFET is turned back on, CBST is then recharged through the boost diode. A 30Ω resistor RBST, which is in series with the BST pin, is required to slow down the turn-on time of the high-side N-channel MOSFET. 2016 Microchip Technology Inc. DS20005524A-page 21 MIC28512 5.0 APPLICATION INFORMATION 5.2 5.1 Output Voltage Setting Components The MIC28512 switching frequency can be adjusted between 200 kHz and 680 kHz by changing the resistor divider network from VIN. Setting the Switching Frequency The MIC28512 requires two resistors to set the output voltage as shown in Figure 5-1. R1 FB gM AMP R2 VREF FIGURE 5-1: Configuration. Voltage Divider The output voltage is determined by Equation 5-1. FIGURE 5-2: Adjustment. Switching Frequency Equation 5-3 gives the estimated switching frequency. EQUATION 5-3: EQUATION 5-1: R4 f SW = f 0 -------------------- R3 + R4 V OUT = V FB 1 + R1 ------- R2 Where: Where: VFB 0.8V A typical value of R1 used on the standard evaluation board is 10 kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using Equation 5-2: f0 Switching frequency when R4 is open; typically 680 kHz. Figure 5-3 shows the switching frequency versus the resistor R4 when R3 = 100 kΩ. EQUATION 5-2: V FB R1 R2 = ----------------------------V OUT – V FB FIGURE 5-3: 5.3 Switching Frequency vs. R4. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and DS20005524A-page 22 2016 Microchip Technology Inc. MIC28512 MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by: EQUATION 5-4: EQUATION 5-7: 2 P L CU = I L RMS DCR The resistance of the copper wire, DCR, increases with the temperature. The value of the winding resistance used should be at the operating temperature. EQUATION 5-8: DCR HT = DCR 20C 1 + 0.0042 T H – T 20C V OUT V IN MAX – V OUT L = ------------------------------------------------------------------V IN MAX I L PP f SW Where: Where: TH Temperature of wire under full load Ambient temperature Room temperature winding resistance (usually specified by the manufacturer) fSW Switching Frequency T20C ∆IL(PP) The peak-to-peak inductor current ripple; typically 20% of the maximum output current DCR(20C) In continuous conduction mode, the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. EQUATION 5-5: I L PK = I OUT + 0.5 I L PP The RMS inductor current is used to calculate the I2R losses in the inductor. EQUATION 5-6: 5.4 Output Capacitor Selection The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Voltage and RMS current capability are also important factors in selecting an output capacitor. Recommended capacitor types are ceramic, tantalum, low-ESR aluminum electrolytic, OS-CON and POSCAP. For high ESR electrolytic capacitors, ESR is the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. For a low ESR ceramic output capacitor, ripple is dominated by the reactive impedance. The maximum value of ESR is calculated by Equation 5-9. 2 I L RMS = 2 I L PP I OUT MAX + -------------------I2 Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC28512 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used, but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized, although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by Equation 5-7: 2016 Microchip Technology Inc. EQUATION 5-9: ESR C OUT V OUT PP --------------------------I L PP Where: ∆VOUT(PP) Peak-to-Peak Output Voltage Ripple ∆IL(PP) Peak-to-Peak Inductor Current Ripple The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated by Equation 5-10. EQUATION 5-10: V OUT PP = 2 I L PP 2 ------------------------------------ C OUT f SW 8 + I L PP ESR COUT Where: D Duty Cycle COUT Output Capacitance Value fSW Switching Frequency DS20005524A-page 23 MIC28512 As described in the Theory of Operation subsection of the Functional Description, the MIC28512 requires at least 20 mV peak-to-peak ripple at the FB pin for the gM amplifier and the error comparator to operate properly. Also, the ripple on FB pin should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Refer to the Ripple Injection subsection for details. EQUATION 5-14: The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated in Equation 5-11. 5.6 EQUATION 5-11: IC OUT RMS I L PP = ----------------12 The power dissipated in the output capacitor is calculated by using: EQUATION 5-12: 2 P DISS COUT = I COUT RMS ESR COUT 5.5 I CIN RMS I OUT MAX D 1 – D The power dissipated in the input capacitor is: EQUATION 5-15: 2 P DISS CIN = I CIN RMS ESR CIN Ripple Injection The VFB ripple required for proper operation of the MIC28512’s gM amplifier and error comparator is 20 mV to 100 mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. If the feedback voltage ripple is so small that the gM amplifier and error comparator can’t sense it, then the MIC28512 will lose control and the output voltage is not regulated. In order to have some amount of VFB ripple, a ripple injection method is applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: • Enough ripple at the feedback voltage due to the large ESR of the output capacitors (Figure 5-4). The converter is stable without any ripple injection. Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: FIGURE 5-4: EQUATION 5-13: Where: V IN = I L PK ESR CIN The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: DS20005524A-page 24 Enough Ripple at FB. The feedback voltage ripple is: EQUATION 5-16: R2 V FB PP = -------------------- ESR C I L PP OUT R1 + R2 ∆IL(PP) Peak-to-Peak Value of the Inductor Current Ripple • Inadequate ripple at the feedback voltage due to the small ESR of the output capacitors. The output voltage ripple is fed into the FB pin through a feed-forward capacitor, CFF in this situation, as shown in Figure 5-5. The typical CFF value is selected by using Equation 5-17. 2016 Microchip Technology Inc. MIC28512 EQUATION 5-17: 10 R1 C FF -------f SW With the feed-forward capacitor, the feedback voltage ripple is very close to the output voltage ripple. EQUATION 5-18: V FB PP ESR I L PP In Equation 5-19 and Equation 5-20, it is assumed that the time constant associated with CFF must be much greater than the switching period: EQUATION 5-21: 1 -=T ------------------ « 1 f SW If the voltage divider resistors R1 and R2 are in the kΩ range, a CFF of 1 nF to 100 nF can easily satisfy the large time constant requirements. Also, a 100 nF injection capacitor CINJ is used in order to be considered as short for a wide range of the frequencies. The process of sizing the ripple injection resistor and capacitors is as follows. FIGURE 5-5: Inadequate Ripple at FB. • Virtually no ripple at the FB pin voltage due to the very low ESR of the output capacitors. In this situation, the output voltage ripple is less than 20 mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor RINJ and a capacitor CINJ, as shown in Figure 5-6. • Select CFF to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of CFF is 1 nF to 100 nF if R1 and R2 are in the kΩ range. • Select RINJ according to the expected feedback voltage ripple using Equation 5-22: EQUATION 5-22: f SW V FB PP K div = ----------------------- ---------------------------V IN D 1 – D The value of RINJ is calculated using Equation 5-23. EQUATION 5-23: 1 - – 1 R INJ = R1//R2 --------K div FIGURE 5-6: Invisible Ripple at FB. The injected ripple is calculated via: • Select CINJ as 100 nF, which could be considered as short for a wide range of the frequencies. EQUATION 5-19: 1 V FB PP = V IN K div D 1 – D ----------------f SW Where: VIN Power stage input voltage D Duty cycle fSW Switching frequency τ (R1//R2//RINJ) x CFF EQUATION 5-20: R1//R2 K div = ---------------------------------R INJ + R1//R2 2016 Microchip Technology Inc. DS20005524A-page 25 MIC28512 6.0 PCB LAYOUT GUIDELINES PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. Figure 6-1 is optimized from a small form-factor point of view and shows the top and bottom layers of a four-layer PCB. It is recommended to use Mid-Layer 1 as a continuous ground plane. operating voltage must be derated by 50%. • In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. 6.3 • Do not route any digital lines underneath or close to the SW node. • Keep the switch node (SW) away from the feedback (FB) pin. 6.4 FIGURE 6-1: Top and Bottom Layers of a Four-Layer Board. The following guidelines should be followed to ensure proper operation of the MIC28512 converter. 6.1 IC • The analog ground pin (AGND) must be connected directly to the ground planes. Do not route the AGND pin to the PGND pin on the top layer. • Place the IC close to the point-of-load (POL). • Use copper planes to route the input and output power lines. • Analog and power grounds should be kept separate and connected at only one location. 6.2 Input Capacitor • Place the input capacitors on the same side of the board and as close to the PVIN and PGND pins as possible. • Place several vias to the ground plane close to the input capacitor ground terminal. • Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. • Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. • If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the SW Node Output Capacitor • Use a copper island to connect the output capacitor ground terminal to the input capacitor ground terminal. • Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. • The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. 6.5 Thermal Measurements Measuring the IC’s case temperature is recommended to ensure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get erroneous results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. Two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire or higher then (smaller wire size) to minimize the wire heat-sinking effect. In addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ICs. However, a IR thermometer from Optris has a 1 mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time. For more information about the Evaluation board layout, please contact Microchip sales. DS20005524A-page 26 2016 Microchip Technology Inc. MIC28512 7.0 PACKAGING INFORMATION 24-Lead FQFN 3 mm x 4 mm Package Outline and Recommended Land Pattern Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging 2016 Microchip Technology Inc. DS20005524A-page 27 MIC28512 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging DS20005524A-page 28 2016 Microchip Technology Inc. MIC28512 APPENDIX A: REVISION HISTORY Revision A (April 2016) • Converted Micrel document MIC28512 to Microchip data sheet template DS20005520A. • Minor text changes throughout. 2015 Microchip Technology Inc. DS20005524A-page 29 MIC28512 NOTES: DS20005524A-page 30 2015 Microchip Technology Inc. MIC28512 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office. PART NO. Device Device: X X Architecture Temperature MIC28512: = = HyperLight Load Hyper Speed Control Temperature: Y = –40°C to +125°C Package: FL = Examples: a) MIC28512-1YFL: Package 70V, 2A Synchronous Buck Regulator 1 2 Architecture: XX Junction Temperature Range, 24LD FQFN b) MIC28512-2YFL: 2015 Microchip Technology Inc. 60VIN, 2A Synchronous Buck Regulator, Hyper Speed Control, –40°C to +125°C Junction Temperature Range, 24LD FQFN Note 1: 24-Pin 3 mm x 4 mm FQFN; Note 1 70VIN, 2A Synchronous Buck Regulator, HyperLight Load, –40°C to +125°C FQFN is a lead-free package. Pb-Free lead finish is Matte Tin. DS20005524A-page 31 MIC28512 NOTES: DS20005524A-page 32 2015 Microchip Technology Inc. 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SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. Microchip received ISO/TS-16949:2009 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. QUALITYMANAGEMENTSYSTEM CERTIFIEDBYDNV == ISO/TS16949== 2016 Microchip Technology Inc. Silicon Storage Technology is a registered trademark of Microchip Technology Inc. in other countries. GestIC is a registered trademarks of Microchip Technology Germany II GmbH & Co. KG, a subsidiary of Microchip Technology Inc., in other countries. 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