2A Synchronous Buck Regulator with Integrated MOSFETs ISL8502A Features The ISL8502A is a synchronous buck controller with internal MOSFETs packaged in a small 4mmx4mm QFN package. The ISL8502A can support a continuous load of 2A and has a very wide input voltage range. With the switching MOSFETs integrated into the IC, the complete regulator footprint can be very small and provide a much more efficient solution than a linear regulator. • Up to 2A Continuous Output Current The ISL8502A is capable of stand-alone operation or it can be used in a master slave combination for multiple outputs that are derived from the same input rail. Multiple slave channels (up to six) can be synchronized. This method minimizes the EMI and beat frequencies effect with multi-channel operation. The switching PWM controller drives two internal N-Channel MOSFETs in a synchronous-rectified buck converter topology. The synchronous buck converter uses voltage-mode control with fast transient response. The switching regulator provides a maximum static regulation tolerance of ±1% over line, load, and temperature ranges. The output is user-adjustable by means of external resistors down to 0.6V. The output is monitored for undervoltage events. The switching regulator also has overcurrent protection. Thermal shutdown is integrated. The ISL8502A features a bi-directional Enable pin that allows the part to pull the enable pin low during fault detection. PGOOD delay for ISL8502A has been decreased to 1ms typical (at 500kHz switching frequency) compared to 250ms (at 500kHz) for ISL8502. • Integrated MOSFETs for Small Regulator Footprint • Adjustable Switching Frequency, 500kHz to 1.2MHz • Tight Output Voltage Regulation, ±1% Over-temperature • Wide Input Voltage Range, 5V ±10% or 5.5V to 14V • Wide Output Voltage Range, from 0.6V • Simple Single-Loop Voltage-Mode PWM Control Design • Input Voltage Feed-Forward for Constant Modulator Gain • Fast PWM Converter Transient Response • Lossless rDS(ON) High Side and Low Side Overcurrent Protections • Undervoltage Detection • Integrated Thermal Shutdown Protection • Power-Good Indication • Adjustable Soft-Start • Start-Up with Pre-Bias Output • Pb-free (RoHS Compliant) Applications • Point of Load Applications • Graphics Cards - GPU and Memory Supplies • ASIC Power Supplies • Embedded Processor and I/O Supplies • DSP Supplies VIN 4.5V TO 5.5V POWER GOOD ENABLE PGOOD EN SYNCH M/S VCC PVCC SS VIN + VIN 5.5V TO 14V BOOT ISL8502A PHASE VOUT + PGND FS POWER GOOD ENABLE PGOOD EN PVCC VCC SS VIN BOOT ISL8502A PHASE SYNCH M/S FS FB SGND COMP FIGURE 1. STAND-ALONE REGULATOR: VIN 5.5V TO 14V October 21, 2011 FN7940.0 1 VOUT + PGND FB SGND + COMP FIGURE 2. STAND-ALONE REGULATOR: V IN 4.5V TO 5.5V CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2011. All Rights Reserved Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries. All other trademarks mentioned are the property of their respective owners. Block Diagram VCC PVCC SS PGOOD VIN (x4) VIN OC MONITOR PVCC 2 SERIES REGULATOR 30μA POR MONITOR BIAS SGND PVCC BOOT EN VOLTAGE MONITOR SYNCH M/S FS GATE DRIVE AND ADAPTIVE SHOOT THRU PROTECTION PHASE (x4) CLOCK AND OSCILLATOR GENERATOR OC MONITOR 0.6V REFERENCE FB COMP PGND (x4) ISL8502A FAULT MONITORING FN7940.0 October 21, 2011 ISL8502A Pin Configuration* VCC PVCC BOOT VIN VIN VIN ISL8502A (24 LD QFN) TOP VIEW 24 23 22 21 20 19 PGOOD 1 18 VIN SGND 2 17 PHASE EN 3 SYNCH 4 M/S 5 14 PHASE FS 6 13 PGND 16 PHASE 10 11 12 PGND PGND FB 9 PGND 8 15 PHASE SS 7 COMP GND 25 *See “Functional Pin Descriptions” beginning on page 13 for pin descriptions. Ordering Information PART NUMBER (Note 2) PART MARKING ISL8502AIRZ (Notes 1, 3) 85 02AIRZ ISL8502AEVAL1Z Evaluation Board TEMP. RANGE (°C) -40 to +85 PACKAGE (Pb-free) 24 Ld 4x4 QFN PKG. DWG. # L24.4x4D NOTES: 1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL8502A. For more information on MSL please see Tech Brief TB363. 3 FN7940.0 October 21, 2011 ISL8502A Typical Application Schematics POWER GOOD PGOOD ENABLE VIN + EN VIN 5.5V TO 14V SYNCH BOOT M/S VCC PVCC ISL8502A VOUT PHASE SS + PGND FS FB COMP SGND FIGURE 3. STAND-ALONE REGULATOR: VIN 5.5V TO 14V VIN 4.5V TO 5.5V POWER GOOD PGOOD ENABLE VIN EN + PVCC BOOT VCC SS ISL8502A VOUT PHASE + SYNCH PGND M/S FS FB SGND COMP FIGURE 4. STAND-ALONE REGULATOR: VIN 4.5V TO 5.5V 4 FN7940.0 October 21, 2011 ISL8502A ISL8502A With Multiple Slaved Channels VIN MASTER M/S SS PVCC VIN FS SYNCH RT EN VOUT1 PHASE + GND ISL8502A ENABLE M/S VIN FS 5k RT VOUT2 SYNCH EN PHASE + GND ISL8502A SLAVE M/S VIN FS 5k RT VOUTN SYNCH EN PHASE + GND ISL8502A SLAVE 5 FN7940.0 October 21, 2011 ISL8502A Absolute Maximum Ratings Thermal Information VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +16.5V VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +6.0V Absolute Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +22.0V Upper Driver Supply Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . +6.0V All other Pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to VCC + 0.3V Thermal Resistance θJA (°C/W) θJC (°C/W) QFN Package (Notes 4, 5) . . . . . . . . . . . . . 38 2 Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . +150°C Maximum Storage Temperature Range . . . . . . . . . . . . . -65°C to +150°C Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions Supply Voltage on VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5V to 14V Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . -40°C to +125°C CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379. 5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications Refer to “Block Diagram”and “Typical Application Schematics”. Operating conditions unless otherwise noted: VIN = 12V, or VCC = 5V ±10%, TA = -40°C to +85°C. Typical are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +85°C PARAMETER SYMBOL TEST CONDITIONS MIN (Note 6) TYP MAX (Note 6) UNITS VIN SUPPLY Input Voltage Range VIN VIN tied to VCC Input Operating Supply Current IQ 5.5 (Note 7) 14 (Note 8) 4.5 5.5 V 7 mA 1.25 2 mA 5.0 5.5 VFB = 1.0V IQ_SBY EN tied to GND, VIN = 14V VCC Voltage VPVCC VIN > 5.6V 4.5 Maximum Output Current IPVCC VIN = 12V 50 Input Standby Supply Current V SERIES REGULATOR VIN = 12V, VCC shorted to PGND VCC Current Limit V mA 300 mA POWER-ON RESET Rising VCC POR Threshold 4.2 4.4 4.49 V Falling VCC POR Threshold 3.85 4.0 4.10 V ENABLE Rising Enable Threshold Voltage VEN_Rising 2.7 V Falling Enable Threshold Voltage VEN_Fall 2.3 V Enable Sinking Current 500 IEN µA OSCILLATOR PWM Frequency fOSC Ramp Amplitude Ramp Amplitude 400 500 600 kHz RT = 40kΩ 960 1200 1440 kHz FS pin tied to VCC 800 kHz ΔVOSC VIN = 14V 1.0 V ΔVOSC VIN = 5V 0.470 V 8 - VVIN/ΔVOSC Modulator Gain RT = 96kΩ By Design Maximum Duty Cycle DMAX fOSC = 500kHz 88 % Maximum Duty Cycle DMAX fOSC = 1.2MHz 76 % REFERENCE VOLTAGE Reference Voltage VREF 6 0.600 V FN7940.0 October 21, 2011 ISL8502A Electrical Specifications Refer to “Block Diagram”and “Typical Application Schematics”. Operating conditions unless otherwise noted: VIN = 12V, or VCC = 5V ±10%, TA = -40°C to +85°C. Typical are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +85°C PARAMETER SYMBOL TEST CONDITIONS System Accuracy MIN (Note 6) TYP MAX (Note 6) +1.0 % ±80 ±200 nA 20 30 40 µA 0.8 1.0 1.2 -1.0 FB Pin Bias Current UNITS SOFT-START Soft-Start Current ISS Enable Soft-Start Threshold Enable Soft-Start Threshold Hysteresis 12 Enable Soft-Start Voltage High 2.8 3.2 V mV 3.8 V ERROR AMPLIFIER DC Gain Gain-Bandwidth Product GBWP Maximum Output Voltage 3.9 Slew Rate SR 88 dB 15 MHz 4.4 V 5 V/µs INTERNAL MOSFETS Upper MOSFET rDS(ON) rDS_UPPER VCC = 5V 180 mΩ Lower MOSFET rDS(ON) rDS_LOWER VCC = 5V 90 mΩ PGOOD PGOOD Threshold VFB/VREF PGOOD Rising Delay (Note 11) tPGOOD_DELAY Rising Edge Hysteresis 1% 107 111 115 Falling Edge Hysteresis 1% 86 90 93 fOSC = 500kHz 1 VPGOOD = 5.5V PGOOD Leakage Current PGOOD Low Voltage VPGOOD PGOOD Sinking Current IPGOOD % % ms 5 0.10 µA V 0.5 mA PROTECTION Positive Current Limit IPOC_peak Negative Current Limit INOC_peak IOC from VIN to PHASE (Notes 9, 10) (TA = 0°C to +85°C) 2.1 3.5 4.5 A IOC from VIN to PHASE (Notes 9, 10) (TA = -40°C to +0°C) 2.0 3.4 4.0 A IOC from PHASE to PGND (Notes 9, 10) (TA = 0°C to +85°C) 2.2 3.0 3.5 A IOC from PHASE to PGND (Notes 9, 10) (TA = -40°C to +85°C) 1.9 2.8 3.7 A 76 80 84 % VFB/VREF Undervoltage Level Thermal Shutdown Setpoint TSD 150 °C Thermal Recovery Setpoint TSR 130 °C NOTES: 6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. 7. Minimum VIN can operate below 5.5V as long as VCC is greater than 4.5V. 8. Maximum VIN can be higher than 14V voltage stress across the upper and lower do not exceed 15.5V in all conditions. 9. Circuit requires 150ns minimum on time to detect overcurrent condition. 10. Limits established by characterization and are not production tested. 11. PGOOD Rising Delay is measured from the point where VOUT reaches regulation to the point where PGOOD rises. It does not include the external soft-start time. The PGOOD Rising Delay specification is measured at 500kHz. 7 FN7940.0 October 21, 2011 ISL8502A Typical Performance Curves VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7µH, CIN = 20µF, 100 100 90 90 80 80 EFFICIENCY (%) EFFICIENCY (%) COUT = 100µF + 22µF, TA = +25° C, unless otherwise noted. VOUT = 2.5V 70 VOUT = 1.8V VOUT = 3.3V 60 50 VOUT = 5.0V 70 VOUT = 3.3V VOUT = 2.5V 60 VOUT = 1.8V 50 40 0.0 0.5 1.0 1.5 2.0 40 0.0 2.5 0.5 OUTPUT LOAD (A) 1.0 1.5 2.0 2.5 OUTPUT LOAD (A) FIGURE 5. EFFICIENCY vs LOAD (V IN = 5V) FIGURE 6. EFFICIENCY vs LOAD (VIN = 12V) 0.6026 1.206 14VIN 1.205 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 0.6025 0.6024 0.6023 14VIN 0.6022 0.6021 9VIN 0.6020 0.6019 9VIN 1.203 1.202 5VIN 1.201 5VIN 0 1.204 1 OUTPUT LOAD (A) 1.200 2 FIGURE 7. VOUT REGULATION vs LOAD (VOUT = 0.6V, 500kHz) 0 1 OUTPUT LOAD (A) 2 FIGURE 8. VOUT REGULATION vs LOAD (VOUT = 1.2V, 500kHz) 1.520 1.815 1.815 5VIN 1.814 5VIN 1.516 1.514 9VIN 14VIN 1.512 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 1.518 1.814 1.813 1.813 14VIN 1.812 9VIN 1.812 1.811 1.811 1.510 0 1 OUTPUT LOAD (A) 2 FIGURE 9. VOUT REGULATION vs LOAD (VOUT = 1.5V, 500kHz) 8 1.810 0 1 OUTPUT LOAD (A) 2 FIGURE 10. VOUT REGULATION vs LOAD (VOUT = 1.8V, 500kHz) FN7940.0 October 21, 2011 ISL8502A Typical Performance Curves VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7µH, CIN = 20µF, COUT = 100µF + 22µF, TA = +25° C, unless otherwise noted. (Continued) 3.355 2.515 3.354 3.353 OUTPUT VOLTAGE (V) 2.513 OUTPUT VOLTAGE (V) 14VIN 2.511 2.509 5VIN 9VIN 3.352 5VIN 3.351 14VIN 3.350 3.349 3.348 3.347 2.507 9VIN 3.346 2.505 0 1 OUTPUT LOAD (A) 3.345 2 0 FIGURE 11. VOUT REGULATION vs LOAD (VOUT = 2.5V, 500kHz) 1 OUTPUT LOAD (A) 2 FIGURE 12. VOUT REGULATION vs LOAD (VOUT = 3.3V, 500kHz) 5.030 2.0 1.8 1.6 POWER DISSIPATION (W) OUTPUT VOLTAGE (V) 5.028 7VIN 5.026 5.024 5.022 14VIN 1.2 1.0 0.8 0.6 5VIN 0.4 14VIN 9VIN 0.2 9VIN 5.020 0 1.4 1 OUTPUT LOAD (A) 0.0 0 2 FIGURE 13. VOUT REGULATION vs LOAD (VOUT = 5V, 500kHz) 1 OUTPUT LOAD (A) 2 FIGURE 14. POWER DISSIPATION vs LOAD (VOUT = 0.6V, 500kHz) 2.5 2.0 1.8 2.0 1.4 1.2 1.0 14VIN 0.8 0.6 5VIN 0.4 POWER DISSIPATION (W) POWER DISSIPATION (W) 1.6 1.5 1.0 14VIN 0.5 5VIN 9VIN 0.2 9VIN 0.0 0.0 0 1 OUTPUT LOAD (A) 2 FIGURE 15. POWER DISSIPATION vs LOAD (VOUT = 1.2V, 500kHz) 9 0 1 2 OUTPUT LOAD (A) FIGURE 16. POWER DISSIPATION vs LOAD (VOUT = 1.5V, 500kHz) FN7940.0 October 21, 2011 ISL8502A Typical Performance Curves VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7µH, CIN = 20µF, 2.5 2.5 2.0 2.0 POWER DISSIPATION (W) POWER DISSIPATION (W) COUT = 100µF + 22µF, TA = +25° C, unless otherwise noted. (Continued) 1.5 14VIN 1.0 0.5 5VIN 0 0.5 1 OUTPUT LOAD (A) 0 2 1 2 OUTPUT LOAD (A) FIGURE 18. POWER DISSIPATION vs LOAD (VOUT = 2.5V, 500kHz) 2.5 2.0 2.0 POWER DISSIPATION (W) 2.5 14VIN 1.5 5VIN 0.0 FIGURE 17. POWER DISSIPATION vs LOAD (VOUT = 1.8V, 500kHz) POWER DISSIPATION (W) 14VIN 1.0 9VIN 9VIN 0.0 1.5 1.0 0.5 14VIN 1.5 1.0 0.5 7VIN 5VIN 9VIN 0.0 9VIN 0.0 0 1 2 0 1 2 OUTPUT LOAD (A) OUTPUT LOAD (A) FIGURE 20. POWER DISSIPATION vs LOAD (VOUT = 5V, 500kHz) FIGURE 19. POWER DISSIPATION vs LOAD (VOUT = 3.3V, 500kHz) 5.5 5.2 5.4 NO LOAD 5.1 5.3 5.2 VCC (V) VCC (V) 5.0 4.9 4.8 5.1 5.0 4.9 100mA LOAD 4.8 4.7 4.7 4.6 4.5 4.6 4.5 0 50 100 150 I VCC (mA) 200 FIGURE 21. VCC LOAD REGULATION 10 250 300 3 4 5 6 7 8 9 10 VIN (V) 11 12 13 14 15 FIGURE 22. VCC REGULATION vs VIN FN7940.0 October 21, 2011 ISL8502A Typical Performance Curves VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7µH, CIN = 20µF, COUT = 100µF + 22µF, TA = +25° C, unless otherwise noted. (Continued) PHASE1 5V/DIV PHASE1 0.5µs 5V 5V/DIV PHASE2 5V/DIV VOUT1 RIPPLE 20mV/DIV VOUT2 RIPPLE 20mV/DIV VOUT1 RIPPLE 20mV/DIV IL1 0.5A/DIV SYNCH1 2V/DIV FIGURE 23. MASTER TO SLAVE OPERATION FIGURE 24. MASTER OPERATION AT NO LOAD PHASE1 10V/DIV PHASE1 5V/DIV VOUT1 RIPPLE 20mV/DIV VOUT1 RIPPLE 20mV/DIV IL1 1A/DIV IL1 1A/DIV SYNCH1 5V/DIV SYNCH1 5V/DIV FIGURE 25. MASTER OPERATION WITH FULL LOAD FIGURE 26. MASTER OPERATION WITH NEGATIVE LOAD EN1 5V/DIV EN1 5V/DIV VOUT1 1V/DIV IL1 2A/DIV VOUT1 0.5V/DIV 2V PRE-BIASED IL1 1A/DIV SS1 2V/DIV FIGURE 27. SOFT-START AT NO LOAD 11 SS1 2V/DIV FIGURE 28. START-UP WITH PRE-BIASED FN7940.0 October 21, 2011 ISL8502A Typical Performance Curves VIN = 12V, VOUT = 2.5V, IO = 2A, fs = 500kHz, L = 4.7µH, CIN = 20µF, COUT = 100µF + 22µF, TA = +25° C, unless otherwise noted. (Continued) PHASE1 10V/DIV EN1 5V/DIV VOUT1 1V/DIV VOUT1 1V/DIV IL1 1A/DIV IL1 1A/DIV PGOOD1 5V/DIV SS1 2V/DIV FIGURE 29. SOFT-START AT FULL LOAD FIGURE 30. POSITIVE OUTPUT SHORT CIRCUIT PHASE1 10V/DIV VOUT1 2V/DIV PHASE1 10V/DIV VOUT1 2V/DIV IL1 2A/DIV IL1 2A/DIV SS1 2V/DIV PGOOD1 5V/DIV FIGURE 31. POSITIVE OUTPUT SHORT CIRCUIT (HICCUP MODE) PHASE1 10V/DIV VOUT1 1V/DIV FIGURE 32. NEGATIVE OUTPUT SHORT CIRCUIT PHASE1 5V/DIV IL1 1A/DIV VOUT1 RIPPLE 50mV/DIV IL1 2A/DIV IOUT1 2A/DIV PGOOD1 5V/DIV FIGURE 33. RECOVER FROM POSITIVE SHORT CIRCUIT 12 FIGURE 34. LOAD TRANSIENT FN7940.0 October 21, 2011 ISL8502A Functional Pin Descriptions PGOOD (Pin 1) PGOOD is an open drain output that pulls to low if the output goes out of regulation or a fault is detected. PGOOD is equipped with a fixed delay upon output power-up. The PGOOD Rising Delay specification is measured at 500 kHz from the point where VOUT reaches regulation to the point where PGOOD rises. This delay is reversely proportional to the switching frequency. SGND (Pin 2) The SGND terminal of the ISL8502A provides the return path for the control and monitor portions of the IC. voltage can be set to any voltage between the power rail (reduced by converter losses) and the 0.6V reference. Loop compensation is achieved by connecting an AC network across the COMP pin and the FB pin. The FB pin is also monitored for undervoltage events. SS (Pin 9) Connect a capacitor from the SS pin to ground. This capacitor, along with an internal 30µA current source, sets the soft-start interval of the converter, tSS, as shown in Equation 2. C SS [ μF ] = 50 ⋅ t SS [ S ] (EQ. 2) PGND (Pins 10-13) EN (Pin 3) The Enable pin is a bi-directional pin. If the voltage on this pin exceeds the enable threshold voltage, the part is enabled. If a fault is detected, the EN pin is pulled low via internal circuitry for a duration of four soft-start periods. For automatic start-up, use 10kΩ to 100kΩ pull-up resistor connecting to VCC. SYNCH (Pin 4) The PGND pins are used as the ground connection of the power train. PHASE (Pins 14-17) The PHASE pins are the PHASE node connections to the inductor. These pins are connected to the source of the control MOSFET and the drain of the synchronous MOSFET. SYNCH is a bi-directional pin used to synchronize slave devices to the master device. As a master device, this pin outputs the clock signal to which the slave devices synchronize. As a slave device, this pin is an input to receive the clock signal from the master device. VIN (Pins 18-21) If configured as a slave device, the ISL8502A is disabled if there is no clock signal from the master device on the SYNCH pin. It is recommended that the DC voltage applied to the VIN pins does not exceed 14V. This recommendation allows for transient spikes and voltage ringing to occur while not exceeding Absolute Maximum Ratings. Leave this pin unconnected if the IC is used in stand-alone operation. BOOT (Pin 22) M/S (Pin 5) As a slave device, tie a 5kΩ resistor between the M/S pin and ground. As a master or a stand-alone device, tie the M/S pin directly to the VCC pin. Do not short the M/S pin to GND. FS (Pin 6) The FS pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from the FS pin to GND, the switching frequency can be programmed as desired between 500kHz and 1.2MHz as shown in Equation 1. 48000 R T [ kΩ ] = -----------------------------f OSC [ kHz ] (EQ. 1) Tying the FS pin to the VCC pin forces the switching frequency to 800kHz. Using resistors with values below 40kΩ (1.2MHz) or with values higher than 97kΩ (500kHz) may damage the ISL8502A. COMP (Pin 7) and FB (Pin 8) The switching regulator employs a single voltage control loop. The FB pin is the negative input to the voltage loop error amplifier. The output voltage is set by an external resistor divider connected to FB. With a properly selected divider, the output 13 Connect the input rail to the VIN pins. These pins are the input to the regulator as well as the source for the internal linear regulator that supplies the bias for the IC. The BOOT pin provides ground-referenced bias voltage to the upper MOSFET driver. A bootstrap circuit is used to create a voltage suitable to drive the internal N-channel MOSFET. The boot diode is included within the ISL8502A. PVCC (Pin 23) The PVCC pin is the output of the internal linear regulator that supplies the bias and gate voltage for the IC. A minimum 4.7µF decoupling capacitor is recommended. VCC (Pin 24) The VCC pin supplies the bias voltage for the IC. This pin should be tied to the PVCC pin through an RC low pass filter. A 10Ω resistor and 0.1µF capacitor are recommended. Functional Description Initialization The ISL8502A automatically initializes upon receipt of input power. The Power-On Reset (POR) function continuously monitors the voltage on the VCC pin. If the voltage on the EN pin exceeds its rising threshold, then the POR function initiates soft-start operation after the bias voltage has exceeded the POR threshold. FN7940.0 October 21, 2011 ISL8502A Stand-alone Operation The ISL8502A can be configured to function as a stand-alone single channel voltage mode synchronous buck PWM voltage regulator. The “Typical Application Schematics” on page 4 show the two configurations for stand-alone operation. The internal series linear regulator requires at least 5.5V to create the proper bias for the IC. If the input voltage is between 5.5V and 15V, simply connect the VIN pins to the input rail, and the series linear regulator creates the bias for the IC. The VCC pin should be tied to a capacitor for decoupling. If the input voltage is 5V ±10%, then tie the VIN pins and the VCC pin to the input rail. The ISL8502A uses the 5V rail as the bias. A decoupling capacitor should be placed as close as possible to the VCC pin. Overcurrent Condition Flag is set from LOW to HIGH. If, on the subsequent cycle, another overcurrent condition is detected, the OC Fault Counter is incremented. If there are eight sequential OC fault detections, the regulator is shut down under an Overcurrent Fault Condition, and the EN pin is pulled LOW. An Overcurrent Fault Condition results, with the regulator attempting to restart in hiccup mode. The delay between restarts is four soft-start periods. At the end of the fourth soft-start wait period, the fault counters are reset, the EN pin is released, and soft-start is attempted again. If the overcurrent condition goes away prior to the OC Fault Counter reaching a count of four, the Overcurrent Condition Flag is set back to LOW. If the Overcurrent Condition Flag is HIGH, the Overcurrent Fault Counter is less than four, and an undervoltage event is detected, the regulator shuts down immediately. Multi-Channel (Master/Slave) Operation UNDERVOLTAGE PROTECTION The ISL8502A can be configured to function in a multi-channel system. “ISL8502A With Multiple Slaved Channels” on page 5 shows a typical configuration for the multi-channel system. If the voltage detected on the FB pin falls 18% below the internal reference voltage, and if the overcurrent condition flag is LOW, then the regulator is shut down immediately under an Undervoltage Fault Condition, and the EN pin is pulled LOW. An Undervoltage Fault Condition results in the regulator attempting to restart in hiccup mode, with the delay between restarts being four soft-start periods. At the end of the fourth soft-start wait period, the fault counters are reset, the EN pin is released, and soft-start is attempted again. In the multi-channel system, each ISL8502A IC regulates a separate rail while sharing the same input rail. By configuring the devices in a master/slave configuration, the clocks of each IC can be synchronized. There can only be one master IC in a multi-channel system. To configure an IC as the master, the M/S pin must be shorted to the VCC pin. The SYNCH pins of all the ISL8502A controller ICs in the multi-channel system must be tied together. The frequency set resistor value (RT) used on the master device must be used on every slave device. Each slave device must have a 5kΩ resistor connecting it from M/S pin to ground. The master device and all slave devices can have their EN pins tied to an enable “bus.” Since the EN pin is bi-directional, it allows for options on how each IC is tied to the enable bus. If the EN pin of any ISL8502A is tied directly to the enable bus, then that device is capable of disabling all the other devices that have their EN pins tied directly to the enable bus. If the EN pin of an ISL8502A is tied to the enable bus through a diode (anode tied to ISL8502A EN pin, cathode tied to enable bus), then the part does not disable other devices on the enable bus if it disables itself for any reason. If the master device is disabled via the EN pin, it continues to send the clock signal from the SYNCH pin. This allows slave devices to continue operating. Fault Protection The ISL8502A monitors the output of the regulator for overcurrent and undervoltage events. The ISL8502A also provides protection from excessive junction temperatures. OVERCURRENT PROTECTION The overcurrent function protects the switching converter from a shorted output by monitoring the current flowing through both the upper and lower MOSFETs. Upon detection of any overcurrent condition, the upper MOSFET is immediately turned off and is not turned on again until the next switching cycle. Upon detection of the initial overcurrent condition, the Overcurrent Fault Counter is set to 1, and the 14 THERMAL PROTECTION If the ISL8502A IC junction temperature reaches a nominal temperature of +150°C, the regulator is disabled. The ISL8502A does not re-enable the regulator until the junction temperature drops below +130°C. SHOOT-THROUGH PROTECTION A shoot-through condition occurs when both the upper and lower MOSFETs are turned on simultaneously, effectively shorting the input voltage to ground. To protect from a shoot-through condition, the ISL8502A incorporates specialized circuitry, which ensures that the complementary MOSFETs are not ON simultaneously. Application Guidelines Operating Frequency The ISL8502A can operate at switching frequencies from 500kHz to 1.2MHz. A resistor tied from the FS pin to ground is used to program the switching frequency (Equation 3). 48000 R T [ kΩ ] = -----------------------------f OSC [ kHz ] (EQ. 3) Output Voltage Selection The output voltage of the regulator can be programmed via an external resistor divider that is used to scale the output voltage relative to the internal reference voltage and feed it back to the inverting input of the error amplifier (see Figure 36). The output voltage programming resistor, R4, depends on the value chosen for the feedback resistor and the desired output FN7940.0 October 21, 2011 ISL8502A voltage of the regulator. The value for the feedback resistor is typically between 1kΩ and 10kΩ. R 1 × 0.6V R 4 = ---------------------------------V OUT – 0.6V (EQ. 4) If the output voltage desired is 0.6V, then R4 is left unpopulated. Output Capacitor Selection An output capacitor is required to filter the inductor current and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (Effective Series Resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. The shape of the output voltage waveform during a load transient that represents the worst-case loading conditions ultimately determines the number of output capacitors and their type. When this load transient is applied to the converter, most of the energy required by the load is initially delivered from the output capacitors. This is due to the finite amount of time required for the inductor current to slew up to the level of the output current required by the load. This phenomenon results in a temporary dip in the output voltage. At the very edge of the transient, the Equivalent Series Inductance (ESL) of each capacitor induces a spike that adds on top of the existing voltage drop due to the Equivalent Series Resistance (ESR). VOUT During removal of the same output load, the energy stored in the inductor is dumped into the output capacitors. This energy dumping creates a temporary hump in the output voltage. This hump, as with the sag, can be attributed to the total amount of capacitance on the output. Figure 35 shows a typical response to a load transient. The amplitudes of the different types of voltage excursions can be approximated using Equation 5. dI tran ΔV ESL = ESL • --------------dt ΔV ESR = ESR • I tran 2 L out • I tran ΔV SAG = -------------------------------------------------C out • ( V in – V out ) 2 L out • I tran ΔV HUMP = -------------------------------C out • V out (EQ. 5) where: Itran = Output Load Current Transient, and Cout = Total Output Capacitance. In a typical converter design, the ESR of the output capacitor bank dominates the transient response. The ESR and ESL typically are the major contributing factors in determining the output capacitance. The number of output capacitors can be determined by using Equation 6, which relates the ESR and ESL of the capacitors to the transient load step and the voltage limit (DVo): ESL • dI tran --------------------------------+ ESR • I tran dt Number of Capacitors = ----------------------------------------------------------------------ΔV o If DVSAG or DVHUMP is found to be too large for the output voltage limits, then the amount of capacitance may need to be increased. In this situation, a trade-off between output inductance and output capacitance may be necessary. The ESL of the capacitors, which is an important parameter in the previous equations, is not usually listed in databooks. Practically, it can be approximated using Equation 7 if an Impedance vs Frequency curve is given for a specific capacitor: 1 ESL = ---------------------------------------2 C ( 2 • π • f res ) DVHUMP DVSAG The ESL of the capacitors becomes a concern when designing circuits that supply power to loads with high rates of change in the current. DVESL ITRAN FIGURE 35. TYPICAL TRANSIENT RESPONSE After the initial spike, attributable to the ESR and ESL of the capacitors, the output voltage experiences sag. This sag is a direct consequence of the amount of capacitance on the output. 15 (EQ. 7) where fres is the frequency at which the lowest impedance is achieved (resonant frequency). DVESR IOUT (EQ. 6) Output Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and to minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current, and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by using Equation 8: DI = VIN - VOUT Fs x L x VOUT VIN DVOUT = DI x ESR (EQ. 8) FN7940.0 October 21, 2011 ISL8502A Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter response time to a load transient. One of the parameters limiting converter response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the ISL8502A provides either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval, the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. Equation 9 gives the approximate response time interval for application and removal of a transient load: tRISE = L x ITRAN VIN - VOUT tFALL = Figure 36 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the reference voltage level. The error amplifier output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). The modulator transfer function is the small-signal transfer function of VOUT/VE/A . This function is dominated by a DC gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR . The DC gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage, DVOSC . The ISL8502A incorporates a feed-forward loop that accounts for changes in the input voltage. This configuration maintains a constant modulator gain. L x ITRAN VOUT (EQ. 9) where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. The worst-case response time can be either at the application or removal of load. Be sure to check both of these equations at the minimum and maximum output levels for the worst-case response time. VIN DRIVER OSC PWM COMPARATOR LO - ΔVOSC DRIVER + PHASE VOUT CO ESR (PARASITIC) ZFB VE/A Input Capacitor Selection ZIN - + Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high-frequency decoupling, and bulk capacitors to supply the current needed each time the upper MOSFET turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of the upper MOSFET and the source of the lower MOSFET. The important parameters for bulk input capacitance are the voltage rating and the RMS current rating. For reliable operation, select bulk capacitors with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. Their voltage rating should be at least 1.25x greater than the maximum input voltage, while a voltage rating of 1.5x is a conservative guideline. For most cases, the RMS current rating requirement for the input capacitor of a buck regulator is approximately one-half the DC load current. The maximum RMS current through the input capacitors can be closely approximated using Equation 10: ⎞ 2⎞ V OUT V OUT ⎛ 2 ⎛ 1 ⎛ V IN – V OUT V OUT -------------- × ⎜ I OUT × 1 – -----------⎞ + ------ × ⎜ ----------------------------- × --------------⎟ ⎟ ⎝ ⎠ V IN V V IN ⎠ ⎠ 12 MAX ⎝ ⎝ L × f OSC IN (EQ. 10) For a through-hole design, several electrolytic capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge current at power-up. Some capacitor series available from reputable manufacturers are surge current tested. 16 Feedback Compensation ERROR AMP REFERENCE DETAILED COMPENSATION COMPONENTS ZFB C1 C2 VOUT ZIN C3 R2 R3 R1 COMP + FB R4 ISL8502A REFERENCE R ⎞ ⎛ V OUT = 0.6 × ⎜ 1 + ------1-⎟ R 4⎠ ⎝ FIGURE 36. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN AND OUTPUT VOLTAGE SELECTION Modulator Break Frequency Equations 1 f LC = ------------------------------------------2π x L O x C O 1 f ESR = -------------------------------------------2π x ESR x C O (EQ. 11) The compensation network consists of the error amplifier (internal to the ISL8502A) and the impedance networks, ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency FN7940.0 October 21, 2011 ISL8502A 1. Pick Gain (R2/R1) for desired converter bandwidth. 2. Place first zero below filter’s double pole (~75% FLC). 100 fP1 fP2 OPEN LOOP ERROR AMP GAIN 60 40 20 20LOG (R2/R1) 20LOG (VIN/ΔVOSC) 0 3. Place second zero at filter’s double pole. -20 4. Place first pole at ESR Zero. -40 5. Place second pole at half the switching frequency. -60 6. Check gain against error amplifier’s open-loop gain. fZ1 fZ2 80 GAIN (dB) (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. Equation 12 relates the compensation network’s poles, zeros, and gain to the components (R1 , R2 , R3 , C1 , C2 and C3) in Figure 36. Use these guidelines for locating the poles and zeros of the compensation network: COMPENSATION GAIN CLOSED LOOP GAIN MODULATOR GAIN fLC 10 100 1k fESR 10k 100k FREQUENCY (Hz) 1M 10M FIGURE 37. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN 7. Estimate phase margin; repeat if necessary. Compensation Break Frequency Equations Layout Considerations 1 f Z1 = -----------------------------------2π x R 2 x C 1 1 f P1 = --------------------------------------------------------⎛ C 1 x C 2⎞ 2π x R 2 x ⎜ ----------------------⎟ ⎝ C1 + C2 ⎠ 1 f Z2 = ------------------------------------------------------2π x ( R 1 + R 3 ) x C 3 1 f P2 = -----------------------------------2π x R 3 x C 3 Layout is very important in high frequency switching converter design. With power devices switching efficiently between 500kHz and 1.2MHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device overvoltage stress. Careful component layout and printed circuit board design minimize these voltage spikes. (EQ. 12) Figure 37 shows an asymptotic plot of the DC/DC converter gain vs frequency. The actual modulator gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 37. Using the guidelines provided should give a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The closed loop gain is constructed on the graph of Figure 37 by adding the modulator gain (in dB) to the compensation gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks, ZFB and ZIN, to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than +45°. Include worst-case component variations when determining phase margin. A more detailed explanation of voltage mode control of a buck regulator can be found in Tech Brief TB417, entitled “Designing Stable Compensation Networks for Single Phase Voltage Mode Buck Regulators.” As an example, consider the turn-off transition of the control MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the lower MOSFET. Any parasitic inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimize the magnitude of voltage spikes. There are two sets of critical components in the ISL8502A switching converter. The switching components are the most critical because they switch large amounts of energy and therefore tend to generate large amounts of noise. Next are the small signal components, which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 38 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer (usually a middle layer of the PC board) for a ground plane, and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane, and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power nodes. Use copper-filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the GATE pins to the MOSFET gates should be kept short and wide enough to easily handle the 1A of drive current. In order to dissipate heat generated by the internal VTT LDO, the ground pad, pin 29, should be connected to the internal ground plane through at least five vias. This allows heat to move away 17 FN7940.0 October 21, 2011 ISL8502A from the IC and also ties the pad to the ground plane through a low impedance path. VIN CIN CBP1 ISL8502A L1 RBP VCC CBP2 VOUT1 PHASE PGND LOAD The switching components should be placed close to the ISL8502A first. Minimize the length of connections between the input capacitors, CIN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper and lower MOSFETs and the load. Make the PGND and the output capacitors as short as possible. VIN PVCC 5V COUT1 COMP C2 The critical small signal components include any bypass capacitors, feedback components, and compensation components. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors should be located as close as possible to the FB pin, with vias tied straight to the ground plane as required. C1 R2 R1 FB R4 C3 R3 GND PAD KEY ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT AND/OR POWER PLANE LAYER VIA CONNECTION TO GROUND PLANE FIGURE 38. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS 18 FN7940.0 October 21, 2011 ISL8502A Revision History The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you have the latest revision. DATE REVISION 10/21/2011 FN7940.0 CHANGE Initial Release Products Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks. Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a complete list of Intersil product families. For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page on intersil.com: ISL8502A To report errors or suggestions for this datasheet, please go to: www.intersil.com/askourstaff FITs are available from our website at: http://rel.intersil.com/reports/search.php For additional products, see www.intersil.com/product_tree Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 19 FN7940.0 October 21, 2011 ISL8502A Package Outline Drawing L24.4x4D 24 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 2, 10/06 4X 2.5 4.00 A 20X 0.50 B PIN 1 INDEX AREA PIN #1 CORNER (C 0 . 25) 24 19 1 4.00 18 2 . 50 ± 0 . 15 13 0.15 (4X) 12 7 0.10 M C A B 0 . 07 24X 0 . 23 +- 0 . 05 4 24X 0 . 4 ± 0 . 1 TOP VIEW BOTTOM VIEW SEE DETAIL "X" 0.10 C C 0 . 90 ± 0 . 1 BASE PLANE ( 3 . 8 TYP ) SEATING PLANE 0.08 C SIDE VIEW ( 2 . 50 ) ( 20X 0 . 5 ) C 0 . 2 REF 5 ( 24X 0 . 25 ) 0 . 00 MIN. 0 . 05 MAX. ( 24X 0 . 6 ) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 indentifier may be either a mold or mark feature. 20 FN7940.0 October 21, 2011