Wide VIN Dual Standard Buck Regulator With 3A/3A Continuous Output Current ISL85033 Features The ISL85033 is a dual standard buck regulator capable of 3A per channel continuous output current. With an input range of 4.5V to 28V, it provides a high frequency power solution for a variety of point of load applications. • Wide Input Voltage Range from 4.5V to 28V • Adjustable Output Voltage with Continuous Output Current up to 3A • Current Mode Control The PWM controller in the ISL85033 drives an internal switching N-Channel power MOSFET and requires an external Schottky diode to generate the output voltage. The integrated power switch is optimized for excellent thermal performance up to 3A of output current. The PWM regulator switches at a default frequency of 500kHz and it can be user programmed or synchronized from 300kHz to 2MHz. The ISL85033 utilizes peak current mode control to provide flexibility in component selection and minimize solution size. The protection features include overcurrent, UVLO and thermal overload protection. • Adjustable Switching Frequency from 300kHz to 2MHz • Independent Power-Good Detection • Selectable In-Phase or Out-of-Phase PWM Operation • Independent, Sequential, Ratiometric or Absolute Tracking Between Outputs • Internal 2ms Soft-start Time • Overcurrent/Short Circuit Protection, Thermal Overload Protection, UVLO The ISL85033 is available in a small 4mmx4mm Thin Quad Flat Pb-free (TQFN) package. • Boot Undervoltage Detection Related Literature Applications • See AN1574 “ISL85033DUALEVAL1Z Wide VIN Dual Standard Buck Regulator With 3A/3A Output Current” • General Purpose Point of Load DC/DC Power Conversion • Pb-Free (RoHS Compliant) • Set-top Boxes • FPGA Power and STB Power • DVD and HDD Drives • LCD Panels, TV Power • Cable Modems 100 EFFICIENCY (%) 90 12VOUT 1MHz 80 70 60 50 40 0.0 0.5 1.0 1.5 2.0 2.5 3.0 OUTPUT LOAD (A) FIGURE 1. EFFICIENCY vs LOAD, VIN = 28V, TA = +25°C September 22, 2011 FN6676.4 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas Inc. 2010, 2011. All Rights Reserved Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries. All other trademarks mentioned are the property of their respective owners. ISL85033 Pin Configuration PGOOD1 FS NC SGND SYNCIN SYNCOUT PGOOD2 ISL85033 (28 LD TQFN) TOP VIEW 28 27 26 25 24 23 22 COMP1 1 21 COMP2 FB1 2 20 FB2 SS1 3 19 SS2 PGND1 4 BOOT1 5 17 BOOT2 PHASE1 6 16 PHASE2 PHASE1 7 15 PHASE2 18 PGND2 9 10 11 12 13 VIN1 EN1 VCC EN2 VIN2 14 VIN2 8 VIN1 PD Pin Descriptions PIN NUMBER SYMBOL 1, 21 COMP1, COMP2 2, 20 FB1, FB2 Feedback pin for the regulator. FB is the negative input to the voltage loop error amplifier. COMP is the output of the error amplifier. The output voltage is set by an external resistor divider connected to FB. In addition, the PWM regulator’s power-good and undervoltage protection circuits use FB1/2 to monitor the regulator output voltage. 3, 19 SS1, SS2 Soft-Start pins for each controller. The SS1/2 pins control the soft-start and sequence of their respective outputs. A single capacitor from the SS pin to ground determines the output ramp rate. See the “Output Tracking and Sequencing” on page 16 for soft-start and output tracking/sequencing details. If SS pins are tied to VCC, an internal soft-start of 2ms will be used. 4, 18 PGND1, PGND2 Power ground connections. Connect directly to the system GND plane. 5, 17 BOOT1, BOOT2 Floating bootstrap supply pin for the power MOSFET gate driver. The bootstrap capacitor provides the necessary charge to turn on the internal N-Channel MOSFET. Connect an external capacitor from this pin to PHASE. 6, 7, 15, 16 PHASE1, PHASE2 Switch node output. It connects the source of the internal power MOSFET with the external output inductor and with the cathode of the external diode. 8, 9, 13, 14 VIN1, VIN2 The input supply for the power stage of the PWM regulator and the source for the internal linear regulator that provides bias for the IC. Place a minimum of 10µF ceramic capacitance from each VIN to GND and close to the IC for decoupling. 10, 12 EN1, EN2 PWM controller’s enable inputs. The PWM controllers are held off when the pin is pulled to ground. When the voltage on this pin rises above 2V, the PWM controller is enabled. 11 VCC Output of the internal 5V linear regulator. Decouple to PGND with a minimum of 4.7µF ceramic capacitor. This pin is provided only for internal bias of ISL85033 (not to be loaded with current over 10mA). 23 SYNCOUT 2 PIN DESCRIPTION COMP1/COMP2 is the output of the error amplifier. Synchronization output. Provides a signal that is the inverse of the SYNCIN signal. FN6676.4 September 22, 2011 ISL85033 Pin Descriptions (Continued) PIN NUMBER SYMBOL PIN DESCRIPTION 24 SYNCIN Connect to an external signal for synchronization from 300kHz to 2MHz (negative edge trigger). SYNCIN is not allowed to be floating. When SYNCIN = logic 0, PHASE1 and PHASE2 are running at 180° out-of-phase. When SYNCIN = logic 1, PHASE1 and PHASE2 are running at 0° in-phase. When SYNCIN = an external clock, PHASE1 and PHASE2 are running at 180° out-of-phase. External SYNC frequency applied to the SYNCIN pin should be at least 2.4 times the internal switching frequency setting. 25 SGND 26 NC This is a no connection pin. 27 FS Frequency selection pin. Tie to VCC for 500kHz switching frequency. Connect a resistor to GND for adjustable frequency from 300kHz to 2MHz. 22, 28 PGOOD1, PGOOD2 - PD Signal ground connections. The exposed pad must be connected to SGND and soldered to the PCB. All voltage levels are measured with respect to this pin. Open drain power-good output that is pulled to ground when the output voltage is below regulation limits or during the soft-start interval. There is an internal 5MΩ internal pull-up resistor. The exposed pad must be connected to the system GND plane with as many vias as possible for proper electrical and thermal performance. Typical Application Schematics R6 8.06k FB2 COMP2 C4 68pF VCC VCC SS1 3 VCC PGOOD2 PGOOD1 L2 7µH VOUT2 3A R2 8.06k C5 470pF C2 470pF R8 69.8k R4 69.8k 21 20 FS 27 SS2 19 C1 68pF 1 2 8/9 VIN1 13/14 C72 ISL85033 28 PHASE2 6/7 C12 10nF C8 10nF VOUT1 3A C9 47µF D1 B340B 11 VCC 10 25 EN1 26 SGND 12 NC PGND1/2 23 SYNCOUT SYNCIN 24 4/18 BOOT2 17 L1 7µH PHASE1 5 BOOT1 EN2 D2 B340B C71 20µF VIN2 10µF 22 15/16 C13 47µF VOUT1 R1 42.2k FB1 R5 25.5k COMP1 VOUT2 4.7µF FIGURE 2. DUAL 3A OUTPUT (VIN RANGE FROM 4.5V TO 28V) 3 FN6676.4 September 22, 2011 ISL85033 Typical Application Schematics (Continued) FB2 VOUT1 R5 42.2k COMP2 C5 1nF 20 VCC FS SS2 SS1 Css1 47nF PHASE2 L2 7µH FB1 COMP1 FB2 1 2 27 8/9 19 13/14 3 PGOOD2 22 PGOOD1 28 VOUT1 6/7 15/16 10 25 C72 VOUT1 6A PHASE1 BOOT1 C8 10nF L1 7µH D1 B340B C9 47µF 11 VCC 26 SGND 12 EN1 23 NC 24 4/18 5 17 EN2 BOOT2 PGND1/2 B340B C71 20µF VIN2 10µF C12 10nF D2 VIN1 ISL85033 SYNCIN C13 47µF 21 SYNCOUT Css2 47nF R8 34k COMP2 R7 0 FB2 C4 68pF R6 8.06k 4.7µF FIGURE 3. SINGLE 6A OUTPUT (VIN RANGE FROM 4.5V TO 28V) CURRENT SHARING 4 FN6676.4 September 22, 2011 ISL85033 BOOT2 COMP2 FB2 PGOOD2 Functional Block Diagram VCC 5MΩ BOOT UV DETECTION + - VCC -10% SOFT-START CONTROL VOLTAGE MONITOR VIN2 CSA2 + - SS2 EA + - COMP2 0.8V REFERENCE FAULT MONITOR EN2 GATE DRIVE CSA2 VIN1 LDO VCC = 5V BOOT REFRESH CONTROL PGND2 + SLOPE COMP POWER-ON RESET MONITOR PHASE2 CSA2 VIN1 THERMAL MONITOR +150°C SYNCOUT CSA1 FS OSCILLATOR SYNCIN CSA1 + SLOPE COMP VIN1 CSA1 EN1 FAULT MONITOR 0.8V REFERENCE DRIVE GATE COMP1 + CONTROL SOFT-START -10% PGND1 BOOT UV DETECTION BOOT1 SGND COMP1 FB1 VCC 5MΩ PGOOD1 VCC PHASE1 BOOT REFRESH CONTROL + SS1 VCC EA + MONITOR VOLTAGE EPAD GND 5 FN6676.4 September 22, 2011 ISL85033 Ordering Information PART NUMBER (Notes 1, 2, 3) ISL85033IRTZ PART MARKING 850 33IRTZ TEMP. RANGE (°C) -40 to +85 PACKAGE (Pb-Free) 28 Ld TQFN PKG. DWG. # L28.4x4 NOTES: 1. Add “-T*” suffix for Tape and Reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL85033. For more information on MSL please see techbrief TB363. 6 FN6676.4 September 22, 2011 ISL85033 Table of Contents Absolute Maximum Ratings ...............................................................................................................................................................8 Thermal Information ...........................................................................................................................................................................8 Recommended Operating Conditions ...............................................................................................................................................8 Electrical Specifications ....................................................................................................................................................................8 Detailed Description ........................................................................................................................................................................ 16 Operation Initialization ................................................................................................................................................................... 16 Power-On Reset and Undervoltage Lockout ....................................................................................................................................................... 16 Enable and Disable................................................................................................................................................................................................. 16 Power Good ............................................................................................................................................................................................................. 16 Output Voltage Selection....................................................................................................................................................................................... 16 Output Tracking and Sequencing.................................................................................................................................................... 16 Protection Features ..........................................................................................................................................................................17 Buck Regulator Overcurrent Protection .............................................................................................................................................................. 17 Thermal Overload Protection ................................................................................................................................................................................ 18 BOOT Undervoltage Protection ............................................................................................................................................................................. 18 Application Guidelines .................................................................................................................................................................... 18 Operating Frequency .............................................................................................................................................................................................. 18 Synchronization Control......................................................................................................................................................................................... 18 Output Inductor Selection...................................................................................................................................................................................... 18 Buck Regulator Output Capacitor Selection....................................................................................................................................................... 18 Current Sharing Configuration .............................................................................................................................................................................. 19 Input Capacitor Selection ...................................................................................................................................................................................... 19 Loop Compensation Design .................................................................................................................................................................................. 19 Theory of Compensation........................................................................................................................................................................................ 20 PWM Comparator Gain Fm ................................................................................................................................................................................... 20 Power Stage Transfer Functions .......................................................................................................................................................................... 20 Rectifier Selection .................................................................................................................................................................................................. 21 Power Derating Characteristics............................................................................................................................................................................ 22 Layout Considerations ........................................................................................................................................................................................... 22 Revision History ............................................................................................................................................................................... 24 Products............................................................................................................................................................................................ 24 Package Outline Drawing ................................................................................................................................................................ 25 7 FN6676.4 September 22, 2011 ISL85033 Absolute Maximum Ratings Thermal Information VIN1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +30V PHASE1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V BOOT1/2 to PHASE1/2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V FS to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V SYNCIN to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V FB1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +2.95V EN1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V PGOOD1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V COMP1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V VCC to GND Short Maximum Duration. . . . . . . . . . . . . . . . . . . . . . . . . . . . .1s SYNCOUT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V SS1/2 to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +5.9V ESD Rating Human Body Model (Tested per JESD22-A114) . . . . . . . . . . . . . . . . . 3kV Charged Device Model (Tested per JESD22-C101E). . . . . . . . . . . . .2.2kV Machine Model (Tested per JESD22-A115). . . . . . . . . . . . . . . . . . . . 300V Latch Up (Tested per JESD-78A; Class 2, Level A) . . . . . . . . . . . . . . 100mA Thermal Resistance θJA (°C/W) θJC (°C/W) QFN Package (Notes 4, 5) . . . . . . . . . . . . . . 38 3 Maximum Junction Temperature (Plastic Package) . . . . . . . . . . . .+150°C Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C Junction Temperature Range . . . . . . . . . . . . . . . . . . . . . . .-55°C to +150°C Operating Temperature Range . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see link below http://www.intersil.com/pbfree/Pb-FreeReflow.asp Recommended Operating Conditions Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5V to 28V CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty. NOTES: 4. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech Brief TB379 for details. 5. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside. Electrical Specifications TA = -40°C to +85°C, VIN = 4.5V to 28V, unless otherwise noted. Typical values are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +85°C PARAMETER SYMBOL TEST CONDITIONS MIN (Note 8) TYP MAX (Note 8) UNITS 28 V 2.2 mA SUPPLY VOLTAGE VIN Voltage Range VIN VIN Quiescent Supply Current IQ 4.5 1.2 VIN Shutdown Supply Current ISD EN1/2 = 0V VCC Voltage VCC VIN = 12V; IOUT = 0mA 20 45 µA 4.5 5.1 5.6 V 3.9 4.4 Falling Edge 3.2 3.7 FS = VCC 420 500 POWER-ON RESET VIN POR Threshold Rising Edge V V OSCILLATOR Nominal Switching Frequency FSW FS Voltage VFS Switching Frequency kHz 300 kHz Resistor from FS to GND = 40.2kΩ 2000 kHz FS = 100kΩ 780 SYNCIN = 600kHz 1.2MHz ≤ SYNCIN ≤ 4MHz Minimum Off-Time 580 Resistor from FS to GND = 383kΩ 800 820 300 600 tOFF mV kHz 2000 130 kHz ns ERROR AMPLIFIER Error Amplifier Transconductance Gain gm FB1, FB2 Leakage Current 125 205 285 µA/V 10 100 nA 0.18 0.21 0.24 V/A 0.792 0.8 0.808 V 1.5 2.5 3.5 ms 1.4 2 2.6 µA VFB = 0.8V Current Sense Amplifier Gain RT Reference Voltage Soft-Start Ramp Time SS1/2 = VDD Soft-Start Charging Current ISS 8 FN6676.4 September 22, 2011 ISL85033 Electrical Specifications TA = -40°C to +85°C, VIN = 4.5V to 28V, unless otherwise noted. Typical values are at TA = +25°C. Boldface limits apply over the operating temperature range, -40°C to +85°C (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN (Note 8) TYP MAX (Note 8) 91 94 82.5 85.5 % % UNITS POWER-GOOD PG1, PG2 Trip Level PG to PGOOD1, PGOOD2 Rise PG1, PG2 Propagation Delay Percentage of the soft-start time 10 PG1, PG2 Low Voltage ISINK = 3mA 100 Fall % 300 mV -1 1 µA 0.8 V Float Level 1.0 1.4 V High Level 2 ENABLE INPUT EN1, EN2 Leakage Current EN1/2 = 0V/5V EN1, EN2 Input Threshold Voltage Low Level V SYNC INPUT/OUTPUT SYNCIN Input Threshold Falling Edge SYNCIN Leakage Current 1.1 1.4 Rising Edge 1.6 Hysteresis 200 SYNCIN = 0V/5V SYNCIN Pulse Width 10 V 1.9 1000 100 SYNCOUT Phase-shift to SYNCIN Measured from rising edge to rising edge, if duty cycle is 50% SYNCOUT Frequency Range ISYNCOUT = 3mA VCC - 0.3 SYNCOUT Output Voltage Low nA ns 180 600 SYNCOUT Output Voltage High V mV ° 4000 VCC -0.08 0.08 kHz V 0.3 V FAULT PROTECTION Thermal Shutdown Temperature TSD Rising Threshold 150 °C THYS Hysteresis 20 °C Overcurrent Protection Threshold (Note 7) 4.1 OCP Blanking Time 5.1 6.1 60 A ns POWER MOSFET Highside RHDS IPHASE = 100mA 75 Internal BOOT1, BOOT2 Refresh Lowside RLDS IPHASE = 100mA 1 tRISE VIN = 25V PHASE Leakage Current EN1/2 = PHASE1/2 = 0V PHASE Rise Time 150 mΩ Ω 300 10 nA ns NOTES: 6. Test Condition: VIN = 28V, FB forced above regulation point (0.8V), no switching, and power MOSFET gate charging current not included. 7. Established by both current sense amplifier gain test and current sense amplifier output test @ IL = 0A. 8. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. 9 FN6676.4 September 22, 2011 ISL85033 Typical Performance Curves 100 100 90 90 80 EFFICIENCY (%) EFFICIENCY (%) Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. 12VOUT 1MHz 9VOUT 1MHz 70 5VOUT 500kHz 60 3.3VOUT 500kHz 5VOUT 70 3.3VOUT 60 50 50 1.8VOUT 300kHz 40 0.0 0.5 1.0 1.5 2.0 OUTPUT LOAD (A) 2.5 40 0.0 3.0 90 3.5 POWER DISSIPATION (W) 4.2 80 9VIN 12VIN 1.0 1.5 2.0 2.5 3.0 FIGURE 5. EFFICIENCY vs LOAD, TA = +25°C, FSW = 500kHz, VIN = 12V 100 70 0.5 OUTPUT LOAD (A) FIGURE 4. EFFICIENCY vs LOAD, TA = +25°C, VIN = 28V EFFICIENCY (%) 80 28VIN 60 50 2.8 2.1 12VIN 1.4 28VIN 0.7 9VIN 40 0 0.0 1 2 3 4 OUTPUT LOAD (A) 5 6 4.8 5.04 4.0 5.03 3.2 2.4 1.6 12VIN 0.8 28VIN 0.0 0 1 2 3 4 OUTPUT LOAD (A) 9VIN 5 FIGURE 8. POWER DISSIPATION vs LOAD, TA = +85°C, CURRENT SHARING 5VOUT, FSW = 500kHz 10 6 1 2 3 4 OUTPUT LOAD (A) 5 6 FIGURE 7. POWER DISSIPATION vs LOAD, TA = +25°C, CURRENT SHARING 5VOUT, FSW = 500kHz OUTPUT VOLTAGE (V) POWER DISSIPATION (W) FIGURE 6. EFFICIENCY vs LOAD, TA = +25°C, CURRENT SHARING 5VOUT, FSW = 500kHz 0 5.02 12VIN 5.01 9VIN 5.00 28VIN 4.99 4.98 0 0.5 1.0 1.5 2.0 OUTPUT LOAD (A) 2.5 3.0 FIGURE 9. VOUT REGULATION vs LOAD, CHANNEL 1, TA = +25°C, 5VOUT, FSW = 500kHz FN6676.4 September 22, 2011 ISL85033 Typical Performance Curves 5.04 3.329 5.03 3.328 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued) 5.02 5.01 5.00 9VIN 28VIN 12VIN 4.99 4.98 0 1 2 3 4 5 3.326 18VIN 3.325 3.323 3.322 3.320 0 6 28VIN 12VIN 0.5 1.0 5.02 5.03 5.01 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) 5.04 5.02 5.01 3A 2A 0A 4.99 4.98 0 5 10 15 2.0 2.5 3.0 FIGURE 11. VOUT REGULATION vs LOAD, CHANNEL 2, TA = +25°C, 3.3VOUT, FSW = 500kHz FIGURE 10. VOUT REGULATION vs LOAD, CURRENT SHARING, TA = +25°C, 5VOUT, FSW = 500kHz 5.00 1.5 OUTPUT LOAD (A) OUTPUT LOAD (A) 20 25 30 5.00 0A 4.99 4.98 4A 6A 4.97 4.96 0 5 10 15 20 25 30 INPUT VOLTAGE (V) INPUT VOLTAGE (V) FIGURE 12. OUTPUT VOLTAGE REGULATION vs VIN, CHANNEL 1, TA = +25°C, 5VOUT, FSW = 500kHz FIGURE 13. OUTPUT VOLTAGE REGULATION vs VIN, CURRENT SHARING, TA = +25°C, 5VOUT, FSW = 500kHz 3.340 OUTPUT VOLTAGE (V) 3.335 LX1 5V/DIV 3.330 3.325 VOUT1 RIPPLE 20mV/DIV 3.320 0A 3.315 3.310 0 5 10 2A 15 3A 20 IL1 0.1A/DIV 25 30 INPUT VOLTAGE (V) FIGURE 14. OUTPUT VOLTAGE REGULATION vs VIN, CHANNEL 2, TA = +25°C, 3.3VOUT, FSW = 500kHz 11 FIGURE 15. STEADY STATE OPERATION AT NO LOAD CHANNEL 1 FN6676.4 September 22, 2011 ISL85033 Typical Performance Curves Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued) LX1 5V/DIV LX2 5V/DIV VOUT1 RIPPLE 20mV/DIV VOUT2 RIPPLE 20mV/DIV IL1 0.2A/DIV IL2 0.1A/DIV FIGURE 16. STEADY STATE OPERATION AT NO LOAD CHANNEL 1 (VIN = 9V) LX1 5V/DIV FIGURE 17. STEADY STATE OPERATION AT NO LOAD CHANNEL 2 LX2 5V/DIV VOUT1 RIPPLE 20mV/DIV IL1 1A/DIV FIGURE 18. STEADY STATE OPERATION WITH FULL LOAD CHANNEL 1 VOUT2 RIPPLE 20mV/DIV IL2 1A/DIV FIGURE 19. STEADY STATE OPERATION WITH FULL LOAD CHANNEL 2 LX2 10V/DIV VOUT1 RIPPLE 20mV/DIV VOUT RIPPLE 20mV/DIV LX1 10V/DIV IL1 2A/DIV FIGURE 20. STEADY STATE OPERATION WITH FULL LOAD CURRENT SHARING 12 FIGURE 21. LOAD TRANSIENT CHANNEL 1 FN6676.4 September 22, 2011 ISL85033 Typical Performance Curves Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued) EN1 5V/DIV VOUT1 2V/DIV VOUT2 RIPPLE 20mV/DIV IL1 0.5A/DIV PG1 5V/DIV IL2 2A/DIV FIGURE 22. LOAD TRANSIENT CHANNEL 2 EN2 5V/DIV VOUT2 2V/DIV IL2 0.5A/DIV PG2 5V/DIV FIGURE 24. SOFT-START WITH NO LOAD CHANNEL 2 FIGURE 23. SOFT-START WITH NO LOAD CHANNEL 1 EN1 5V/DIV VOUT1 2V/DIV IL1 2A/DIV PG1 5V/DIV FIGURE 25. SOFT-START AT FULL LOAD CHANNEL 1 EN2 5V/DIV VOUT2 2V/DIV EN1 5V/DIV VOUT1 1V/DIV IL2 2A/DIV IL1 0.5A/DIV PG2 5V/DIV PG 5V/DIV FIGURE 26. SOFT-START AT FULL LOAD CHANNEL 2 13 FIGURE 27. SOFT-DISCHARGE SHUTDOWN CHANNEL 1 FN6676.4 September 22, 2011 ISL85033 Typical Performance Curves Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued) VOUT1 2V/DIV EN2 5V/DIV VOUT2 2V/DIV VOUT2 0.5V/DIV EN1, 2 2V/DIV IL2 0.5A/DIV PG 5V/DIV FIGURE 28. SOFT-DISCHARGE SHUTDOWN CHANNEL 2 FIGURE 29. INDEPENDENT START-UP SEQUENCING AT NO LOAD VOUT1 2V/DIV VOUT1 2V/DIV VOUT2 2V/DIV VOUT2 2V/DIV EN1, 2 2V/DIV FIGURE 30. RATIOMETRIC START-UP SEQUENCING AT NO LOAD LX1 10V/DIV EN1, 2 2V/DIV FIGURE 31. ABSOLUTE START-UP SEQUENCING AT NO LOAD LX1 10V/DIV VOUT1 RIPPLE 20mV/DIV VOUT2 RIPPLE 20mV/DIV LX2 10V/DIV LX2 10V/DIV SYNC 5V/DIV SYNC 5V/DIV FIGURE 32. STEADY STATE OPERATION CHANNEL 1 AT FULL LOAD WITH SYNC FREQUENCY = 4MHz 14 FIGURE 33. STEADY STATE OPERATION CHANNEL 2 AT FULL LOAD WITH SYNC FREQUENCY = 4MHz FN6676.4 September 22, 2011 ISL85033 Typical Performance Curves Circuit of Figure 2. VIN = 12V, VOUT1 = 5V, VOUT2 = 3.3V, IOUT1 = 3A, IOUT2 = 3A, TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued) PHASE1 10V/DIV IL1 2A/DIV VOUT1 2V/DIV PHASE1 10V/DIV VOUT1 2V/DIV IL1 2A/DIV PG1 5V/DIV PG1 5V/DIV FIGURE 34. OUTPUT SHORT CIRCUIT CHANNEL 1 PHASE2 10V/DIV IL2 2A/DIV VOUT2 2V/DIV FIGURE 35. OUTPUT SHORT CIRCUIT HICCUP AND RECOVERY FOR CHANNEL 1 PHASE2 10V/DIV VOUT2 2V/DIV IL2 2A/DIV PG2 5V/DIV PG2 5V/DIV FIGURE 36. OUTPUT SHORT CIRCUIT CHANNEL 2 15 FIGURE 37. OUTPUT SHORT CIRCUIT HICCUP AND RECOVERY FOR CHANNEL 2 FN6676.4 September 22, 2011 ISL85033 Detailed Description The ISL85033 combines a standard buck PWM controller with an integrated switching MOSFET. The buck controller drives an internal N-Channel MOSFET and requires an external diode to deliver load current up to 3A. A Schottky diode is recommended for improved efficiency and performance over a standard diode. The standard buck regulator can operate from an unregulated DC source, such as a battery, with a voltage ranging from +4.5V to +28V. The converter output can be regulated to as low as 0.8V. These features make the ISL85033 ideally suited for FPGA, set-top boxes, LCD panels, DVD drives, and wireless chipset power applications. The ISL85033 employs a peak current mode control loop which simplifies feedback loop compensation and rejects input voltage variation. External feedback loop compensation allows flexibility in output filter component selection. The regulator switches at a default 500kHz and it can be adjusted from 300kHz to 2MHz with a resistor from FS to GND. The ISL85033 is also synchronizable from 300kHz to 2MHz. forces PG low until the fault condition is cleared by attempts to soft-start. There is an internal 5MΩ internal pull-up resistor. Output Voltage Selection The regulator output voltages is easily programmed using an external resistor divider to scale VOUT relative to the internal reference voltage. The scaled voltage is applied to the inverting input of the error amplifier; refer to Figure 38. The output voltage programming resistor, R2, depends on the value chosen for the feedback resistor, R3, and the desired output voltage, VOUT, of the regulator. Equation 1 describes the relationship between VOUT and resistor values. R3 is often chosen to be in the 1kΩ to 10kΩ range. R 2 = ( V OUT – 0.8 ) • R 3 ⁄ 0.8 If the desired output voltage is 0.8V, then R3 is left unpopulated and R2 is zero ohm. VOUT Operation Initialization FB EA R2 + - The power-ON reset circuitry and enable inputs prevent false start-up of the PWM regulator output. Once all input criteria are met, the controller soft-starts the output voltage to the programmed level. (EQ. 1) R3 0.8V REFERENCE Power-On Reset and Undervoltage Lockout The ISL85033 automatically initializes upon receipt of input power supply. The power-on reset (POR) function continually monitors VIN1 voltage. While below the POR threshold, the controller inhibits switching of the internal power MOSFET. Once exceeded, the controller initializes the internal soft-start circuitry. If VIN1 supply drops below their falling POR threshold during softstart or operation, the buck regulator is disabled until the input voltage returns. Enable and Disable When EN1 and EN2 are pulled low, the device enters shutdown mode and the supply current drops to a typical value of 20µA. All internal power devices are held in a high-impedance state while in shutdown mode. The EN pin enables the controller of the ISL85033. When the voltage on the EN pin exceeds its logic rising threshold, the controller initiates the 2ms soft-start function for the PWM regulator. If the voltage on the EN pin drops below the falling threshold, the buck regulator shuts down. Power Good PG is the open-drain output of a window comparator that continuously monitors the buck regulator output voltage via the FB pin. PG is actively held low when EN is low and during the buck regulator soft-start period. After the soft-start period terminates, PG becomes high impedance as long as the output voltage (monitored on the FB pin) is above 90% of the nominal regulation voltage set by FB. When VOUT drops 10% below the nominal regulation voltage, the ISL85033 pulls PG low. Any fault condition 16 FIGURE 38. EXTERNAL RESISTOR DIVIDER Output Tracking and Sequencing Output tracking and sequencing between channels can be implemented by using the SS1 and SS2 pins. Figures 39, 40 and 41 show several configurations for output tracking/sequencing for a 2.5V and 1.8V application. Independent soft-start for each channel is shown in Figure 39 and measured in Figure 29. The output ramp-time for each channel (tSS) is set by the soft-start capacitor (CSS). C SS [ μF ] = 2.5*t SS ( s ) (EQ. 2) Ratiometric tracking is achieved in Figure 40 by using the same value for the soft-start capacitor on each channel; it is measured in Figure 30. By connecting a feedback network from VOUT1 to the SS2 pin with the same ratio that sets VOUT2 voltage, absolute tracking shown in Figure 41 is implemented. The measurement is shown in Figure 31. If the output of Channel 1 is shorted to GND, it will enter overcurrent hiccup mode, SS2 will be pulled low through the added resistor between VOUT1 and SS2 and this will force Channel 2 into hiccup as well. If the output of Channel 2 is shorted to GND with VOUT1 in regulation, it will enter overcurrent hiccup mode with a very short hiccup waiting time. The reason is that VOUT1 is still in regulation and can pull-up SS2 very quickly via the resistor added between VOUT1 and SS2. FN6676.4 September 22, 2011 ISL85033 Figure 42 illustrates output sequencing. When EN1 is high and EN2 is floating, OUT1 comes up first and OUT2 won't start until OUT1 > 90% of its regulation point. If EN1 is floating and EN2 is high, OUT2 comes up first and OUT1 won't start until OUT2 > 90% of its regulation point. If EN1 = EN2 = high, OUT1 and OUT2 come up at the same time. Please refer to Table 1 for conditions related to Figure 42 (Output Sequencing). VOUT1 SS1 C1 0.22µF SS2 ISL85033 VOUT2 TABLE 1. OUTPUT SEQUENCING EN2 VOUT1 VOUT2 High Floating First After VOUT1 > 90% Floating High After VOUT2 > 90% First High High Same time as VOUT2 Same time as VOUT1 Floating Floating C1 0.1µF NOTE R2 8.06k Not Allowed VOUT1 R1 25.5k FIGURE 41. ABSOLUTE START-UP 5.0V C3 C3 C1 0.1µF ISL85033 VOUT2 5.0V VOUT1 SS1 SS2 C2 0.22µF 3.3V C4 EN1 SS1 5.0V C3 SS2 3.3V C4 C2 0.1µF EN1 ISL85033 VOUT2 EN2 3.3V C4 FIGURE 39. INDEPENDENT START-UP FIGURE 42. OUTPUT SEQUENCING VOUT1 SS1 5.0V C3 C1 0.1µF The ISL85033 limits the current in all on-chip power devices. Overcurrent protection limits the current on the two buck regulators and internal LDO for VCC. SS2 ISL85033 VOUT2 3.3V C4 C2 0.1µF FIGURE 40. RATIOMETRIC START-UP Protection Features Buck Regulator Overcurrent Protection During the PWM on-time, the current through the internal switching MOSFET is sampled and scaled through an internal pilot device. The sampled current is compared to a nominal 5A overcurrent limit. If the sampled current exceeds the overcurrent limit reference level, an internal overcurrent fault counter is set to 1 and an internal flag is set. The internal power MOSFET is immediately turned off and will not be turned on again until the next switching cycle. The protection circuitry continues to monitor the current and turns off the internal MOSFET as described. If the overcurrent condition persists for 17 sequential clock cycles, the overcurrent fault counter overflows indicating an overcurrent fault condition exists. The regulator is shut down and power-good goes low. The buck controller attempts to recover from the overcurrent condition after waiting 8 soft-start cycles. The internal overcurrent flag and counter are reset. A normal soft-start cycle is attempted and normal operation continues if the fault 17 FN6676.4 September 22, 2011 ISL85033 condition has cleared. If the overcurrent fault counter overflows during soft-start, the converter shuts down and this hiccup mode operation repeats. Thermal Overload Protection Thermal overload protection limits maximum junction temperature in the ISL85033. When the junction temperature (TJ) exceeds +150°C, a thermal sensor sends a signal to the fault monitor. The fault monitor commands the buck regulator to shut down. When the junction temperature has decreased by 20°C, the regulator will attempt a normal soft-start sequence and return to normal operation. For continuous operation, the +125°C junction temperature rating should not be exceeded. BOOT Undervoltage Protection If the BOOT capacitor voltage falls below 2.5V, the BOOT undervoltage protection circuit will pull the phase pin low through a 1Ω switch for 400ns to recharge the capacitor. This operation may arise during long periods of no switching as in no load situations. Application Guidelines The ISL85033 operates at a default switching frequency of 500kHz if FS is tied to VCC. Tie a resistor from FS to GND to program the switching frequency from 300kHz to 2MHz, as shown in Equation 3. (EQ. 3) Where: t is the switching period in µs. 300 RFS (kΩ) The frequency of operation can be synchronized up to 2MHz by an external signal applied to the SYNCIN pin. The falling edge on the SYNCIN triggers the rising edge of PHASE1/2. The switching frequency for each output is half of the SYNCIN frequency. Output Inductor Selection The inductor value determines the converter’s ripple current. Choosing an inductor current requires a somewhat arbitrary choice of ripple current, ΔI. A reasonable starting point is 30% of total load current. The inductor value can then be calculated using Equation 4: V IN – V OUT V OUT L = ---------------------------- × ------------Fs × ΔI V IN (EQ. 4) Increasing the value of inductance reduces the ripple current and thus ripple voltage. However, the larger inductance value may reduce the converter’s response time to a load transient. The inductor current rating should be such that it will not saturate in overcurrent conditions. Buck Regulator Output Capacitor Selection An output capacitor is required to filter the inductor current. Output ripple voltage and transient response are 2 critical factors when considering output capacitance choice. The current mode control loop allows the usage of low ESR ceramic capacitors and thus smaller board layout. Electrolytic and polymer capacitors may also be used. Operating Frequency R FS [ kΩ ] = 122kΩ∗ ( t – 0.17μs ) Synchronization Control 200 Additional consideration applies to ceramic capacitors. While they offer excellent overall performance and reliability, the actual in-circuit capacitance must be considered. Ceramic capacitors are rated using large peak-to-peak voltage swings and with no DC bias. In the DC/DC converter application, these conditions do not reflect reality. As a result, the actual capacitance may be considerably lower than the advertised value. Consult the manufacturers data sheet to determine the actual in-application capacitance. Most manufacturers publish capacitance vs DC bias so that this effect can be easily accommodated. The effects of AC voltage are not frequently published, but an assumption of ~20% further reduction will generally suffice. The result of these considerations can easily result in an effective capacitance 50% lower than the rated value. Nonetheless, they are a very good choice in many applications due to their reliability and extremely low ESR. The following equations allow calculation of the required capacitance to meet a desired ripple voltage level. Additional capacitance may be used. 100 0 500 750 1000 1250 1500 FS (kHz) FIGURE 43. RFS SELECTION vs FS 1750 2000 For the ceramic capacitors (low ESR): = ΔI V OUTripple = ----------------------------------8∗ F SW∗ C OUT (EQ. 5) where ΔI is the inductor’s peak to peak ripple current, FSW is the switching frequency and COUT is the output capacitor. If using electrolytic capacitors then: V OUTripple = ΔI*ESR 18 (EQ. 6) FN6676.4 September 22, 2011 ISL85033 Regarding transient response needs, a good starting point is to determine the allowable overshoot in VOUT if the load is suddenly removed. In this case, energy stored in the inductor will be transferred to COUT causing its voltage to rise. After calculating capacitance required for both ripple and transient needs, choose the larger of the calculated values. The following equation determines the required output capacitor value in order to achieve a desired overshoot relative to the regulated voltage. If capacitors other than MLCC are used, attention must be paid to ripple and surge current ratings. I RMS ------------ = Io D – D2 (EQ. 9) where D = VO/VIN The input ripple current is graphically represented in Figure 45. 0.6 I OUT 2 * L C OUT = ------------------------------------------------------------------------------------V OUT 2 * ( V OUTMAX ⁄ V OUT ) 2 – 1 ) (EQ. 7) 0.5 I OUT 2 * L C OUT = ---------------------------------------------------V OUT 2 * ( 1.05 2 – 1 ) (EQ. 8) 0.4 IRMS/IO where VOUTMAX/VOUT is the relative maximum overshoot allowed during the removal of the load. For an overshoot of 5%, the equation becomes: 0.3 0.2 The graph in Figure 44 shows the relationship of COUT and % overshoot at 3 different output voltages. L is assumed to be 7µH and IOUT is 3A. 0.1 0 0 0.2 0.4 0.6 0.8 D FIGURE 45. IRMS/IO vs DUTY CYCLE COUT (µF) 80 A minimum of 10µF ceramic capacitance is required on each VIN pin. The capacitors must be as close to the IC as physically possible. Additional capacitance may be used. 60 3.3VOUT Loop Compensation Design 40 20 0 1.02 5VOUT ISL85033 uses a constant frequency current mode control architecture to achieve simplified loop compensation and fast loop transient response. 12VOUT 1.04 1.06 1.08 1.10 VOUTMAX/VOUT FIGURE 44. COUT vs OVERSHOOT VOUTMAX/VOUT Current Sharing Configuration In current sharing configuration, FB1 is connected to FB2, EN1 to EN2, COMP1 to COMP2 and VOUT1 to VOUT2 as shown in Figure 3. As a result, the equivalent gm doubles its single channel value. Since the two channels are out-of-phase, the frequency will be 2X the channel switching frequency. Ripple current cancellation will reduce the ripple current seen by the output capacitors and thus lower the ripple voltage. This results in the ability to use less capacitance than would be required by a single phase design of similar rating. Ripple current cancellation also reduces the ripple current seen at the input capacitors. Input Capacitor Selection To reduce the resulting input voltage ripple and to minimize EMI by forcing the very high frequency switching current into a tight local loop, an input capacitor is required. The input capacitor must have adequate ripple current rating which can be approximated by the Equation 9. 19 The compensator schematic is shown in Figure 47. As mentioned in the COUT selection, ISL85033 allows the usage of low ESR output capacitor. Choice of the loop bandwidth fc is somewhat arbitrary but should not exceed 1/4 of the switching frequency. As a starting point, the lower of 100kHz or 1/6 of the switching frequency is reasonable. The following equations determine initial component values for the compensation, allowing the designer to make the selection with minimal effort. Further detail is provided in “Theory of Compensation” on page 20 to allow fine tuning of the compensator. Compensation resistor R1 is given by Equation 10: 2πf c V o C o R T R 1 = --------------------------------g m V FB (EQ. 10) which when applied to ISL85033 becomes: R 1 [ kΩ ] = 0.008247∗ f c∗ V o∗ C o (EQ. 11) where Co is the output capacitor value [µF], fc = loop bandwidth [kHz] and Vo is the output voltage [V]. Compensation capacitors C1 [nF], C2 [pF] are given by Equation 12: 3 6 C o × V o × ( 10 ) C o × R c × ( 10 ) C 1 = ---------------------------------------- ,C 2 = ---------------------------------------Io × R1 R1 (EQ. 12) FN6676.4 September 22, 2011 ISL85033 where Io [A] is the output load current, R1 (Ω) and Rc (Ω) is the ESR of the output capacitor Co. Example: Vo = 5V, Io = 3A, fs = 500kHz, fc = 50kHz, Co = 47µF/Rc = 5mΩ, then the compensation resistance R1 = 96kΩ. 2 C1 = 815pF, C2 = 2.5pF (There is approximately 3pF parasitic capacitance from VCOMP to GND; therefore, C2 is optional). Theory of Compensation The sensed current signal is injected into the voltage loop to achieve current mode control to simplify the loop compensation design. The inductor is not considered as a state variable for current mode control and the system becomes a single order system. It is much easier to design a compensator to stabilize the voltage loop than voltage mode control. Figure 46 shows the small signal model of the synchronous buck regulator. ^ VIN + ^i IN ILd^ 1:D RT Transfer function F1(S) from control to output voltage is calculated in Equation 16: S 1 + -----------ω esr vˆ o - = V in -------------------------------------F 1 ( S ) = ----2 dˆ S S ------- + -------------- + 1 2 ω Q o p ωo (EQ. 16) C 1 1 Where ω esr = ------------- ,Q p ≈ Ro -----o- ,ω o = -------------L LC o (EQ. 17) 1 Ti(S) K Fm + Power Stage Transfer Functions S 1 + -----ˆI V in ωz o F 2 ( S ) = ---ˆ- = ------------------- -------------------------------------Ro + RL 2 d S S ------- + -------------- + 1 2 ω Q o p ωo Rc Ro Co d^ 2 Transfer function F2(S) from control to inductor current is given by Equation 17: VINd^ + (EQ. 15) Where Qn and ωn are given by Qn = – --π-, = ω n = πf s . Rc Co ^ VO L In current loop, the current signal is sampled every switching cycle. Equation 15 shows the transfer function: S S H e ( S ) = ------- + -------------- + 1 2 ω Q n n ωn The compensation capacitors are: ^ iL CURRENT SAMPLING TRANSFER FUNCTION He(S) Where ω z = ------------Ro Co . Current loop gain Ti(S) is expressed as Equation 18: ^ VCOMP The voltage loop gain with open current loop is calculated in Equation 19: -Av(S) FIGURE 46. SMALL SIGNAL MODEL OF SYNCHRONOUS BUCK REGULATOR The PWM comparator gain Fm for peak current mode control is given by Equation 13: (EQ. 13) Where Se is the slew rate of the slope compensation and Sn is given by Equation 14. V in – V o S n = R t -------------------L (EQ. 14) Where: RT is trans-resistance, and is the product of the current sensing resistance and gain of the current amplifier in current loop. 20 (EQ. 19) T v ( S ) = KF m F 1 ( S )A v ( S ) The voltage loop gain with current loop closed is given by Equation 20: PWM Comparator Gain Fm 1 dˆ - = -----------------------------F m = --------------( S e + S n )T s vˆ comp (EQ. 18) T i ( S ) = R T F m F 2 ( S )H e ( S ) Tv(S) He(S) Tv ( S ) L v ( S ) = ----------------------1 + Ti ( S ) (EQ. 20) V FB K = --------- , V FB is the feedback voltage of the voltage Vo Where error amplifier. If Ti(S)>>1, then Equation 20 can be simplified as shown in Equation 21: S 1 + -----------V FB R o + R L ω esr A v ( S ) 1 L v ( S ) = --------- ------------------- ---------------------- --------------- , ω p ≈ ------------RT Vo Ro Co S He ( S ) 1 + ------ωp (EQ. 21) From Equation 21, it is shown that the system is a single order system, which has a single pole located at ω P before the half switching frequency. Therefore, a simple type II compensator can be easily used to stabilize the system. FN6676.4 September 22, 2011 ISL85033 Put the compensator zero at 6.6kHz (~1.5x CoRo), and put the compensator pole at ESR zero, which is 1.45MHz. The compensator capacitors are: Vo R2 C3 V FB - V REF R3 C1 = 470pF, C2 = 3pF (There is approximately 3pF parasitic capacitance from VCOMP to GND; therefore, C2 is optional). V COMP GM Figure 48A shows the simulated voltage loop gain. It is shown that it has 80kHz loop bandwidth with 69° phase margin and 15dB gain margin. Optional addition phase boost can be added to the overall loop response by using C3. + R1 C2 C1 60 45 FIGURE 47. TYPE II COMPENSATOR 30 Figure 47 shows the type II compensator and its transfer function is expressed as Equation 22: 15 S ⎞⎛ S ⎛ 1 + -----------1 + ------------⎞ ⎝ gm ω cz1⎠ ⎝ ω cz2⎠ vˆ comp - = ------------------- --------------------------------------------------------A v ( S ) = --------------C1 + C2 S vˆ FB S ⎛ 1 + ----------⎞ ⎝ ω ⎠ GAIN (dB) 0 (EQ. 22) cp -15 -30 Where: 100 1•103 1•104 1•105 1•106 1•105 1•106 FIGURE 48A. C1 + C2 1 1 ω cz1 = -------------- , ω cz2 = --------------, ω cp = --------------------R1 C1 C2 R1 C1 R2 C3 (EQ. 23) 100 the compensator design goal is: 80 High DC gain ⎛1 1⎞ - f Loop bandwidth fc: ⎝ --4- to -----10⎠ s 60 Gain margin: >10dB 40 Phase margin: 40° 20 The compensator design procedure is shown in Equation 24: 1 Put compensator zero ω cz1 = ( 1to3 ) --------------R C (EQ. 24) O O Put one compensator pole at zero frequency to achieve high DC gain, and put another compensator pole at either ESR zero frequency or half switching frequency, whichever is lower. The loop gain Tv(S) at crossover frequency of fc has unity gain. Therefore, the compensator resistance R1 is determined by Equation 25: 2πf c V o C o R T R 1 = --------------------------------g m V FB (EQ. 25) where gm is the trans-conductance of the voltage error amplifier, typically 200µA/V. Compensator capacitor C1 is then given by Equation 26: 1 1 C 1 = ----------------- ,C 2 = ------------------------R 1 ω cz 2πR 1 f esr (EQ. 26) Example: VIN = 12V, Vo = 5V, Io = 3A, fs = 500kHz, Co = 220µF/5mΩ, L = 5.6µH, gm = 200µs, RT = 0.21, VFB = 0.8V, Se = 1.1×105V/s, Sn = 3.4×105V/s, fc = 80kHz, then compensator resistance R1 = 72kΩ. 21 PHASE (°) 0 -20 100 1•103 1•104 FIGURE 48B. Rectifier Selection Current circulates from ground to the junction of the external Schottky diode and the inductor when the high-side switch is off. As a consequence, the polarity of the switching node is negative with respect to ground. This voltage is approximately -0.5V (a Schottky diode drop) during the off-time. The rectifier's rated reverse breakdown voltage must be at least equal to the maximum input voltage, preferably with a 20% derating factor. The power dissipation when the Schottky diode conducts is expressed in Equation 27: V OUT⎞ ⎛ P D [ W ] = I OUT ⋅ V D ⋅ ⎜ 1 – -------------⎟ V IN ⎠ ⎝ (EQ. 27) Where: VD is the voltage drop of the Schottky diode. Selection of the Schottky diode is critical in terms of the high temperature reverse bias leakage current which is very dependent on VIN and FN6676.4 September 22, 2011 ISL85033 exponentially increasing with temperature. Due to the nature of reverse bias leakage vs temperature, the diode should be carefully selected to operate in the worst case circuit conditions. Catastrophic failure is possible if the diode chosen experiences thermal runaway at elevated temperatures. Please refer to Application Note for diode selection. Power Derating Characteristics To prevent the ISL85033 from exceeding the maximum junction temperature, some thermal analysis is required. The temperature rise is given by Equation 28: (EQ. 28) T RISE = ( PD ) ( θ JA ) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by Equation 29: (EQ. 29) T J = ( T A + T RISE ) where TA is the ambient temperature. For the QFN package, the θJA is +38°C/W. The actual junction temperature should not exceed the absolute maximum junction temperature of +125°C When considering the thermal design, remember to consider the thermal needs of the rectifier diode. MAXIMUM AMBIENT TEMPERATURE (°C) The ISL85033 delivers full current at ambient temperatures up to +85°C if the thermal impedance from the thermal pad maintains the junction temperature below the thermal shutdown level, depending on the Input Voltage/Output Voltage combination and the switching frequency. The device power dissipation must be reduced to maintain the junction temperature at or below the thermal shutdown level. Figure 49 illustrates the power derating versus ambient temperature for the ISL85033 EVAL kit. Note that the EVAL kit derating curve is based on total circuit dissipation, not IC dissipation alone. 120 110 100 90 80 70 60 50 40 30 20 10 0 0 2 3 4 5 6 7 8 9 10 ISL85033EVAL1ZB EVAL BOARD TOTAL POWER DISSIPATION (W) As an example, consider the turn-off transition of the upper MOSFET. Prior to turn-off, the MOSFET is carrying the full load current. During turn-off, current stops flowing in the MOSFET and is picked up by the Schottky diode. Any parasitic inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide traces minimizes the magnitude of voltage spikes. There are two sets of critical components in the ISL85033 switching converter. The switching components are the most critical because they switch large amounts of energy, and therefore tend to generate large amounts of noise. Next, are the small signal components which connect to sensitive nodes or supply critical bypass current and signal coupling. A multi-layer printed circuit board is recommended. Figure 50 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer, usually a middle layer of the PC board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminals to the output inductor short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. In order to dissipate heat generated by the internal LDO and MOSFET, the ground pad should be connected to the internal ground plane through at least four vias. This allows the heat to move away from the IC and also ties the pad to the ground plane through a low impedance path. The switching components should be placed close to the ISL85033 first. Minimize the length of the connections between the input capacitors, CIN, and the power switches by placing them nearby. Position both the ceramic and bulk input capacitors as close to the upper MOSFET drain as possible. Position the output inductor and output capacitors between the upper and Schottky diode and the load. ΘJA = 38°C/W 1 can degrade efficiency, radiate noise into the circuit, and lead to device overvoltage stress. Careful component layout and printed circuit board design minimizes these voltage spikes. 11 12 The critical small signal components include any bypass capacitors, feedback components, and compensation components. Place the PWM converter compensation components close to the FB and COMP pins. The feedback resistors should be located as close as possible to the FB pin with vias tied straight to the ground plane as required. FIGURE 49. POWER DERATING CURVE Layout Considerations Layout is very important in high frequency switching converter design. With power devices switching efficiently between 100kHz and 600kHz, the resulting current transitions from one device to another cause voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes 22 FN6676.4 September 22, 2011 D1 Cout1 ISL85033 SL85033 .. .. .. vias Cin1 Cin2 LX2 trace L2 D2 Cout2 VOUT2 VOUT2 VIN1 VIN2 VOUT1 Cboot LX1 trace Fb2 Cboot Comp1 Fb1 L1 Comp2 ISL85033 FIGURE 50. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS 23 FN6676.4 September 22, 2011 ISL85033 Revision History The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you have the latest Rev. DATE 9/14/11 8/9/11 4/5/11 REVISION CHANGE FN6676.4 In the “Pin Descriptions” on page 3, for “SYNCIN”, replaced “Set the internal switching frequency 20% lower than the external SYNC frequency applied to the SYNCIN pin" with "External SYNC frequency applied to the SYNCIN pin should be at least 2.4 times the internal switching frequency setting" On page 8, changed parameter name from “Syncronization Frequency” to “Switching Frequency”. FN6676.3 Converted to new template Updated Intersil Trademark statement at bottom of page 1 per directive from Legal. Page 2 in the pin table definition, please add the following sentence to the Pin 11 (VCC) description after “Output of the internal 5V linear regulator. Decouple to PGND with a minimum of 4.7μF ceramic capacitor.” “This pin is provided only for internal bias of ISL85033 (not to be loaded with current over 10mA).” Page 8 all Absolute Max Ratings that are “5.5” should be changed to “5.9” 10/15/10 FN6676.2 Added the following sentence to the “SYNCIN” description in the “Pin Descriptions” table on page 3: “Set the internal switching frequency 20% lower than the external SYNC frequency applied to the SYNCIN pin.” Added the following sentence to “Synchronization Control” on page 18: “The switching frequency for each output is half of the SYNCIN frequency.” Revised tape and reel note in “Ordering Information” on page 6 from: “Add “-T” suffix for Tape and Reel. Please refer to TB347 for details on reel specifications” to: “Add “-T*” suffix for Tape and Reel. Please refer to TB347 for details on reel specifications” This is in order to delineate all tape and reel options. 9/14/10 Corrected Eq. 1 on page 16 from: R 2 x0.8V R 3 = ------------------------------V OUT – 0.8V to: R 2 = ( V OUT – 0.8 ) • R 3 ⁄ 0.8 Revised preceding paragraph from: “The output voltage programming resistor, R3, depends on the value chosen for the feedback resistor, R2, and the desired output voltage, VOUT, of the regulator. Equation 1 describes the relationship between VOUT and resistor values. R2 is often chosen to be in the 1kΩ to 10kΩ range.” to: “The output voltage programming resistor, R2, depends on the value chosen for the feedback resistor, R3, and the desired output voltage, VOUT, of the regulator. Equation 1 describes the relationship between VOUT and resistor values. R3 is often chosen to be in the 1kΩ to 10kΩ range.” 6/21/10 FN6676.1 Changed MIN/MAX for “Soft-Start Charging Current” on page 8 from 1.5/2.5µA to 1.4/2.6µA 6/18/10 FN6676.0 Initial Release. 24 FN6676.4 September 22, 2011 ISL85033 Products Intersil Corporation is a leader in the design and manufacture of high-performance analog semiconductors. The Company's products address some of the industry's fastest growing markets, such as, flat panel displays, cell phones, handheld products, and notebooks. Intersil's product families address power management and analog signal processing functions. Go to www.intersil.com/products for a complete list of Intersil product families. *For a complete listing of Applications, Related Documentation and Related Parts, please see the respective device information page on intersil.com: ISL85033 To report errors or suggestions for this datasheet, please go to www.intersil.com/askourstaff FITs are available from our website at http://rel.intersil.com/reports/search.php For additional products, see www.intersil.com/product_tree Intersil products are manufactured, assembled and tested utilizing ISO9000 quality systems as noted in the quality certifications found at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. 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For information regarding Intersil Corporation and its products, see www.intersil.com 25 FN6676.4 September 22, 2011 ISL85033 Package Outline Drawing L28.4x4 28 LEAD THIN QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 0, 9/06 A 4 . 00 2 . 50 PIN #1 INDEX AREA CHAMFER 0 . 400 X 45° 0 . 40 22 28 1 0 . 40 15 3 . 20 2 . 50 4 . 00 21 0 . 4 x 6 = 2.40 REF B PIN 1 INDEX AREA 7 0 . 10 2X 14 8 0 . 20 ±0 . 05 0 . 10 M C A B 0 . 4 x 6 = 2 . 40 REF TOP VIEW 3 . 20 BOTTOM VIEW SEE DETAIL X'' 0 . 10 C (3 . 20) C PACKAGE BOUNDARY MAX. 0 . 80 SEATING PLANE (28X 0 . 20) 0 . 00 - 0 . 05 0 . 20 REF 0 . 08 C (3 . 20) (2 . 50) SIDE VIEW (0 . 40) C (0 . 40) 0 . 20 REF 5 0 ~ 0 . 05 (2 . 50) (28X 0 . 60) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Controlling dimensions are in mm. Dimensions in ( ) for reference only. 2. Unless otherwise specified, tolerance : Decimal ±0.05 Angular ±2° 3. Dimensioning and tolerancing conform to AMSE Y14.5M-1994. 4. Bottom side Pin#1 ID is diepad chamfer as shown. 5. Tiebar shown (if present) is a non-functional feature. 26 FN6676.4 September 22, 2011