Application Note, V1.3, April 2007 ICE1PCS01/02 Boost Type CCM PFC Design with ICE1PCS01/02 Power Management & Supply N e v e r s t o p t h i n k i n g . Edition 2007-04-11 Published by Infineon Technologies Asia Pacific, 168 Kallang Way, 349253 Singapore, Singapore © Infineon Technologies AP 2005. All Rights Reserved. Attention please! The information herein is given to describe certain components and shall not be considered as a guarantee of characteristics. Terms of delivery and rights to technical change reserved. We hereby disclaim any and all warranties, including but not limited to warranties of non-infringement, regarding circuits, descriptions and charts stated herein. Information For further information on technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies Office (www.infineon.com). Warnings Due to technical requirements components may contain dangerous substances. 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ICE1PCS02 Revision History: Previous Version: Page 2007-04 none Subjects (major changes since last revision) Boost Type CCM PFC Design with ICE1PCS01/02 License to Infineon Technologies Asia Pacific Pte Ltd Luo Junyang Liu Jianwei Jeoh Meng Kiat We Listen to Your Comments Any information within this document that you feel is wrong, unclear or missing at all? Your feedback will help us to continuously improve the quality of this document. Please send your proposal (including a reference to this document) to: mailto:[email protected] V1.3 ICE1PCS01 Table of Contents Page 1 Introduction ...................................................................................................................................5 2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 2.10 2.11 2.12 2.13 Boost PFC design with ICE1PCS01/02 .......................................................................................7 Target specification .........................................................................................................................7 Bridge rectifier .................................................................................................................................7 Power MOSFET and Gate Drive Circuit .........................................................................................7 Boost Diode.....................................................................................................................................8 Boost inductor .................................................................................................................................9 AC line current filter.......................................................................................................................10 Boost Output Bulk Capacitance ....................................................................................................11 Current Sense Resistor.................................................................................................................11 Output voltage sensing divider......................................................................................................12 Frequency setting (only for ICE1PCS01)......................................................................................12 AC Brown-out Shutdown (only for ICE1PCS02) ...........................................................................13 IC supply .......................................................................................................................................15 Voltage loop and current loop compensation................................................................................15 Application Note 4 2007-04-11 Abstract Continuous conduction mode (CCM) PFC controllers, named ICE1PCS01/02, are developed based on a new control scheme. Compared to the conventional PFC solution, the new ICs does not need the direct sinewave sensing reference signal from the AC mains. Average current control is implemented to achieve the unity power factor. In this application note, the design process for the boost PFC with ICE1PCS01/02 is presented and the design details for a 300W output power PFC with the universal input voltage range of 85~265VAC are included. 1 Introduction The Pin layout of ICE1PCS01 and ICE1PCS02 is shown in Figure 1. GND 1 8 GATE ICOMP 2 7 VCC ISENSE 3 6 VSENSE FREQ 4 5 VCOMP ICE1PCS01 Figure 1 GND 1 8 GATE ICOMP 2 7 VCC ISENSE 3 6 VSENSE VINS 4 5 VCOMP ICE1PCS02 Pin Layout of ICE1PCS01 and ICE1PCS02 From the layout, it can be seen that most of Pins in ICE1PCS02 are the same as ICE1PCS01 except Pin 4. In ICE1PCS01, Pin 4 is to set the switching frequency. However, for ICE1PCS02, Pin 4 is for AC brown out detection and the switching frequency is fixed by internal oscillator at 65kHz. The typical application circuits of ICE1PCS01 and ICE1PCS02 are shown in Figure 2 and Figure 3 respectively. Application Note 5 2007-04-11 Rectifier EMI Filter D1 L1 COUT R1 VOUT =400VDC T1 R2 RSENSE VIN=85V ...265V AC R3 ISENSE Auxiliary Supply GATE ICE1PCS01 VCC ICOMP VCOMP FREQ R4 C2 RFREQ C1 Figure 2 GND VSENSE C3 Typical application circuit of ICE1PCS01 Rectifier EMI Filter D1 L1 COUT R1 VOUT =400VDC T1 D2 R3 ISENSE GATE R5 R6 Figure 3 GND VSENSE ICE1PCS02 VINS Application Note R2 RSENSE VIN=85V ...265V AC ICOMP C1 C4 Auxiliary Supply VCC VCOMP R4 C2 C3 Typical application circuit of ICE1PCS02 6 2007-04-11 2 Boost PFC design with ICE1PCS01/02 2.1 Target specification The fundamental electrical data of the circuit are the input voltage range Vin, the output power Pout, the output voltage Vout, the operating switching frequency fSW and the value of the high frequency ripple of the AC line current Iripple. Table 1 shows the relevant values for the system calculated in this Application Note. The efficiency at rated output power Pout is estimated to 90 % over the complete input voltage range. Input voltage 85VAC~265VAC Input frequency 50Hz Output voltage and current 390VDC, 0.76A Output power 300W Efficiency >90% at full load Switching Frequency 65kHz Maximum Ambient temperature around PFC 70ºC Table 1 Design parameter for the proposed design 2.2 Bridge rectifier In order to obtain 300W output power at 85 V minimum AC input voltage, the maximum input RMS current is I in _ RMS = Pout Vin _ min ⋅ η = 300 = 3.92 A 85 ⋅ 90% and the sinusoidal peak value of AC current is I in _ pk = 2 ⋅ I in _ RMS = 2 ⋅ 3.92 = 5.54 A For these values a bridge rectifier with an average current capability of 6A or higher is a good choice. Please note here, that due to a power dissipation of approximately PBR = 2 ⋅ VF ⋅ I in _ RMS = 2 ⋅ 1V ⋅ 3.92 A = 7.84W the rectifier bridge should be connected to an appropriate heatsink. Assuming a maximum junction temperature TJmax of 125°C, a maximum ambient temperature TAmax of 70°C, the thermal junction-to-case RthJC of approximate 2.5 K/W and the thermal case to heatsink RthCHS of approximate 1K/W, the heatsink must have a maximum thermal resistance of RthHS _ BR = TJ max − T A max 125 − 70 − RthJC − RthCHS = − 2.5 − 1 = 3.52 K / W PBR 7.84 2.3 Power MOSFET and Gate Drive Circuit Due to the switch mode operation, the losses are only active during the on-time of the MOSFET. The duty cycle of the transistor in boost converters operating in CCM at minimum AC input RMS voltage is Don = 1 − Vin _ min Vout Application Note = 1− 85 = 0.782 390 7 2007-04-11 Since rms-values have the same effect on a system as DC-values, it is possible to calculate a characteristic duty cycle for the rms-value. Therefore, the on-state losses of the MOSFET in CCM-mode at a junctiontemperature of 125°C are 2 Pcond = I in _ RMS ⋅ Don ⋅ Rdson (125C ) the MOSFET switching loss can be estimated as PSW = ( E on + E off ) ⋅ f SW where, Eon and Eoff are the switch-on and switch-off energy loss which data can be found in MOSFET datasheet, fSW is the switching frequency. For 300W design, if SPP20N60C3 is used, the conduction loss is Pcond = 3.92 2 ⋅ 0.782 ⋅ 0.42 = 5.05W assuming the switching current is about 6A and gate drive resistance Rg=3.6Ω, then the switching loss is PSW = (0.007mWs + 0.015mWs) * 65kHz = 1.43W the total loss is PMOS _ total = Pcond + PSW = 6.48W the required heatsink for the MOSFET is RthHS _ MOS = TJ max − T A max 125 − 70 − RthJC _ MOS − RthCHS = − 0.6 − 1 = 6.89 K / W PMOS _ total 6.48 the gate drive resistance is used to drive MOSFET as fast as possible but also keep dv/dt within EMI specification. In this 300W example, 3.6Ω gate resistor is chosen for SPP20N60C3 MOSFET. Beside gate drive resistance, one 10kΩ resistor is also commonly connected between MOSFET gate and source to discharge gate capacitor. 2.4 Boost Diode The boost diode D1 has a big influence on the system’s performance due to the reverse recovery behaviour. So the Ultra-fast diode with very low trr and Qrr is necessary to reduce the switching loss. The new diode technology of silicon carbide (SiC) Schottky shows its outstanding performance with almost no reverse recovery behaviour. The switching loss due to the boost diode can be ignored with choosing SiC Schottky diode. Only conduction loss is calculated as below. Pdiode = VF ⋅ I in _ RMS ⋅ (1 − Don ) = 2V ⋅ 3.92 A ⋅ (1 − 0.782) = 1.71W For a rule of thumb, the SiC diodes provide a output power Pout of a CCM-PFC-system of 100 W to120 W per rated ampere of the diode. This means for example, that the SDT04S60 from Infineon Technologies which is rated for a forward current IF = 4 A is capable for a system of Pout = 4*100 W = 400 W system in minimum. Therefore, this diode would be suitable for the proposed design. The required heatsink for boost diode is RthHS _ diode = TJ max − T A max 125 − 70 − RthJC _ diode − RthCHS = − 4.1 − 1 = 27.06 K / W Pdiode 1.71 Application Note 8 2007-04-11 The SiC boost diodes often have a poor surge current capability, so that they may break down. Therefore a so called bypass diode is necessary such as the diode D3 as Figure 4. For the proposed system, 1N5408 is suitable. D3 Rectifier D1 L1 R1 COUT T1 R2 RSENSE Figure 4 2.5 inrush current bypass diode Boost inductor The peak current that the inductor must carry is the peak line current at the lowest input voltage plus the high frequency ripple current. The high frequency ripple current peak to peak, IHF, can be related to maximum input power and minmum input voltage as equation below. I HF = k ⋅ 2 ⋅ Pin _ max Vin _ min Where, k must be kept reasonably small, and is usually optimized in the range of 15% to 25% for cost effective design based on the current magnetic component status. If k is too high, the larger AC input filter is required to filter out this ripple noise. If k is too low, the value of the inductance is too large and leads to big size of the magnetic core. For example, we choose k 22%. Then, I HF = 22% ⋅ ⋅ 2 ⋅ Pin _ max Vin _ min = 1 .2 A The peak current passing through inductor is I L _ pk = I in _ peak + I HF 1 .2 = 5.54 + = 6.14 A 2 2 The boost inductance must be Lboost ≥ D ⋅ (1 − D ) ⋅ Vout I HF ⋅ f SW D=0.5 will generate the maximum value for the above equation. Lboost ≥ 0.5 ⋅ (1 − 0.5) ⋅ 390V = 1.25mH 1.2 A ⋅ 65kHz The magnetic core of the boost choke can be either magnetic powder or ferrite material. (1) sendust powder toroid core The required effective magnetic volume of the core, Ve, is Application Note 9 2007-04-11 Ve ≥ µ r µ 0 Lboost ( I L _ pk Bmax ) 2 = 125 ⋅ 1.257 e − 6 ⋅ 1.25mH ( 6.14 A 2 ) = 11.6e − 6m 3 = 11.6cm 3 0.8T where, µr is the relative permeability which is fixed by core manufacturer; µ0 is magnetic field constant which is equal to 1.257e-6; Bmax is the maximum magnetic flux density for the selected magnetic material (for sendust, Bmax is up to 0.8T.) Select a core with similar Ve value from the magnetic core datasheet. For example, the core type CS468125 from Chang Sung Corporation is suitable for this case. The parameters of CS468125 are Ve=15.584cm3, Ae=1.34cm2, C=11.63cm, µr=125. The turn number of the boost choke winding is Lboost ⋅ C µ r µ 0 Ae N toroid _ boost = where, C is the magnetic path length and Ae is the effective magnetic cross section area. The copper loss of the winding wire can be calculated on Iin_RMS. 2 PL _ boost = I in _ RMS ⋅ RL _ boost Selecting the proper wire type to fullfil the loss and thermal requirement for the choke. (2) ferrite core To make sure the ferrite core will not go into saturation, the turn number of the boost choke winding with ferrite core is N ferrite _ boost ≥ I L _ pk ⋅ Lboost Bmax ⋅ Amin where, Bmax is up to 0.3T according to ferrite material specification; Amin is the minimum magnetic cross section area. The winding wire copper loss calculation is the same as in the above section of sendust powder toroid core. 2.6 AC line current filter As decribed in section 2.5, there is high frequency ripple current peak to peak IHF passing through boost choke. This ripple will also go into AC line power network. The current filter is necessary to reduce the amplitude of high frequency current component. The filtering circuit consists of a capacitor and an inductor as shown in Figure 5. Rectifier IHF_spec Current Filter IHF Lfilter Cfilter VIN=85V ...265VAC Figure 5 AC line current filter The required Lfilter is Application Note 10 2007-04-11 I HF L filter ≥ I HF _ spec +1 (2πf SW ) 2 C filter normally there is one EMI X2 capacitor which can act as Cfilter. In this example, if we define IHF_spec as 0.2A peak to peak and asumming X2 capacitance 0.47µF, then L filter 1.2 A +1 0 . 2 A ≥ = 89 µH (2π ⋅ 65kHz ) 2 ⋅ 0.47 µF The leakage inductance of EMI common mode choke can be used for current filter. If the leakage inductance is large enough, no need to add the additional differential mode inductor for filtering. Otherwise, a current filter choke is necessary. The calculation method for the current filter choke is the same as for boost choke. 2.7 Boost Output Bulk Capacitance The bulk capacitance has to fullfil two requirements, output double line frequency ripple and holdup time. (1) output double line frequency ripple limit. The inherent PFC always presents 2*fL ripple. The amplitude of ripple voltage is dependant on output current and bulk capacitance as below. C out ≥ I out π ⋅ 2 * f L ⋅ Vout _ ripple _ pp where, Iout is the PFC output current, Vout_ripple_pp is the output voltage ripple (peak to peak), and fL is the AC line frequency. Please note that ICE1PCS01/02 has enhance dynamic block which is active when Vout exceed ±5% of regulated level. The enchanc dynamic block should be designed to work only during load or line change. During steady state with constant load, the enhance dynamic block should not be triggered, otherwise THD will be deteriorated. That means the target Vout_ripple_pp must be lower than 10% of Vout. For this example, Vout=390VDC, then Vout_ripple_pp must be lower than 39V. if we define Vout_ripple_pp=8V, then C out ≥ I out = 306µF π ⋅ 2 * f L ⋅ Vout _ ripple _ pp (2) holdup time requirement After the PFC stage, there is commonly a PWM stage to provide isolated DC output for end user. Some applications, especially computing, have the holdup time requirement. It means that PWM stage should be able to provide the isolated output even if AC input voltage become zero for a short holdup time. The common specification for this holdup time is 20ms. If minimum input voltage for PWM stage is defined as 250VDC, then the bulk capacitance will be C out ≥ 2 ⋅ Pout ⋅ t holdup 2 Vout − Vout _ min 2 = 2 ⋅ 300W ⋅ 20ms = 134 µF 390 2 − 250 2 the final Cout capacitance should be higher value calculated from the above two requirement. 2.8 Current Sense Resistor Application Note 11 2007-04-11 The current sense resistance is calculated based on the IC soft over current control threshold and peak current carried by boost choke. When the Isense signal reaches the soft over control threshold, IC will reduce the internal control voltage and accordingly the duty cycle is reduced in the following cycles. Finally the boost choke current is limited. According to IC datasheet, soft over current control threshold is -0.66V maximum. So the current sense resistor should be Rsense ≤ 0.66V 0.66V = = 0.11Ω 6.14 A I L _ pk According to Figure 2 and Figure 3, the transistor current as well as the diode current is sensed with Rsense. That means that also the inrush current is sensed there leading to a large negative voltage drop at Rsense, because the inrush current is in the range of about 150 A to 200 A. It is therefore necessary to limit the current into Pin 2 (ISENSE) to 1 mA, which is realized with resistor R3. A value of R3 = 220Ω is sufficient for this resistor. 2.9 Output voltage sensing divider The output voltage is set with the voltage divider represented by R1 and R2 in Figure 2 and Figure 3. First, choose the value of the lower resistor R2. Then the value of the upper resistor R1 is R1 = Vout − Vref Vref ⋅ R2 where, Vref is IC internal reference voltage for voltage sensing, 5V typical. If R2=10kΩ, R1 = 390 − 5 ⋅ 10kΩ = 770kΩ 5 It is recommended to take resistor values with a tolerance of 1% for R1 and R2. Due to the voltage stress of R1, it is recommended to split this value into few resistors in series. 2.10 Frequency setting (only for ICE1PCS01) The frequency of the ICE1PCS01 is adjustable in the range of 50 kHz up to 200 kHz. The external resistor RFREQ according to Figure 2 programs a current which controls the oscillator. The given points of the resistorfrequency-characteristic are (250 kHz / 18 k.), (125 kHz / 33 k.) and (50 kHz / 82k.) and Figure 6 shows the curve through those points. Application Note 12 2007-04-11 450000 400000 350000 Freq /Hz 300000 250000 200000 150000 100000 50000 0 0 50000 100000 150000 200000 Resistance /ohm Figure 6 2.11 Resistor-frequency characteristic AC Brown-out Shutdown (only for ICE1PCS02) Brown-out occurs when the input voltage VAC falls below the minimum input voltage of the design (i.e. 85V for universal input voltage range) and the VCC has not entered into the VCCUVLO level yet. For a system without input brown out protection (IBOP), the boost converter will increasingly draw a higher current from the mains at a given output power which may exceed the maximum design values of the input current and lead to over heat of MOSFET and boost diode. ICE1PCS02 provides a new IBOP feature whereby it senses directly the input voltage for Input Brown-Out condition via an external resistor/capacitor/diode network as shown in Figure 7. This network provides a filtered value of VIN which turns the IC on when the voltage at Pin 4 (VINS) is more than 1.5V. The IC enters into the standby mode and gate is off when VINS goes below 0.8V. The hysteresis prevents the system to oscillate between normal and standby mode. Note also that input voltage needs to at least 16% of the rated VOUT in order to overcome open loop protection and powerup the system (referred to application note of ICE1PCS01). Application Note 13 2007-04-11 VBR D2 ICE1PCS02 R5 1.5V C5 VINS R S Protection Logic C4 0.8V R6 Figure 7 C4 Block diagram of voltage loop Because of the high input impedence of comparator of C4 and C5, R5 can be high ohmic resistance to reduce the loss. From the datasheet, the bias current on VINS Pin is 1µA maximum. In order to have the design consistence, the current passing through R5 and R6 has to be much higher than this bias current, for example 7µA. Then R6 is: R6 = 0.8V = 114kΩ 7 µA R6 is selected 120KΩ. R5 is selcted by R5 = 2 ⋅ V AC _ on − 1.5V 1.5V ⋅ R6 where, VAC_on is the minimum AC input voltage (RMS) to start PFC, for example 70VAC. R5 = 2 ⋅ 70V − 1.5V ⋅ 120kΩ = 7.8MΩ 1.5V Due to the voltage stress of R5, it is recommended to split this value into few resistors in series. C4 is used to modulate the ripple at the VINS pin. The timing diagram of VINS pin when IC enters brown-out shutdown is shown in Figure 8. Application Note 14 2007-04-11 VBR 2*VINS_AVE-0.8V VINS_AVE 0.8V Figure 8 tdischarge Timing diagram of VINS Pin when IC enters brown-out shutdown If the bottom level of the ripple voltage touches 0.8V, PFC is in standby mode and gate is off. The ripple voltage defines PFC brown out off threshold of AC input voltage (RMS), VAC_off. C4 can be obtained from the following equation. Assuming V INS _ AVE = R6 ⋅ V AC _ off , where, VAC_off is the maximum AC input voltage R5 + R6 (RMS) to switch off PFC, for example 65VAC. − R6 (2 ⋅ ⋅ V AC _ off − 0.8) ⋅ e R5 + R6 t disch arg e R6C4 = 0.8V assuming tdischarge is equal to half cycle time of line frequency, ie. t disch arg e = R6 2⋅ V AC _ off − 0.8V R5 + R6 C 4 = 2 f L R6 ln 0.8V −1 120kΩ 2⋅ 65V − 0.8V 7 . 8 M 120 k Ω + Ω C 4 = 2 ⋅ 50 Hz ⋅ 120kΩ ln 0.8V 2.12 1 , then 2 fL −1 = 219nF IC supply Because of the low Vcc turn-on/off hysteresis, ICE1PCS01/02 is not supposed to be supplied by self supply method with auxiliary winding on boost choke. The ICs can only be supplied by external DC applying to Vcc. One Electrolyte capacitor 47uF and one ceramic capacitor 100nF is recommended to connect between Vcc and GND to filter out the noise. 2.13 Voltage loop and current loop compensation Application Note 15 2007-04-11 Please refer to Reference [5] for detail. References [1] Infineon Technologies: ICE1PCS01 - Standalone Power Factor Correction Controller in Continuous Conduction Mode; Preliminary datasheet; Infineon Technologies; Munich; Germany; Sept. 2002. [2] Luo Junyang, Jeoh Meng Kiat, Huang Heng Cheong, A New Continuous Conduction Mode PFC IC With Average Current Mode Control, PEDS 2003; pp. 1110-1114, Nov 2003. [3] Luo Junyang, Jeoh Meng Kiat, Yew Ming Lik, 300W CCM PFC Evaluation Board with ICE1PCS01, CoolMOS™ and SiC Diode thinQ!™, Application note, Infineon Technologies, Munich, Germany, Oct. 2003. [4] Infineon Technologies: ICE1PCS02 - Standalone Power Factor Correction (PFC) Controller in Continuous Conduction Mode (CCM) at Fixed Frequency, Preliminary datasheet; Infineon Technologies; Munich; Germany; Sept. 2004. [5] Luo Junyang, Huang Heng Cheong, Jeoh Meng Kiat, ICE1PCS01 Based Boost Type CCM PFC Design Guide - Control Loop Modeling, Application note, Infineon Technologies, Munich, Germany, July, 2004. Application Note 16 2007-04-11