DATASHEET

DATASHEET
Dual (180°Out-of-Phase) PWM and Linear Controller
ISL6446
Features
The ISL6446 is a high-performance, triple output controller
that provides a single high-frequency power solution primarily
for broadband, DSL and networking applications. This device
integrates complete control, monitoring and protection
functions for two synchronous buck PWM controllers and one
linear controller. Input voltage ripple and total RMS input
current is substantially reduced by synchronized 180°
out-of-phase operation of the two PWMs.
• 4.5V to 5.5V or 5.5V to 24V input voltage range
The two PWM buck converters provide simple voltage mode
control. The output voltage of the converters can be precisely
regulated to as low as 0.6V, with a maximum tolerance of
±1.5% over-temperature and line variations. Programmable
switching frequency down to 100kHz provides optimized low
cost solution for ATX power supplies. It’s also able to operate
up to 2.5MHz to deliver compact solutions. The linear
controller provides a low-current output.
• Three programmable power output voltages
- Two PWM controllers with out-of-phase operation
- Voltage-mode PWM control
- One linear controller
• Programmable switching frequency from 100kHz to 2.5MHz
• Fast transient response
- High-bandwidth error amplifier
• Extensive circuit protection functions
- Undervoltage and over-temperature
- Overvoltage with latch-off mode
- Programmable overcurrent limit with latch-off mode
- Lossless current sensing (no sense resistor needed)
Each PWM controller has soft-start and independent enable
functions combined on a single pin. A capacitor from SS/EN to
ground sets the soft-start time; pulling SS/EN pin below 1V
disables the controller. Both outputs can soft-start into a
prebiased load.
• Externally adjustable soft-start time
- Independent enable control
- Voltage tracking capability
- Able to soft-start into a prebiased load
The ISL6446 incorporates robust protection features. An
adjustable overcurrent protection circuit monitors the output
current by sensing the voltage drop across the upper MOSFET
rDS(ON). Latch-off mode overcurrent operation protects the
DC/DC converters from damage under overload and
short-circuit conditions. A PGOOD signal is issued when
soft-start is complete and PWM outputs are within 10% of
their regulated values and the linear regulator output is higher
than 75% of its nominal value. Thermal shutdown circuitry
turns the device off if the IC temperature exceeds +150°C.
• PGOOD output with delay
VCC
Applications
• ATX power supplies
• DSP, ASIC and FPGA point of load regulation
• Industrial and security networking applications
VIN
CVCC
VCC
COMP1
FB1
R1
VOUT2
Feedback
and
Compensation
Network
COMP2
FB2
ROCSET1
VIN1 = VIN
VIN
COCSET1
OCSET1 C
BOOT1
BOOT1
Q1 CIN1
UGATE1
VOUT1
PHASE1
L1
LGATE1
COUT1
Q2
ROCSET2
ISL6446
VIN2 = VIN
COCSET2
OCSET2 C
BOOT2 BOOT2
Q3 C
IN2
UGATE2
R2
PGOOD
RT
SS1/EN1
SS2/EN2
SGND
PHASE2
LGATE2
L2
Q4
LCDR
LCFB
PGND
VOUT2
COUT2
100
98
96
94
EFFICIENCY (%)
VOUT1
Feedback
and
Compensation
Network
VOUT = 5V
92
VOUT = 3.3V
90
88
86
84
82
80
0
5
10
15
20
25
LOAD CURRENT (A)
FIGURE 1. TYPICAL APPLICATION
October 12, 2015
FN7944.2
1
FIGURE 2. EFFICIENCY vs LOAD CURRENT (OBTAINED FROM
ISL6446EVAL1Z)
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Copyright Intersil Americas LLC 2012, 2013, 2015. All Rights Reserved
Intersil (and design) is a trademark owned by Intersil Corporation or one of its subsidiaries.
All other trademarks mentioned are the property of their respective owners.
ISL6446
Pin Configuration
ISL6446
(24 LD QSOP)
TOP VIEW
OCSET1 1
24 VIN
SS1/EN1 2
23 BOOT1
COMP1 3
22 UGATE1
FB1 4
21 PHASE1
RT 5
20 LGATE1
SGND 6
19 VCC
LCDR 7
18 PGND
LCFB 8
17 LGATE2
FB2 9
16 PHASE2
COMP2 10
15 UGATE2
SS2/EN2 11
14 BOOT2
OCSET2 12
13 PGOOD
Pin Descriptions
SYMBOL
PIN #
DESCRIPTION
BOOT2, 1
14, 23 These pins power are the upper MOSFET drivers of each PWM converter. The anode of each internal bootstrap diode is connected
to the VCC pin. The cathode of the bootstrap diode is connected to this pin, which should also connect to the bootstrap capacitor.
UGATE2, 1
15, 22 These pins provide the gate drive for upper MOSFETs, bootstrapped from the VCC pin.
PHASE2, 1
16, 21 These are the junction points of the upper MOSFET sources, the output filter inductor and lower MOSFET drains. Connect these
pins accordingly to the respective converter.
LGATE2, 1
17, 20 These are the outputs of the lower N-Channel MOSFET drivers, sourced from the VCC pin.
PGND
18
This pin provides the power ground connection for the lower gate drivers. This pin should be connected to the source of the lower
MOSFET for PWM1 and PWM2 and the negative terminals of the external input capacitors.
FB1, 2
4, 9
These pins are connected to the feedback resistor divider and provide the voltage feedback signals for the respective controller. They
set the output voltage of the converter. In addition, the PGOOD circuit and OVP circuit use these inputs to monitor the output voltage
status.
COMP1, 2
3, 10
These pins are the error amplifier outputs for the respective PWM. They are used, along with the FB pins, as the compensation point
for the PWM error amplifier.
PGOOD
13
This is an open drain logic output used to indicate the status of the output voltages. This pin is pulled low when either of the two
PWM outputs is not within 10% of the respective nominal voltage or when the linear output drops below 75% of its nominal voltage.
To maintain the PGOOD function if the linear output is not used, connect LCFB to VCC.
SGND
6
This is the signal ground, common to both controllers, and must be routed separately from the high current grounds (PGND). All
voltage levels are measured with respect to this pin.
VIN
24
This pin powers the controllers with an internal linear regulator (if VIN > 5.5V) and must be closely decoupled to ground using a
ceramic capacitor as close to the VIN pin as possible. The VIN is the input voltage applied to the upper FET of both converters.
TABLE 1. INPUT SUPPLY CONFIGURATION
INPUT
PIN CONFIGURATION
5.5V to 24V
Connect the input supply to the VIN pin. The VCC
pin will provide a 5V output from the internal
voltage regulator.
5V ±10%
VCC
19
Connect the input supply to the VCC pin.
This pin supplies the bias for the regulators, powers the low-side gate drivers and external boot circuitry for high-side gate drivers.
The IC may be powered directly from a single 5V (±10%) supply at this pin; when used as a 5V supply input, this pin must be
externally connected to VIN. When VIN > 5.5, VCC is the output of the internal 5V linear regulator output. The VCC pin must always
be decoupled to power ground with a minimum of 1µF ceramic capacitor, placed very close to the pin.
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ISL6446
Pin Descriptions
(Continued)
SYMBOL
PIN #
DESCRIPTION
RT
5
This is the operating frequency adjustment pin. By placing a resistor from this pin to the SGND, the oscillator frequency can be
programmed from 100kHz to 2.5MHz.
SS1/EN1
SS2/EN2
2, 11
These pins provide enable/disable and soft-start function for their respective controllers. The output is held off when the pin is
pulled to ground. When the chip is enabled, the regulated 30µA pull-up current source charges the capacitor connected from the
pin to ground. The output voltage of the converter follows the ramping voltage on the SS/EN pin. See “Soft-Start and Voltage
Tracking” on page 11 for more details.
LCFB
8
This pin is the feedback pin for the linear controller. An external voltage divider network connected to this pin sets the output
voltage of the linear controller. If the linear controller is not used, tie this pin to VCC.
LCDR
7
Open drain output PNP Transistor or P-channel MOSFET Driver. The LCDR connects to the base of an external PNP pass transistor
or the gate of the MOSFET to form a positive linear regulator. A small resistor can be inserted between the LCDR and the base
of the PNP pass transistor or the gate of the MOSFET to alleviate thermal stress at output short condition.
OCSET1, 2
1, 12
These pins are the overcurrent set points for the respective PWM controllers. Connect a resistor (ROCSET) from this pin to the
drain of the upper MOSFET. ROCSET, an internal 110µA current source, and the upper MOSFET ON-resistance rDS(ON) set the
converter overcurrent (OC) trip point according to Equation 1:
I OCSET  R OCSET
I OC = --------------------------------------------------r DS  ON 
(EQ. 1)
IOC includes the DC load current, as well as the ripple current. An overcurrent trip initiates hiccup mode.
The voltage on the OCSET pin should not exceed 0.7V above the VIN pin voltage for proper current sensing when the UGATE is
turned on.
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
ISL6446IAZ
PART
MARKING
ISL 6446IAZ
TEMP
RANGE (°C)
-40 to +85
PACKAGE
(RoHS Compliant)
24 Ld QSOP
PKG.
DWG. #
M24.15
NOTES:
1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6446. For more information on MSL please see tech brief TB363.
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ISL6446
Block Diagram
VIN
VCC
VCC
REFERENCE
BIAS CURRENT
POWER-ON
RESET
AND CONTROL
30µA 100µA 0.6V
5V LINEAR
REGULATOR
110µA
OCSET1
VCC
BOOT1
UVP1
OVP1
PG1
EN1
OUTPUT1
DRIVERS
COMP1
-
FB1
0.6V
+
+
VCC5
VCC5
30µA
-
FAULT1
30µA
SS1
UVP1
OVP1
PG1
EN1
STARTUP
SS1/EN1
SS1
EN1
SS2/EN2
SS2
EN2
PWM1
UGATE1
GATE CONTROL
LOGIC
VCC
DEAD TIME
CONTROL
UVP2
OVP2
PG2
EN2
SS2
PGND
RAMP1
0°
110µA
CLOCK AND
SAWTOOTH
GENERATOR
OCSET2
-
UGATE2
OUTPUT2
DRIVERS
PWM2
GATE CONTROL
LOGIC
VCC
DEAD TIME
CONTROL
+
-
BOOT2
UVP2
OVP2
PG2
EN2
RAMP2
180°
0.6V +
FB2
LGATE1
OVERCURRENT
VCC
FAULT2
PHASE1
PHASE2
LGATE2
OVERCURRENT
PGND
COMP2
FAULT3
PG3 LCFB
RT
PG1
PG2
PG3
PGOOD
gm*VE
+
0.6V
LCFB
LCDR
PGND
SGND
FIGURE 3. BLOCK DIAGRAM
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ISL6446
Typical Application Schematic
VOLTAGE INPUTS REQUIRED
VIN (4.5V TO 24V) = VIN1 = VIN2
VCC
VCC (5V; INTERNAL IF VIN > 5.6V)
VIN
OPTIONAL
CONNECTION
(FOR VIN = VCC = 5V)
VIN3 (VCC) FOR LINEAR
CVCC CVIN
TYPE-3 COMPENSATION SHOWN
ROCSET1
C102
VCC
R102
COMP1
C101
OCSET1
BOOT1
R101
VOUT1
VIN1 = VIN
VIN
COCSET1
CBOOT1
FB1
C103
R103
UGATE1
R100
C202
R202
R203
0.9µH
LGATE1
COMP2
C201
C203
2x680µF/18m
2x100µF
1x47µF
Q102
FB2
ROCSET2
OCSET2
TYPE-3 COMPENSATION SHOWN
VIN2 = VIN
BOOT2
COCSET2
CBOOT2
Q201
CIN2
UGATE2
0.9µH
VOUT2
VCC
L200
PHASE2
COUT2
Q202
LGATE2
RPGOOD
2x680µF/18m
2x100µF
R304
PGOOD
RT
VIN3
C301
SS1/EN1
R303
R302
CIN3
LCDR
SS2/EN2
Q301
VOUT3
LCFB
RRT
COUT1
ISL6446
R200
CSS1/EN1 CSS2/EN2
VOUT1
L100
PHASE1
R201
VOUT2
CIN1
Q101
SGND
PGND
R300
R301
COUT3
FIGURE 4. ISL6446 TYPICAL APPLICATION
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ISL6446
Absolute Maximum Ratings
Thermal Information
(Note 4)
SS1/EN1, SS2/EN2, COMP1, COMP2 to SGND . . . . . . . . . . -0.3V to +6.0V
VCC, FB1, FB2, RT, PGOOD to SGND . . . . . . . . . . . . . . . . . . . -0.3V to +6.0V
LCDR, LCFB to SGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.0V
VIN, OCSET1 and OCSET2 to PGND . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V
BOOT1 and BOOT2 to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
BOOT1 to PHASE1 and BOOT2 to PHASE2 . . . . . . . . . . . . . . -0.3V to +6.0V
UGATE1 to PHASE1 . . . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to (BOOT1 +0.3V)
UGATE2 to PHASE2 . . . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to (BOOT2 +0.3V)
LGATE1, LGATE2 to PGND . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to (VCC+0.3V)
PHASE1, PHASE2 to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-1V to 28V
SGND to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 0.3V
ESD Rating
Human Body Model (Tested per JESD22-A114E) . . . . . . . . . . . . . . .2.5kV
Machine Model (Tested per JESD22-115-A) . . . . . . . . . . . . . . . . . . . 150V
Latch-up (Tested per JEDEC-78B Level II Class A) . . . . ±100mA at +85°C
Thermal Resistance (Typical)
JA (°C/W) JC (°C/W)
QSOP Package (Notes 5, 6). . . . . . . . . . . . .
75
36
Maximum Junction Temperature (Plastic Package) . . . .-55°C to +150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -40°C to +85°C
Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Recommended Operating Conditions
VCC Supply Voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5V ±10%
VIN Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5.5V to 24V
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. All voltages are measured with respect to GND.
5. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
6. For JC, the “case temp” location is taken at the package top center.
Electrical Specifications Operating conditions unless otherwise noted: VIN = 12V, or VCC = 5V ±10%, TA = -40°C to +85°C. Typical values
are at +25°C. Boldface limits apply across the operating temperature range, -40°C to +85°C
PARAMETER
SYMBOL
TEST CONDITIONS
Input Operating Supply Current
ICC_op
VIN = 5.5V or 12V; LGATEx, UGATEx Open, FB
forced above regulation point (no switching)
Input Standby Supply Current
ICC_sb
VIN = 5.5V, 12V, 24V;
SS1/EN1 = SS2/EN2 = 0V
Output Voltage
VVCC
VIN = 5.6V, SS1/EN1 = SS2/EN2 = 0V
No additional load
Output Voltage
VVCC
VIN = 24V, SS1/EN1 = SS2/EN2 = 0V
No additional load
Output Voltage
VVCC
VIN = 12V, SS1/EN1 = SS2/EN2 = 0V
IVCC = 80mA
MIN
(Note 9)
TYP
MAX
(Note 9)
UNIT
4.5
7.5
mA
1.25
3
mA
VIN SUPPLY
VCC INTERNAL REGULATOR
VCC Current Limit (Note 7)
IICC_CL
VCC is pulled to PGND (Note 8)
VREF1,
VREF2
VIN = 5V or 12V; TA = +25°C
4.5
5.35
5.36
4.5
V
5.6
V
5.2
V
300
mA
REFERENCE AND SOFT-START
Reference Voltage at FB1, FB2
VREF1,
VREF2
Reference Voltage at FB1, FB2
0.6000
V
VIN = 5V or 12V; TA = 0°C to +85°C
0.5925
0.6085
V
VIN = 5V or 12V; TA = -40°C to +85°C
0.5900
0.6085
V
0.6100
V
VIN = 24V; TA = +25°C
0.6015
VIN = 24V; TA = 0°C to +85°C
0.5930
VIN = 24V; TA = -40°C to +85°C
0.5915
V
0.6100
V
ENx/SSx Soft-start Current
ISSx
20
30
40
µA
ENx/SSx Enable Threshold
VENx
850
940
1050
mV
ENx/SSx Enable Threshold Hysteresis
VENx_hys
(Note 7)
15
mV
ENx/SSx Soft-Start Top of Ramp
Voltage
VSSx_top
(Note 7)
3.12
V
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ISL6446
Electrical Specifications Operating conditions unless otherwise noted: VIN = 12V, or VCC = 5V ±10%, TA = -40°C to +85°C. Typical values
are at +25°C. Boldface limits apply across the operating temperature range, -40°C to +85°C (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 9)
TYP
MAX
(Note 9)
UNIT
POWER-ON RESET ON VCC
Rising Threshold
VPOR_r
4.2
4.4
4.48
V
Falling Threshold
VPOR_f
3.85
4.0
4.1
V
PWM CONVERTERS
tUGATE_min (Note 7)
Minimum UGATE On-time
100
ns
Maximum Duty Cycle
DCmax
VIN = 5V or 12V; fSW = 300kHz
95
%
Maximum Duty Cycle
DCmax
VIN = 5V; fSW = 2.58MHz
FBx Pin Bias Current
IFBx
VFB1 = VFB2 = 600mV
Low-end Frequency
fSW
VIN = 12V; RT = 163kΩ
Oscillation Frequency
fSW
VIN = 5V or 12V; RT = 52.3kΩ
270
300
330
kHz
VIN = 24V; RT = 52.3kΩ
270
305
340
kHz
80
-250
%
30
250
nA
OSCILLATOR
High-end Frequency
fSW
Frequency Adjustment Range
VIN = 5V; RT = 4.75kΩ
2.25
2.5
2.75
MHz
VIN = 12V; RT = 4.75kΩ
2.25
2.59
2.85
MHz
RT = 163kΩ; (Note 7)
0.1
MHz
2.6
MHz
(Note 8)
1.25
V
VPWM_OFF (Note 8)
1.25
V
VP-P
PWM Sawtooth Ramp Offset
kHz
RT = 4.75kΩ; (Note 7)
fSW
PWM Sawtooth Ramp Amplitude
(Peak-to-peak)
103
PWM CONTROLLER GATE DRIVERS (Note 7)
Upper Gate Pull-up Resistance
Upper Gate Pull-down Resistance
Lower Gate Pull-up Resistance
Upper Gate Pull-down Resistance
2.6
Ω
2
Ω
2.6
Ω
2
Ω
Rise Time
CL = 3300pF
25
ns
Fall Time
CL = 3300pF
25
ns
20
ns
Dead Time Between Drivers
ERROR AMPLIFIERS
DC Gain
Gain-bandwidth Product
Slew Rate
Gain
(Note 8)
88
dB
GBWP
(Note 8)
15
MHz
5
V/µs
4.2
V
SR
COMP = 10pF (Note 8)
Maximum Output Voltage
VEA_H
ICOMP_SRC = 40µA
Minimum Output Voltage
VEA_L
ICOMP_SINK = 40µA
3.9
0.8
1.1
V
PROTECTION AND OUTPUT MONITOR
Overvoltage Threshold
OV
111
116
121
%
Undervoltage Threshold
UV
77
82
88
%
OCSET Current Source
IOCSET
VOCSET = 4.5V, TA = -40°C
OCSET Current Source
IOCSET
VOCSET = 4.5V, TA = +25°C
OCSET Current Source
IOCSET
VOCSET = 4.5V, TA = +85°C
Drive Sink Current
ILCDR
LCDR
LCFB Feedback Threshold
VLCFB
TA = +25°C
80
µA
110
µA
140
µA
LINEAR CONTROLLER
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50
mA
0.595
V
TA = -40°C to +85°C
0.570
0.620
V
TA = 0°C to +70°C
0.580
0.610
V
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ISL6446
Electrical Specifications Operating conditions unless otherwise noted: VIN = 12V, or VCC = 5V ±10%, TA = -40°C to +85°C. Typical values
are at +25°C. Boldface limits apply across the operating temperature range, -40°C to +85°C (Continued)
PARAMETER
SYMBOL
LCFB Input Leakage Current
ILCFB
Error Amplifier Transconductance
gm
TEST CONDITIONS
MIN
(Note 9)
TYP
MAX
(Note 9)
UNIT
(Note 7)
80
nA
VLCFB = 0.6V, ILCDR = 21mA (Note 7)
2
A/V
PGOOD
Power-good Lower Threshold
PG_lowx
LCFB = VCC, LDO disabled
PGOOD for Ch1 and Ch2 only
88
93
97
%
Power-good Higher Threshold
PG_hix
LCFB = VCC, LDO disabled
PGOOD for Ch1 and Ch2 only
105
110
115
%
Power-good Lower Threshold
PG_low3
LDO enabled, PGOOD for LDO;
Ch1 and Ch2 disabled; (Note 7)
72
%
PGOOD Delay
tPGOOD
fSW = 1.4MHz (Note 7)
46
ms
PGOOD Leakage Current
IPGOOD
VPULLUP = 5.5V
5
µA
PGOOD Voltage Low
VPG_low
IPGOOD = -4mA
0.5
V
THERMAL
Shutdown Temperature
(Note 8)
150
°C
Shutdown Hysteresis
(Note 8)
20
°C
NOTES:
7. Limits established by characterization.
8. Design guideline only; not production tested.
9. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
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ISL6446
Typical Performance Curves Oscilloscope plots are taken using the ISL6446EVAL1Z evaluation board, VIN = 12V,
VOUT1 = 5V, VOUT2 = 3.3V, fSW = 300kHz, unless otherwise noted.
PHASE 1
48mVP-P
PHASE 2
VOUT1/AC
43mVP-P
FIGURE 5. OUTPUT RIPPLE (PWM1)
VIN = 12V
VOUT = 5V
IOUT = 5A~15A
VOUT2/AC
FIGURE 6. OUTPUT RIPPLE (PWM2)
VIN = 12V
VOUT1 = 5V
VOUT2 = 3.3V
VOUT1/AC
PHASE 1
IOUT1 = 0A
IOUT2 = 0A
ISTEP 1A/50mV
IL1
PHASE 2
FIGURE 8. PWM INTERLEAVING
FIGURE 7. LOAD TRANSIENT
VIN = 12V
VOUT1 = 5V
VOUT2 = 3.3V
PHASE 1
IOUT1 = 25A
IOUT2 = 25A
PHASE 2
FIGURE 9. PWM INTERLEAVING
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FN7944.2
October 12, 2015
ISL6446
Typical Performance Curves Oscilloscope plots are taken using the ISL6446EVAL1Z evaluation board, VIN = 12V,
VOUT1 = 5V, VOUT2 = 3.3V, fSW = 300kHz, unless otherwise noted. (Continued)
VIN = 12V
VOUT1 = 5V
VOUT2 = 3.3V
EN/SSx TIED
TOGETHER
EN/SSx
EN/SSx TIED
TOGETHER
VIN = 12V
VOUT1 = 5V
VOUT2 = 3.3V
VOUT1
VOUT1
VOUT2
VOUT2
IOUT1 = 0A
IOUT2 = 0A
IOUT1 = 25A
IOUT2 = 25A
FIGURE 11. EN/SS START-UP
FIGURE 10. EN/SS START-UP
VIN = 12V
VOUT1 = 5V
VOUT2 = 3.3V
EN/SSx
EN/SSx
VIN = 12V
VOUT1 = 5V
VOUT2 = 3.3V
EN/SSx
VOUT1
VOUT1
VOUT2
IOUT1 = 0A
IOUT2 = 0A
FIGURE 12. EN/SS SHUTDOWN
VIN = 12V
VOUT1 = 5V
IOUT = 0A
VOUT2
IOUT1 = 25A
IOUT2 = 25A
FIGURE 13. EN/SS SHUTDOWN
VIN = 12V
VOUT1 = 5V
IOUT = 10A~20A
EN/SSx
EN/SS
VOUT
PHASE
IL
LGATE
VOUT
PHASE
FIGURE 14. PREBIASED START-UP (VOUT PRE-BIASED AT 3.5V)
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FIGURE 15. OVERCURRENT PROTECTION
FN7944.2
October 12, 2015
ISL6446
Typical Performance Curves Oscilloscope plots are taken using the ISL6446EVAL1Z evaluation board, VIN = 12V,
VOUT1 = 5V, VOUT2 = 3.3V, fSW = 300kHz, unless otherwise noted. (Continued)
VIN = 12V
VOUT1 = 5V
IOUT = S/C
VIN = 12V
VOUT1 = 5V
IOUT = 10, FB = 1V
EN/SSx
EN/SSx
VOUT
IL
IL
VOUT
PHASE
PHASE
FIGURE 17. OVERVOLTAGE PROTECTION
FIGURE 16. START-UP WITH OC
Functional Description
Soft-Start and Voltage Tracking
After the VCC pin exceeds its rising POR trip point (nominal 4.4V),
the chip operation begins. Both 30µA current sources will start
charging up the soft-starting capacitors respectively. The charging
continues until the voltage across the soft-start capacitor reaches
about 3.2V. From 1.0V to 1.6V, the outputs will ramp individually
from zero to full-scale. Now, if V = 0.6V, C = 0.1µF, and I = 30µA,
then t = 2ms. Figure 18 shows the typical waveforms for SS2/EN2
and VOUT2; SS1/EN1 and VOUT1 are similar (see Figure 19).
Finally, there is a delay after 1.6V, until the ramp gets to ~3.2V,
which signals that the ramp is done; when both ramps are done,
the PGOOD delay begins. To guarantee the soft-start is
completed, please make sure the EN/SSx pin voltage is able to
reach above 3.2V at normal operation.
VOUT2 (1V/DIV)
VOUT (1V/DIV)
GND>
1.6V
SS2/EN2 (0.5V/DIV)
1.0V
SS1/EN1 (0.5V/DIV)
1.6V
1.0V
SS2/EN2 (0.5V/DIV)
GND>
VOUT2 (2V/DIV)
GND>
FIGURE 18. SOFT-START
The soft-start ramps for each output can be selected
independently. The basic timing is shown in Equation 2:
dV
t = C  ------I
(EQ. 2)
Where:
FIGURE 19. VOLTAGE TRACKING
Figure 20 shows prebiased outputs before soft-start. The solid
blue curve shows no prebias; the output starts ramping from
GND. The magenta dotted line shows the output prebiased to a
voltage less than the final output. The FETs don’t turn on until the
soft-start ramp voltage exceeds the output voltage; then the
output starts ramping seamlessly from there. The cyan dotted
line shows the output prebiased above the final output (but
below the OVP (Overvoltage Protection)). The FETs will not turn on
until the end of the soft-start ramp; then the output will be
quickly pulled down to the final value.
If the output is prebiased above the OVP level, the ISL6446 will go
into OVP at the end of soft-start, which will keep the FETs off. See
“Protection Mechanisms” on page 13 for more details.
t is the charge time
C is the external capacitance
dV is the voltage charged
I is the charging current (nominal 30µA)
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October 12, 2015
ISL6446
VOUT1 has the same functionality as previously described for
VOUT2. Each output should react independently of the other,
unless they are related by the circuit configuration.
SS2/EN2 (0.5V/DIV)
.
PGOOD (5V/DIV)
GND>
VOUT3 (2V/DIV)
GND>
VOUT2 (2V/DIV)
GND>
VOUT2 OVERCHARGED
VOUT2 (2V/DIV)
VOUT2 PREBIASED
VOUT1 (2V/DIV)
GND>
FIGURE 21. PGOOD DELAY
GND>
Switching Frequency
The linear output does not have a soft-start ramp; however, it
may follow the ramp of its input supply, if timed to coincide with
its rise, after the VCC rising POR trip. If the input to the linear is
from one of the two switcher outputs, then it will share the same
ramp rate as the switcher.
PGOOD
A group of comparators (separate from the protection
comparators) monitor the output voltages (via the FB pins) for
PGOOD. Each switcher has a lower and upper boundary
(nominally around 90% and 110% of the target value) and the
linear has a lower boundary (around 75% of the target). Once
both switcher output ramps are done, and all 3 outputs are
within their expected ranges, the PGOOD will start an internal
timer, with Equation 3:
0.065
t PGOOD = --------------f SW
(EQ. 3)
Where:
tPGOOD is the delay time (in sec)
fSW is the switching frequency (in MHz)
Once the time-out is complete, the internal pull-down device will
shut off, allowing the open-drain PGOOD output to rise through an
external pull-up resistor, to a 5V (or lower) supply, which signals
that the “Power is GOOD”. Figure 21 shows the three outputs
turning on, and the delay for PGOOD. If any of the conditions is
subsequently violated, then PGOOD goes low. Once the voltage
returns to the normal region, a new delay will start after which the
PGOOD will go high again.
The switching frequency of the ISL6446 is determined by the
external resistor placed from the RT pin to SGND. See Figure 22
for a graph of Frequency vs RT Resistance. Use Equation 4 to
calculate the approximate RT resistor value for the desired
switching frequency. The typical resistance for 100kHz operation
is 163kΩ. Running at both high frequency and high VIN voltages
is not recommended, due to the increased power dissipation
on-chip (mostly from the internal VCC regulator, which supplies
gate drivers). The user should check the maximum acceptable IC
temperature, based on their particular conditions.
f SW – 1.093
R T =  ----------------
 11290
(EQ. 4)
300
100
RT VALUE (kΩ)
FIGURE 20. SOFT-START WITH PREBIAS
50
30
10
3
100k
200k
500k
1M
2M
SWITCHING FREQUENCY (Hz)
FIGURE 22. FREQUENCY vs RT RESISTOR
The PGOOD delay is inversely proportional to the clock frequency.
If the clock is running as slow as 524kHz, the delay will be
125ms long. There is no way to adjust the PGOOD delay
independently of the clock.
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October 12, 2015
ISL6446
Output Regulation
VOUT
FB
EA
+
RUP
COMP
0.6V
VOUT must be greater than 0.6V and 2 resistors are needed, and
their accuracy directly affect the regulator tolerance.
(EQ. 5)
Use Equation 6 to choose the resistor values. RUP is part of the
compensation network for the switchers and should be selected
to be compatible; 1kΩ to 5kΩ is a good starting value. Find FB
from the “Electrical Specifications” table on page 7 (for the right
condition), plug in the desired value for VOUT, and solve for RLOW.
(EQ. 6)
The maximum duty cycle of the ISL6446 approaches 100% at
low frequency, but falls off at higher frequency; see the
“Electrical Specifications” table on page 7. In addition, there is a
minimum UGATE pulse width, in order to properly sense
overcurrent. The two switchers are 180° out of phase.
Linear Regulator
The linear regulator controller is a transconductance amplifier
with a nominal gain of 2A/V. The N-Channel MOSFET output
buffer can sink a minimum of 50mA.
The reference voltage is 0.6V. With 0V differential at its input, the
controller sinks 21mA of current. For better load regulation, it is
recommended that the resistor from the LDO input to the base of
the PNP (or gate of the PFET) is set so that the sink current at G4
pin is within 9mA to 31mA over the entire load and temperature
range.
An external PNP transistor or P-Channel MOSFET pass device can
be used. The dominant pole for the loop can be placed at the
base of the PNP (or gate of the PFET), as a capacitor from
emitter-to-base (source to gate of a PFET). Better load transient
response is achieved however, if the dominant pole is placed at
the output with a capacitor to ground at the output of the
regulator.
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13
40
30
20
10
0.6
0.62
0.63
0.61
FEEDBACK VOLTAGE (V)
0.64
0.65
FIGURE 24. LINEAR CONTROLLER GAIN
FIGURE 23. OUTPUT REGULATION
FB  R UP
R LOW = -----------------------------V OUT – FB
50
0
0.59
RLOW
R LOW
FB = V OUT  ----------------------------------R UP + R LOW
ERROR AMPLIFIER SINK
CURRENT (mA)
60
Figure 23 shows the generic feedback resistor circuit for any of
the two PWM VOUT’s; the VOUT is divided down to equal the
reference. All three use a 0.6V internal reference (check the
“Electrical Specifications” Table on page 6 for the exact reference
value at 24V). The RUP is connected to the VOUT; the RLOW to
GND; the common point goes to the FB pin.
Protection Mechanisms
OCP - (Function independent for both PWM). The overcurrent
function protects the PWM converter from a shorted output by
using the upper MOSFET’s ON-resistance, rDS(ON) to monitor the
current. This method enhances the converter’s efficiency and
reduces cost by eliminating a current sensing resistor. The
overcurrent function latches off the outputs to provide fault
protection. A resistor connected to the drain of the upper MOSFET
and OCSET pin programs the overcurrent trip level. The PHASE
node voltage will be compared against the voltage on the OCSET
pin, while the upper MOSFET is on. A current (typically 110µA) is
pulled from the OCSET pin to establish the OCSET voltage. If
PHASE is lower than OCSET while the upper MOSFET is on then an
overcurrent condition is detected for that clock cycle. The upper
gate pulse is immediately terminated and a counter is
incremented. If an overcurrent condition is detected for 32
consecutive clock cycles, the ISL6446 output is latched off with
gate drivers three-stated. The switcher will restart when the SS/EN
pin is externally driven below 1V, or if power is recycled to the chip.
During soft-start, both pulse termination current limiting and the
32-cycle counter are enabled.
UVP - (Function independent for both PWM). If the voltage on the
FB pin falls to 82% (typical) of the reference voltage for 8
consecutive PWM cycles, then the circuit enters into soft-start
hiccup mode. During hiccup, the external capacitor on the SS/EN
pin is discharged, then released and a soft-start cycle is initiated.
The UVP comparator is separate from the one sensing for PGOOD,
which should have already detected a problem, before the UVP
trips.
OVP - (Function independent for both PWM). OVP function is
enabled after the soft-start has finished. If voltage on the FB pin
rises to 116% (typical) of the reference voltage, the lower gate
driver is turned on continuously. If the overvoltage condition
continues for 32 consecutive PWM cycles, then that output is
latched off with the gate drivers three-stated. The capacitor on the
SS/EN pin will not be discharged. The switcher will restart when
the SS/EN pin is externally driven below 1V, or if power is recycled
to the chip. The OVP comparator is separate from the one sensing
for PGOOD, which should have already detected a problem,
before the OVP trips.
FN7944.2
October 12, 2015
ISL6446
Application Guidelines
L OUT  I TRAN
t FALL = --------------------------------------V OUT
PWM Controller
DISCUSSION
The PWM must be compensated such that it achieves the
desired transient performance goals, stability and DC regulation
requirements.
The first parameter that needs to be chosen is the switching
frequency, fSW. This decision is based on the overall size
constraints and the frequency plan of the end equipment.
Smaller space requires higher frequency. This allows the output
inductor, input capacitor bank and output capacitor bank to be
reduced in size and/or value. The power supply must be designed
such that the frequency and its distribution over component
tolerance, time and temperature causes minimal interference in
RF stages, IF stages, PLL loops, mixers, etc.
INDUCTOR SELECTION
The output inductor is selected to meet the output voltage ripple
requirements and minimize the converter’s response time to the
load transient. The inductor value determines the converter’s
ripple current, and the ripple voltage is a function of the ripple
current. The ripple current and voltage are approximated by the
following Equations 7 and 8, where ESR is the output
capacitance ESR value.
V IN - V OUT V OUT
I = --------------------------------  ---------------f SW  L
V IN
(EQ. 7)
VOUT = I x ESR
(EQ. 8)
Increasing the value of inductance reduces the ripple current and
voltage. However, the large inductance value reduces the
converter’s response time to a load transient (and usually
increases the DCR of the inductor, which decreases the
efficiency). Increasing the switching frequency (fSW) for a given
inductor also reduces the ripple current and voltage.
One of the parameters limiting the converter’s response to a load
transient is the time required to change the inductor current.
Given a sufficiently fast control loop design, the ISL6446 will
provide either 0% or 100% duty cycle in response to a load
transient. The response time is the time required to slew the
inductor current from an initial current value to the transient
current level. During this interval, the difference between the
inductor current and the transient current level must be supplied
by the output capacitor. Minimizing the response time can
minimize the output capacitance required.
The response time to a transient is different for the application of
load and the removal of load. Equations 9 and 10 give the
approximate response time interval for application and removal
of a transient load:
L OUT  I TRAN
t RISE = --------------------------------------V IN – V OUT
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(EQ. 9)
(EQ. 10)
Where: ITRAN is the transient load current step, tRISE is the
response time to the application of load, and tFALL is the
response time to the removal of load. With a +5V input source,
the worst case response time can be either at the application or
removal of load and dependent upon the output voltage setting.
Be sure to check both of these equations at the minimum and
maximum output levels for the worst case response time.
Finally, check that the inductor Isat rating is sufficiently above the
maximum output current (DC load plus ripple current).
OUTPUT CAPACITOR SELECTION
An output capacitor is required to filter the output and supply the
load transient current. The filtering requirements are a function
of the switching frequency and the ripple current. The load
transient requirements are a function of the slew rate (di/dt) and
the magnitude of the transient load current. These requirements
are generally met with a mix of capacitors and careful layout.
Modern microprocessors produce transient load rates above
1A/ns. High frequency capacitors initially supply the transient
and slow the current load rate seen by the bulk capacitors. The
bulk filter capacitor values are generally determined by the ESR
(effective series resistance) and voltage rating requirements
rather than actual capacitance requirements.
High frequency decoupling capacitors should be placed as close to
the power pins of the load as physically possible. Be careful not to
add inductance in the circuit board wiring that could cancel the
usefulness of these low inductance components. Consult with the
manufacturer of the load on specific decoupling requirements. Keep
in mind that not all applications have the same requirements; some
may need many ceramic capacitors in parallel; others may need
only one.
Use only specialized low-ESR capacitors intended for
switching-regulator applications for the bulk capacitors. The bulk
capacitor’s ESR will determine the output ripple voltage and the
initial voltage drop after a high slew-rate transient. An aluminum
electrolytic capacitor's ESR value is related to the case size with
lower ESR available in larger case sizes. However, the Equivalent
Series Inductance (ESL) of these capacitors increases with case
size and can reduce the usefulness of the capacitor to high slew
rate transient loading. Unfortunately, ESL is not a specified
parameter. Work with your capacitor supplier and measure the
capacitor’s impedance with frequency to select a suitable
component. In most cases, multiple electrolytic capacitors of
small case size perform better than a single large case capacitor.
INPUT CAPACITOR SELECTION
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic capacitors for
high frequency decoupling and bulk capacitors to supply the
current needed each time Q1 (upper FET) turns on. Place the
small ceramic capacitors physically close to the MOSFETs and
between the drain of Q1 and the source of Q2 (lower FET).
The important parameters for the bulk input capacitor are the
voltage rating and the RMS current rating. For reliable operation,
select the bulk capacitor with voltage and current ratings above
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FN7944.2
October 12, 2015
ISL6446
the maximum input voltage and largest RMS current required by
the circuit. The capacitor voltage rating should be at least 1.25x
greater than the maximum input voltage and a voltage rating of
1.5x is a conservative guideline. The RMS current rating
requirement for the input capacitor of a buck regulator is
approximately 1/2 the DC load current.
Feedback Compensation Equations
This section highlights the design consideration for a voltage
mode controller requiring external compensation. To address a
broad range of applications, a type-3 feedback network is
recommended (see Figure 25).
C2
SWITCHER MOSFET SELECTION
VIN for the ISL6446 has a wide operating voltage range allowed,
so both FETs should have a source-to-drain breakdown voltage
(VDS) above the maximum supply voltage expected; 20V or 30V
are typical values available.
The ISL6446 gate drivers (UGATEx and LGATEx) were designed to
drive single FETs (for up to ~10A of load current) or smaller dual
FETs (up to 4A). Both sets of drivers are sourced by the internal VCC
regulator (unless VIN = VCC = 5V, in which case the gate driver
current comes from the external 5V supply). The maximum current
of the regulator (ICC_max) is listed in the “Electrical Specifications”
table on page 6; this may limit how big the FETs can be. In addition,
the power dissipation of the regulator is a major contributor to the
overall IC power dissipation (especially as Cin of the FET or VIN or
fSW increases).
Since VCC is around 5V, that affects the FET selection in two
ways. First, the FET gate-source voltage rating (VGS) can be as
low as 12V (this rating is usually consistent with the 20V or 30V
breakdown chosen above). Second, the FETs must have a low
threshold voltage (around 1V), in order to have its rDS(ON) rating
at VGS = 4.5V in the 10mΩ to 40mΩ range that is typically used
for these applications. While some FETs are also rated with gate
voltages as low as 2.7V, with typical thresholds under 1V, these
can cause application problems. As LGATE shuts off the lower
FET, it does not take much ringing in the LGATE signal to turn the
lower FET back on, while the upper FET is starting to turn on,
causing some shoot-through current. Therefore, avoid FETs with
thresholds below 1V.
If the power efficiency of the system is important, then other FET
parameters are also considered. Efficiency is a measure of power
losses from input to output, and it contains two major
components: losses in the IC (mostly in the gate drivers) and
losses in the FETs. For low duty cycle applications (such as 12V in
to 1.5V out), the upper FET is usually chosen for low gate charge,
since switching losses are key, while the lower FET is chosen for
low rDS(ON), since it is on most of the time. For high duty cycles
(such as 5V in to 3.3V out), the opposite may be true.
C1
R2
COMP
FB
C3
R1
ISL6446
R3
VOUT
FIGURE 25. COMPENSATION CONFIGURATION FOR ISL6446 CIRCUIT
Figure 26 highlights the voltage-mode control loop for a
synchronous-rectified buck converter, applicable to the ISL6446
circuit. The output voltage (VOUT) is regulated to the reference
voltage, VREF. The error amplifier output (COMP pin voltage) is
compared with the oscillator (OSC) modified sawtooth wave to
provide a pulse-width modulated wave with an amplitude of VIN
at the PHASE node. The PWM wave is smoothed by the output
filter (L and C). The output filter capacitor bank’s equivalent
series resistance is represented by the series resistor E.
The modulator transfer function is the small-signal transfer
function of VOUT /VCOMP. This function is dominated by a DC gain,
given by dMAXVIN /VOSC , and shaped by the output filter, with a
double pole break frequency at FLC and a zero at FCE . For the
purpose of this analysis, L and D represent the channel
inductance and its DCR, while C and E represent the total output
capacitance and its equivalent series resistance.
C2
COMP
R2
C3
R3
C1
-
E/A
+
FB
Ro
R1
VREF
VOUT
OSCILLATOR
VIN
PWM
CIRCUIT
VOSC
UGATE
HALF-BRIDGE
DRIVE
L
D
PHASE
LGATE
ISL6446
C
E
EXTERNAL CIRCUIT
FIGURE 26. VOLTAGE-MODE BUCK CONVERTER COMPENSATION
DESIGN
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ISL6446
(EQ. 11)
1
F CE = -----------------------2  C  E
(EQ. 12)
The compensation network consists of the error amplifier
(internal to the ISL6446) and the external R1 through R3, C1
through C3 components. The goal of the compensation network
is to provide a closed loop transfer function with high 0dB
crossing frequency (F0; typically 0.1 to 0.3 of fSW) and adequate
phase margin (better than 45°). Phase margin is the difference
between the closed loop phase at F0dB and 180°. The equations
that follow relate the compensation network’s poles, zeros and
gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in
Figure 26. Use the following guidelines for locating the poles and
zeros of the compensation network:
1. Select a value for R1 (1kΩ to 5kΩ, typically). Calculate the
value for R2 for desired converter bandwidth (F0). If setting
the output voltage via an offset resistor connected to the FB
pin, RO in Figure 26, the design procedure can be followed as
presented in Equation 13.
(EQ. 13)
2. Calculate C1 such that FZ1 is placed at a fraction of the FLC, at
0.1 to 0.75 of FLC (to adjust, change the 0.5 factor to desired
number). The higher the quality factor of the output filter and/or
the higher the ratio FCE/FLC, the lower the FZ1 frequency (to
maximize phase boost at FLC).
1
C 1 = ----------------------------------------------2  R 2  0.5  F LC
(EQ. 14)
3. Calculate C2 such that FP1 is placed at FCE.
C1
C 2 = -------------------------------------------------------2  R 2  C 1  F CE – 1
(EQ. 15)
4. Calculate R3 such that FZ2 is placed at FLC. Calculate C3 such
that FP2 is placed below fSW (typically, 0.5 to 1.0 times fSW).
fSW represents the switching frequency. Change the
numerical factor to reflect desired placement of this pole.
Placement of FP2 lower in frequency helps reduce the gain of
the compensation network at high frequency, in turn reducing
the HF ripple component at the COMP pin and minimizing
resultant duty cycle jitter.
R1
R 3 = -------------------f SW
----------–1
F LC
1
C 3 = ----------------------------------------------2  R 3  0.7  f SW
(EQ. 17)
1 + s  f   R2  C1
G FB  f  = ---------------------------------------------------- 
s  f   R1   C1 + C2 
1 + s  f    R1 + R3   C3
 -------------------------------------------------------------------------------------------------------------------------
 C1  C2  
 1 + s  f   R 3  C 3    1 + s  f   R2   --------------------- 

 C 1 + C 2 
(EQ. 18)
G CL  f  = G MOD  f   G FB  f 
(EQ. 19)
Where:
s  f  = 2  f  j
COMPENSATION BREAK FREQUENCY EQUATIONS
1
F Z1 = ------------------------------2  R 2  C 1
(EQ. 20)
1
F Z2 = ------------------------------------------------2   R 1 + R 3   C 3
(EQ. 21)
1
F P1 = --------------------------------------------C1  C2
2  R 2  --------------------C1 + C2
(EQ. 22)
1
F P2 = ------------------------------2  R 3  C 3
(EQ. 23)
Figure 27 shows an asymptotic plot of the DC/DC converter’s gain vs
frequency. The actual Modulator Gain has a high gain peak
dependent on the quality factor (Q) of the output filter, which is not
shown. Using the previously mentioned guidelines should yield a
compensation gain similar to the curve plotted. The open loop error
amplifier gain bounds the compensation gain. Check the
compensation gain at FP2 against the capabilities of the error
amplifier. The closed loop gain, GCL, is constructed on the log-log
graph of Figure 27 by adding the modulator gain, GMOD (in dB), to
the feedback compensation gain, GFB (in dB). This is equivalent to
multiplying the modulator transfer function and the compensation
transfer function and then plotting the resulting gain.
FZ1 FZ2
FP1
FP2
MODULATOR GAIN
COMPENSATION GAIN
CLOSED LOOP GAIN
OPEN LOOP E/A GAIN
(EQ. 16)
It is recommended a mathematical model is used to plot the
loop response. Check the loop gain against the error amplifier’s
open-loop gain. Verify phase margin results and adjust as
necessary. The following equations describe the frequency
response of the modulator (GMOD), feedback compensation
(GFB) and closed-loop response (GCL):
R2
20 log  --------
R1
d MAX  V
IN
20 log ----------------------------V OSC
0
GFB
GCL
LOG
V OSC  R 1  F 0
R 2 = --------------------------------------------d MAX  V IN  F LC
d MAX  V IN
1 + sf  E  C
G MOD  f  = ------------------------------  ---------------------------------------------------------------------------------------2
V OSC
1 + s f  E + D  C + s f  L  C
GAIN
1
F LC = --------------------------2  L  C
GMOD
LOG
FLC
FCE
F0
FREQUENCY
FIGURE 27. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
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October 12, 2015
ISL6446
Layout Considerations
As in any high frequency switching converter, layout is very
important. Switching current from one power device to another can
generate voltage transients across the impedances of the
interconnecting bond wires and circuit traces. These
interconnecting impedances should be minimized by using wide,
short printed circuit traces. The critical components should be
located as close together as possible using ground plane
construction or single point grounding.
Figure 28 shows the critical power components of the converter.
To minimize the voltage overshoot, the interconnecting wires
indicated by heavy lines should be part of ground or power plane
in a printed circuit board. The components shown in Figure 28
should be located as close together as possible. Please note that
the capacitors CIN and COUT each represent numerous physical
capacitors. Locate the ISL6446 within 1 inch of the MOSFETs, Q1
and Q2. The circuit traces for the MOSFETs’ gate and source
connections from the ISL6446 must be sized to handle up to 2A
peak current.
BOOT
CBOOT C
IN
PGND
Q1 L
OUT
ISL6446
SS
PHASE
+VIN
RT
Q2 COUT
VOUT
VIN
RT
SGND PGND
CSS
CVIN
PGND
SGND
FIGURE 29. PRINTED CIRCUIT BOARD POWER AND GROUND
PLANES OR ISLANDS
PIN 18
PGND
PIN 19
VCC
CVCC
FIGURE 30. DECOUPLING CAPACITOR
Q1
Figure 29 shows the circuit traces that require additional layout
consideration. Use single point and ground plane construction for
the circuits shown. Locate the RT resistor as close as possible to
the RT pin and the SGND pin. Provide a decoupling capacitor,
CVCC, between VCC and PGND pins and place it as close to the
VCC and PGND pins as possible (shown in Figure 30).
Q2
LOUT
CIN
VOUT
COUT
PGND
RETURN
FIGURE 28. PRINTED CIRCUIT BOARD POWER AND GROUND
PLANES OR ISLANDS
LOAD
PHASE
LGATE
+VIN
VCC
VIN
ISL6446
UGATE
CVCC
LOAD
A stable control loop has a gain crossing with close to a
-20dB/decade slope and a phase margin greater than 45°.
Include worst case component variations when determining
phase margin. The mathematical model presented makes a
number of approximations and is generally not accurate at
frequencies approaching or exceeding half the switching
frequency. When designing compensation networks, select
target crossover frequencies in the range of 10% to 30% of the
switching frequency, fSW.
For each switcher, minimize any leakage current paths on the
SS/EN pin and locate the capacitor, CSS close to the SS/EN pin
because the internal current source is only 30µA. All of the
compensation network components for each switcher should be
located near the associated COMP and FB pins. Locate the
capacitor, CBOOT as close as practical to the BOOT and PHASE
pins (but keep the noisy PHASE plane away from the IC (except
for the PHASE pin connection).
The OCSET circuits (see Figure 4 on page 5) should have a
separate trace from the upper FET to the OCSET R and C; that will
more accurately sense the VIN at the FET than just tying them to
the VIN plane. The OCSET R and C should be placed near the IC
pins.
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ISL6446
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to web to make sure you
have the latest revision.
DATE
REVISION
CHANGE
October 12, 2015
FN7944.2
Updated “Layout Considerations” on page 17.
October 15, 2013
FN7944.1
Figure 1 on page 1: Changed CVCC from common ground tied to PGND earth ground.
Figure 4 on page 5: Changed CVIN from common ground tied to PGND earth ground.
Figure 29 on page 17: Changed CVCC from common ground tied to PGND earth ground.
“Absolute Maximum Ratings” on page 6 ESD rating: changed HBM from 2000V to 2500V and changed MM
from 200V to 150V.
Converted to new POD format. Added land pattern.
July 10, 2012
FN7944.0
Initial Release.
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FN7944.2
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ISL6446
Package Outline Drawing
M24.15
24 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE (QSOP/SSOP) 0.150” WIDE BODY
Rev 3, 2/13
24
6.19
5.80
INDEX
3.98
3.81
AREA
5
4
0.25(0.010) M
B M
-B-
1
TOP VIEW
DETAIL “X”
SEATING PLANE
-A-
8.74
8.55
3
1.75
1.35
GAUGE
PLANE
-C-
SIDE VIEW 1
1.27
0.41
0.25
0.10
0.635 BSC
7
0.30
0.20
0.25
0.010
0.10(0.004)
0.49
x 45° 5
0.26
0.17(0.007) M C A M B S
7.11
8°
0°
5.59
1.54
SIDE VIEW 2
0.25
0.18
4.06
0.38
0.635
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication
Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Package length does not include mold flash, protrusions or gate burrs. Mold
flash,
protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side.
4. Package width does not include interlead flash or protrusions. Interlead flash and
protrusions shall not exceed 0.25mm (0.010 inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature
must be located within the crosshatched area.
6. Terminal numbers are shown for reference only.
7. Lead width does not include dambar protrusion. Allowable dambar protrusion
shall be 0.10mm (0.004 inch) total in excess of “B” dimension at maximum material condition.
8. Controlling dimension: MILLIMETER.
TYPICAL RECOMMENDED LAND PATTERN
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October 12, 2015