DATASHEET

ISL59830
S
SI G N
W DE
E
N
R
D FO
9833
ENDE R TO ISL5 RADE
M
M
Data
UPGSheet
E FE
R EC O
NOT PLEASE R PATIBLE
C OM
A PIN
F OR
®
May 4, 2006
True Single Supply Video Driver
Features
The ISL59830 is a revolutionary device that allows true singlesupply operation of video amplifiers. The device runs off a
single 3.3V supply and generates the required negative
voltage internally. This allows for DC-accurate coupling of
video onto a 75Ω double-terminated line. Since the buffers
have an integrated 6dB gain, no external gain setting resistors
are required. An input reference voltage can be supplied to
shift the analog video level down by an amount equal to the
reference (typically 0.6V).
• Triple single-supply buffer
Ordering Information
• 200MHz -3dB bandwidth
PART NUMBER
PART
TAPE &
MARKING REEL
PACKAGE
PKG.
DWG. #
• Operates from single +3.3V supply
• No output DC blocking capacitor needed
• Fixed gain of 2 output buffer
• Output three-statable
• Enable/disable function
• 50MHz 0.1dB bandwidth
• Pb-free plus anneal available (RoHS compliant)
ISL59830IA
59830IA
-
16 Ld QSOP M16.15A
Applications
ISL59830IA-T7
59830IA
7”
16 Ld QSOP M16.15A
• Driving video
ISL59830IA-T13
59830IA
13”
16 Ld QSOP M16.15A
Pinout
ISL59830IAZ
(See Note)
59830IAZ
-
16 Ld QSOP M16.15A
(Pb-Free)
ISL59830IAZ-T7
(See Note)
59830IAZ
7”
16 Ld QSOP M16.15A
(Pb-Free)
ISL59830IAZ-T13 59830IAZ
(See Note)
13”
16 Ld QSOP M16.15A
(Pb-Free)
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100%
matte tin plate termination finish, which are RoHS compliant and
compatible with both SnPb and Pb-free soldering operations. Intersil
Pb-free products are MSL classified at Pb-free peak reflow
temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
ISL59830
(16 LD QSOP)
TOP VIEW
RIN 1
16 ROUT
GIN 2
15 GOUT
BIN 3
14 BOUT
REF 4
13 VCC
VEE 5
12 EN
GND 6
11 VCC
VEEOUT 7
DGND 8
1
FN7489.6
10 NC
9 DVCC
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright © Intersil Americas Inc. 2005, 2006. All Rights Reserved.
All other trademarks mentioned are the property of their respective owners.
ISL59830
Absolute Maximum Ratings (TA = 25°C)
VCC, Supply Voltage between VS and GND . . . . . . . . . . . . . . . . .5V
VIN, VREF . . . . . . . . . . . . . . . . . . . . . . . . . . . .VCC+0.3V, VEE-0.3V
Voltage between VIN and VREF . . . . . . . . . . . . . . . . . . . . . . . . . .±2V
Maximum Continuous Output Current . . . . . . . . . . . . . . . . . . . 30mA
Operating Temperature . . . . . . . . . . . . . . . . . . . . . . .-40°C to +85°C
Maximum Die Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Lead Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
Power Dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Curves
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
IMPORTANT NOTE: All parameters having Min/Max specifications are guaranteed. Typical values are for information purposes only. Unless otherwise noted, all tests
are at the specified temperature and are pulsed tests, therefore: TJ = TC = TA
AC Electrical Specifications
PARAMETER
BW -3dB
VCC = DVCC = +3.3V, REF = GND, TA = 25°C, RL = 150Ω, unless otherwise specified.
DESCRIPTION
3dB Bandwidth
CONDITIONS
MIN
TYP
MAX
UNIT
VOUT = 200mVPP
200
MHz
VOUT = 2VPP
100
MHz
50
MHz
BW 0.1dB
0.1dB Bandwidth
VOUT = 2VPP
SR
Slew Rate
VOUT = 2VPP
dG
Differential Gain
0.07
%
dP
Differential Phase
0.06
°
XT
Hostile Crosstalk
6MHz
-90
dB
I
Input to Output Isolation
6MHz
-70
dB
VN
Input Noise Voltage
20
nV/√Hz
Fcp
Charge Pump Switch Frequency
168
MHz
Load Reg
VRIPPLE
500
IEE = 0mA to 10mA
12
Output Amp Ripple Voltage
With Bead Core to DVCC
DC Electrical Specifications
PARAMETER
V/µs
60
mV
30
mV
10
mV
VCC = DVCC = +3.3V, REF = GND, TA = 25°C, RL = 150Ω, unless otherwise specified.
DESCRIPTION
CONDITIONS
MIN
TYP
3.0
MAX
UNIT
3.6
V
1.5
%
V+
Supply Range
VG%
Gain Error
RL = 150Ω, VIN = +2.5V to -1V
ΔG
Gain Matching
RL = 150Ω
RIN
Input Resistance
VIN = 0V to 1.5V
1.0
1.7
15
MΩ
VOS
Output Offset Voltage
VREF = 0
-25
7
+25
mV
IOUT +
Output Current
RL = 10Ω, VIN = 1.2V
50
IOUT -
Output Current
RL = 10Ω, VIN = -0.3V
ZOUT
Output Impedance
Enabled
1
Ω
Three-stated
10
MΩ
90
dB
PSRR
Power Supply Rejection Ratio
IS
Supply Current
RREF
Input Reference Resistor
2
0.5
%
mA
-18
60
Amp Enabled
120
Amp Disabled
80
4
5
150
mA
mA
mA
6
kΩ
FN7489.6
May 4, 2006
ISL59830
Pin Descriptions
PIN NUMBER
PIN NAME
1
RIN
PIN FUNCTION
EQUIVALENT CIRCUIT
Analog input
VCC
VEE
CIRCUIT 1
2
GIN
Analog input
Reference Circuit 1
3
BIN
Analog input
Reference Circuit 1
4
REF
Reference input
VCC
RIN
GIN
BIN
ROUT
GOUT
BOUT
+
-
3
REF
VEE
CIRCUIT 2
5
VEE
Chip substrate
VCC
VEE OUT
- +
DVCC
VEE
CHARGE
PUMP
DGND
CIRCUIT 3
6
GND
Analog ground
7
VEE OUT
Charge pump output
Reference Circuit 3
8
DGND
Charge pump ground
Reference Circuit 3
9
DVCC
Charge pump supply voltage
Reference Circuit 3
10
NC
11, 13
VCC
12
EN
Not connected
Positive power supply
Chip enable
VCC
VEE
CIRCUIT 4
3
FN7489.6
May 4, 2006
ISL59830
Pin Descriptions (Continued)
PIN NUMBER
PIN NAME
14
BOUT
PIN FUNCTION
EQUIVALENT CIRCUIT
Analog output
VCC
VEE
CIRCUIT 5
15
GOUT
Analog output
Reference Circuit 5
16
ROUT
Analog output
Reference Circuit 5
Typical Performance Curves
5
AV=+2
CL=0pF
2
AV=+2
RL=500Ω
1
NORMALIZED GAIN (dB)
NORMALIZED GAIN (dB)
3
1kΩ
0
500Ω
-1
150Ω
-2
9pF
4.7pF
3
2.2pF
1
0pF
-1
-3
75Ω
-3
1M
10M
100M
-5
100K
1G
FREQUENCY (Hz)
100M
1G
FREQUENCY (Hz)
FIGURE 1. GAIN vs FREQUENCY FOR VARIOUS RLOAD
FIGURE 2. GAIN vs FREQUENCY FOR VARIOUS CLOAD
5
300
AV=+2
CL=0pF
RL=500Ω
0
-5
-10
-15
-20
-25
AV=+2
RL=500Ω
GAIN ROLL-OFF (MHz)
NORMALIZED OUTPUT (dB)
10M
1M
240
-3dB ROLL-OFF
180
120
60
-0.1dB ROLL-OFF
-30
-35
1
100
200
300
400
500
FREQUENCY (MHz)
FIGURE 3. VREF PIN OUTPUT FREQUENCY RESPONSE
4
0
2.25
2.8
3.35
3.9
4.45
5
TOTAL SUPPLY VOLTAGE, VCC - VEE (V)
FIGURE 4. GAIN ROLL-OFF
FN7489.6
May 4, 2006
ISL59830
Typical Performance Curves (Continued)
-30
1.6
AV=+2
-40 RL=500Ω
AV=+2
RL=500Ω
CL=3.9pF
-50
CROSS TALK (dB)
PEAKING (dB)
1.2
0.8
0.4
-60
ENABLED
-70
-80
DISABLED
-90
-100
-110
0
2.2
2.6
2.4
2.8
3
3.2
3.4
3.6
3.8
-120
100K
4
1M
FIGURE 6. CROSS TALK CHANNEL TO CHANNEL (TYPICAL)
FIGURE 5. PEAKING vs SUPPLY VOLTAGE
-20
120
SUPPLY CURRENT (mA)
AV=+2
-30 RL=500Ω
ISOLATION (dB)
-40
-50
-60
-70
-80
-90
10M
1M
100M
AV=+2
RL=500Ω
100
80
60
40
20
0
1G
1
1.5
FREQUENCY (Hz)
2.5
2
3
3.5
SUPPLY VOLTAGE (V)
FIGURE 7. INPUT TO OUTPUT ISOLATION vs FREQUENCY
FIGURE 8. SUPPLY CURRENT vs SUPPLY VOLTAGE
200
95
AV=+2
RL=500Ω
VCL=3.3V
160
AV=+2
RL=500Ω
120
80
40
SUPPLY CURRENT (mA)
-3dB
BANDWIDTH (MHz)
1G
FREQUENCY (Hz)
SUPPLY VOLTAGE (V)
-100
100K
100M
10M
90
85
80
-0.1dB
0
25
85
55
115
TEMPERATURE (°C)
FIGURE 9. BANDWIDTH vs TEMPERATURE
5
145
75
25
55
85
115
145
TEMPERATURE (°C)
FIGURE 10. SUPPLY CURRENT vs TEMPERATURE
FN7489.6
May 4, 2006
ISL59830
Typical Performance Curves (Continued)
100
-10
-30
PSRR (dB)
IMPEDANCE (Ω)
10
1
-50
PSRR-70
PSRR+
0.1
-90
0.01
10K
100K
1M
-110
1K
100M
10M
10K
FREQUENCY (Hz)
10M
100M
FIGURE 12. POWER SUPPLY REJECTION RATIO vs
FREQUENCY
-30
HARMONIC DISTORTION (dBc)
1K
VOLTAGE NOISE (nV/√Hz),
CURRENT NOISE (pA/√Hz)
1M
FREQUENCY (Hz)
FIGURE 11. OUTPUT IMPEDANCE vs FREQUENCY
100
eN
10
IN+
1
IN0.1
10
100K
-40
THD
-50
-60
-70
2ND HD
3RD HD
-80
-90
-100
100
1K
10K
100K
1M
0
10M
10
20
30
40
FUNDAMENTAL FREQUENCY (MHz)
FREQUENCY (Hz)
FIGURE 13. VOLTAGE AND CURRENT NOISE vs FREQUENCY
FIGURE 14. HARMONIC DISTORTION vs FREQUENCY
-30
-40
THD
FIN=10MHz
-60
DIFFERENTIAL
GAIN (%)
THD (dBc)
-50
-70
THD
FIN=1MHz
-80
-90
0.5
1
1.5
2
2.5
OUTPUT VOLTAGE (VP-P)
FIGURE 15.
6
3
3.5
0
-0.02
-0.04
-0.06
-0.08
IRE
FIGURE 16. DIFFERENTIAL GAIN
FN7489.6
May 4, 2006
ISL59830
VOLTS (500mV/DIV)
DIFFERENTIAL
PHASE (°)
Typical Performance Curves (Continued)
0
-0.02
-0.04
-0.06
-0.08
TIME (2µs/DIV)
IRE
FIGURE 18. DISABLE TIME
VOLTS (50mV/DIV)
VOLTS (500mV/DIV)
FIGURE 17. DIFFERENTIAL PHASE
TIME (200ns/DIV)
FIGURE 20. SMALL SIGNAL RISE & FALL TIMES
VOLTS (10mV/DIV)
VOLTS (500mV/DIV)
FIGURE 19. ENABLE TIME
TIME (10ns/DIV)
TIME (10ns/DIV)
FIGURE 21. LARGE SIGNAL RISE & FALL TIMES
7
TIME (20ns/DIV)
FIGURE 22. AMP OUTPUT NOISE (CHARGE PUMP
OSCILLATION)
FN7489.6
May 4, 2006
ISL59830
Typical Performance Curves (Continued)
1.6
BACKDRIVE CURRENT (mA)
OUTPUT RANGE (V)
3.25
3
2.75
2.5
50
AV=+2
CL=3.9pF
450
250
650
850
BACKDRIVE ACROSS 5Ω RESISTOR
TYPICAL CHANNEL
1.2
VCC = 3.3V
0.8
0.4
0
0
1050
2
1
3
5
BACKDRIVE VOLTAGE (V)
LOAD RESISTANCE (Ω)
FIGURE 23. MAXIMUM OUTPUT MAGNITUDE vs LOAD
RESISTANCE
FIGURE 24. BACKDRIVE VOLTAGE vs CURRENT
AMP DISABLED OUTPUT LOADING
Block Diagram
JEDEC JESD51-3 LOW EFFECTIVE THERMAL
CONDUCTIVITY TEST BOARD
VCC
1.4
POWER DISSIPATION (W)
4
1.2
1
791mW
0.8
QS
OP
θJ
16
A =1
58
°C
/W
0.6
0.4
Y
RIN
+
ROUT
6dB
REFERENCE
0.2
Pb
GIN
GOUT
+
6dB
0
0
25
50
75 85 100
125
-
150
AMBIENT TEMPERATURE (°C)
FIGURE 25. PACKAGE POWER DISSIPATION vs AMBIENT
TEMPERATURE
Pr
BIN
+
BOUT
6dB
-
DVCC
JEDEC JESD51-7 HIGH EFFECTIVE THERMAL
CONDUCTIVITY TEST BOARD
CHARGE
PUMP
1.8
POWER DISSIPATION (W)
1.6
VEE-OUT
VEE
1.4
VOUT = 2VIN - VREFERENCE
1.116W
1.2
1
θJ
QS
A =1
0.8
OP
12
0.6
16
°C
/W
0.4
0.2
0
0
25
75 85
50
100
125
150
AMBIENT TEMPERATURE (°C)
FIGURE 26. PACKAGE POWER DISSIPATION vs AMBIENT
TEMPERATURE
8
FN7489.6
May 4, 2006
ISL59830 + DC-Restore Solution
1 IN1
IN2 16
2 COM1
COM2 15
3 NC1
NC2 14
9
4 V-
R7
2kΩ
V+ 13
5 GND
NC 12
(No Connect)
6 NC4
NC3 11
7 COM4
COM3 10
YO
R1
75Ω
8 IN4
R2
75Ω
R3
75Ω
R9
2kΩ
R11
499Ω
C4
0.1µF
C5
0.1µF
C6
0.1µF
CN = Option for lower
charge pump noise
R10
2kΩ
1 RIN
ROUT 16
2 GIN
GOUT 15
3 BIN
BOUT 14
4 REF
VCC 13
5 VEE
EN 12
6 GND
VCC 11
REF
MMBP
3904
C7
1.0µF
VEE (-1.6V)
1kΩ
R12
R8
D1
1N4148
C11
0.1µF
(or similar)
7 VEEOUT
8 DGND
R5
75Ω
R6
75Ω
C12
20pF
C13
20pF
C1
0.1µF
ENABLE
2
1
NC 10
DVCC 9
75Ω
VCC
VCC
REFERENCE
CONTROL
R4
C15
0.1µF
3
C14
20pF
VCC
YO
Pb
Pr
VCC
+ C16
VCC
1µF
GND
ISL59830
C4
0.1µF
1 COMP
VDD 8
2 COMP
OUT 7
SYNC OUT
VIDEO IN
3 VSYNC
OUT
C10
0.1µF
4 GND
RESET 6
BACK
PORCH 5
OUT
EL1881
C8
0.1µF
R13
C9
0.1µF 681kΩ
Option: Panasonic 120Ω Bead
EXC3BP121H
Lower Amp output noise from charge pump
ISL59830
Pb
Pr
IN3 9
ISL43140
FN7489.6
May 4, 2006
ISL59830
Demo Board Schematic
RED_IN
R1
75Ω
RED_OUT
GREEN_IN
R2
75Ω
BLUE_IN
R7
1kΩ
C4
1.0µF
R3
75Ω
R4
VCC
499Ω
R8
1kΩ
VCC
REFERENCE
CONTROL
1 RIN
ROUT 16
2 GIN
GOUT 15
3 BIN
BOUT 14
4 REF
VCC 13
5 VEE
EN 12
6 GND
VCC 11
7 VEEOUT
C2
0.1µF
8 DGND
D1
1N4148
(or similar)
Description of Operation and Application
Information
Theory Of Operation
The ISL59830 is a highly practical and robust marriage of
three high bandwidth, high speed, low power, rail-to-rail
voltage feedback amplifiers with a charge pump, to provide a
negative rail without an additional power supply. Designed to
operate with a single supply voltage range of from 0V to
3.3V, the ISL59830 eliminates the need for a split supply with
the incorporation of a charge pump capable of generating a
bottom rail as much as 1.6V below ground; for a 4.9V range
on a single 3.3V supply. This performance is ideal for NTSC
video with its negative-going sync pulses.
The Amplifier
The ISL59830 fabricated on a dielectrically isolated high
speed 5V Bi-CMOS process with 4GHz PNPs and NPN
transistor exceeding 20GHz - perfect for low distortion, low
power demand and high frequency circuits. While the
ISL59830 utilizes somewhat standard voltage mode
feedback topologies, there are many non-standard analog
features providing its outstanding bandwidth, rail-to-rail
operation, and output drive capabilities. The input signal
initially passes through a folded cascode, a topology
providing enhanced frequency response essentially by fixing
the base collector voltage at the junction of the input and
gain stage. The collector of each input device looks directly
into an emitter that is tied closely to ground through a
resistor and biased with a very stable DC source. Since the
voltage of this collector is "locked stable" the effective
10
75Ω
R5
75Ω
R6
75Ω
GREEN_OUT
VCC
BLUE_OUT
C3
0.1µF
ENABLE
2
1
NC 10
DVCC 9
R4
C5
0.1µF
VCC
3
Option: Panasonic 120Ω Bead
EXC3BP121H
Lower Amp output noise from charge pump
bandwidth limiting of the Miller capacitance is greatly
reduced. The signal is then passed through a second fullyrealized differential gain stage and finally through a
proprietary common emitter output stage for improved railto-rail output performance. The result is a highly-stable, low
distortion, low power, and high frequency amplifier capable
of driving moderately capacitive loads with near rail-to-rail
performance.
Input Output Range
The three amplifier channels have an input common mode
voltage range from 0.15V below the bottom rail to within
100mV of the positive supply, VS+ pin (Note: bottom rail is
established by the charge pump at negative one half the
positive supply). As the input signal moves outside the
specified range, the output signal will exhibit increasingly
higher levels of harmonic distortion. And of course, as load
resistance becomes lower, the current drive capability of the
device will be challenged and its ability to drive close to each
rail is reduced. For instance, with a load resistance of 1kΩ
the output swing is within a 100mV of the rails, while a load
resistance of 150Ω limits the output swing to within around
300mV of the rails.
Amplifier Output Impedance
To achieve near rail-to-rail performance, the output stage of
the ISL59830 uses transistors in the common emitter
configuration, typically producing higher output impedance
than the standard emitter follower output stage. The
exceptionally high open loop gain of the ISL59830 and local
feedback reduces output impedance to less than a 2Ω at low
frequency. However, since output impedance of the device is
FN7489.6
May 4, 2006
ISL59830
IN+
INOUT
BIAS
exponentially modulated by the magnitude of the open loop
gain, output impedance increases with frequency as the
open loop gain decreases with frequency. This inductive-like
effect of the output impedance is countered in the ISL59830
with proprietary output stage topology, keeping the output
impedance low over a wide frequency range and making it
possible to easily and effectively drive relatively heavy
capacitive loads.(See Figure 11).
The Charge Pump
The ISL59830 charge pump provides a bottom rail up to
1.65V below ground while operating on a 0V to 3.3V power
supply. The charge pump is internally regulated to one-half
the potential of the positive supply. This internal multi-phase
charge pump is driven by a 160MHz differential ring
oscillator driving a series of inverters and charge storage
circuitry. Each series inverter charges and places parallel
adjoining charge circuitry slightly out of phase with the
immediately preceding block. The overall effect is sequential
discharge and generation of a very low ripple of about 10mV
that is applied to the amplifiers providing a negative rail of up
to -1.65V.
There are two options to reduce the output supply noise.
• Add a 120Ω bead in series between VCC and DVCC to
further reduce ripple.
Add a 20pF capacitor between the back load 75Ω resistor
and ground (see the ISL59830A + DC-Restore Solution
schematic on page 10).
11
VOLTS (10mV/DIV)
FIGURE 27.
TIME (20ns/DIV)
FIGURE 28. CHARGE PUMP OSCILLATION (AMP OUTPUT)
The system operates at sufficiently high frequencies that any
related charge pump noise is far beyond standard video
bandwidth requirements. Still, appropriate bypassing
discipline must be observed, and all pins related to either the
power supply or the charge pump must be properly
bypassed. See "Power Supply Bypassing and Printed Circuit
Board Layout" in this section.
To maximize resistance to latch-up, a diode should be added
between the VEEOUT pin (anode) and GND (cathode), as
shown in the Demo Board Schematic. This prevents VEE
from rising more than 0.7V above ground during startup.
(VEE > 1V above GND can cause latchup under some
conditions.)
FN7489.6
May 4, 2006
ISL59830
The VREF Pin
Applying a voltage to the VREF pin simply places that
voltage on what would usually be the ground side of the gain
resistor of the amplifier, resulting in a DC-level shift of the
output signal. Applying 100mV to the Vref pin would apply a
-100mV DC level shift to the outgoing signal. The charge
pump provides sufficient bottom room to accommodate the
shifted signal. VREF may be connected to ground for back
porch at ground.
Note: The VREF input is the common point of the 3 amps
minus input resistors. Any common resistance on VREF
input will share the voltage induced on it with all the other
amps, so using a resistor source to get offset will cause
cross talk and gain change for the offset for all amps and
amp +input gain change. Offset on the VREF pin must be low
impedance to prevent gain error and cross talk. A transistor
emitter follower should work like an NPN MMBT3904 with
the emitter connected to the VREF pin and 1k pull down to Vwith 1µF cap bypass to ground and the collector to V+ and
base to V offset source. If better tempco is needed then a
diode may be used in series with the pot to ground. A 499Ω
resistor may be added in series with the collector to prevent
damage when testing.
See the Block Diagram on page 8.
The VEE Pin
The VEE pin is the output pin for the charge pump. A
voltmeter applied to this pin will display the output of the
charge pump. This pin does not affect the functionality of the
part. One may use this pin as an additional voltage source.
Keep in mind that the output of this pin is generated by the
internal charge pump and a fully regulated supply that must
be properly bypassed. We recommend a 0.1µF ceramic
capacitor placed as close to the pin and connected to the
ground plane of the board.
Input, Output, and Supply Voltage Range
The ISL59830 is designed to operate with a single supply
voltage range of from 0V to 3.3V. The need for a split supply
has been eliminated with the incorporation of a charge pump
capable of generating a bottom rail as much as 1.6V below
ground, for a 4.9V range on a single 3.3V supply. This
performance is ideal for NTSC video with its negative-going
sync pulses.
Video Performance
For good video performance, an amplifier is required to
maintain the same output impedance and the same
frequency and phase response as DC levels are changed at
the output. This is especially difficult when driving a standard
video load of 150Ω because of the change in output current
with changing DC levels. Special circuitry has been
incorporated into the ISL59830 for the reduction of output
impedance variation with the current output. This results in
outstanding differential gain and differential phase
12
specifications of 0.06% and 0.1°, while driving 150Ω at a
gain of +2. Driving higher impedance loads would result in
similar or better differential gain and differential phase
performance.
NTSC
The ISL59830, generating a negative rail internally, is ideally
suited for NTSC video with its accompanying negative-going
sync signals; easily handled by the ISL59830 without the
need of an additional supply as the ISL59830 generates a
negative rail with an internal charge pump referenced at
negative 1/2 the positive supply.
YPbPr
YPbPr signals originating from a DVD player requiring three
channels of very tightly-controlled amplifier gain accuracy
present no difficulty for the ISL59830. Specifically, this
standard encodes sync on the Y channel and it is a negativegoing signal; easily handled by the ISL59830 without the
need of an additional supply as the ISL59830 generates a
negative rail placed at negative 1/2 the positive supply.
Additionally, the Pb and Pr are bipolar analog signals and
the video signals are negative-going; and again easily
handled by the ISL59830.
Driving Capacitive Loads and Cables
The ISL59830, internally-compensated to drive 75Ω cables,
will drive 10pF loads in parallel with 1kΩ with less than 5dB
of peaking. If less peaking is required, a small series resistor,
usually between 5Ω to 50Ω, can be placed in series with the
output. This will reduce peaking at the expense of a slight
closed loop gain reduction. When used as a cable driver,
double termination is always recommended for reflectionfree performance. For those applications, a back-termination
series resistor at the amplifier's output will isolate the
amplifier from the cable and allow extensive capacitive drive.
However, other applications may have high capacitive loads
without a back-termination resistor. Again, a small series
resistor at the output can help to reduce peaking. The
ISL59830 is a triple amplifier designed to drive three
channels; simply deal with each channel separately as
described in this section.
DC-Restore
When the ISL59830 is AC-coupled it becomes necessary to
restore the DC reference for the signal. This is accomplished
with a DC-restore system applied between the capacitive
"AC" coupling and the input of the device. Refer to
Application Circuit for reference DC-restore solution.
Amplifier Disable
The ISL59830 can be disabled and its output placed in a
high impedance state. The turn-off time is around 25ns and
the turn-on time is around 200ns. When disabled, the
amplifier's supply current is reduced to 80mA typically,
reducing power consumption. The amplifier's power-down
can be controlled by standard TTL or CMOS signal levels at
FN7489.6
May 4, 2006
ISL59830
the EN pin. The applied logic signal is relative to GND pin.
Letting the EN pin float or applying a signal that is less than
0.8V above GND will enable the amplifier. The amplifier will
be disabled when the signal at EN pin is 2V above GND. The
VEE charge pump remains active.
Output Drive Capability
The ISL59830 does not have internal short-circuit protection
circuitry. A short-circuit current of 80mA sourcing and 150mA
sinking for the output is connected to half way between the
rails with a 10Ω resistor. If the output is shorted indefinitely,
the power dissipation could easily increase such that the part
will be destroyed. Maximum reliability is maintained if the
output current never exceeds ±40mA, after which the
electro-migration limit of the process will be exceeded and
the part will be damaged. This limit is set by the design of the
internal metal interconnections.
Power Dissipation
With the high output drive capability of the ISL59830, it is
possible to exceed the 150°C absolute maximum junction
temperature under certain load current conditions.
Therefore, it is important to calculate the maximum junction
temperature for an application to determine if load conditions
or package types need to be modified to assure operation of
the amplifier in a safe operating area.
The maximum power dissipation allowed in a package is
determined according to:
T JMAX – T AMAX
PD MAX = --------------------------------------------Θ JA
Where:
TJMAX = Maximum junction temperature
TAMAX = Maximum ambient temperature
Where:
VS = Supply voltage
ISMAX = Maximum quiescent supply current
VOUT = Maximum output voltage of the application
RLOAD = Load resistance tied to ground
ILOAD = Load current
i = Number of output channels
By setting the two PDMAX equations equal to each other, we
can solve the output current and RLOAD to avoid the device
overheat.
Power Supply Bypassing and Printed Circuit
Board Layout
Strip line design techniques are recommended for the input
and output signal traces. As with any high frequency device,
a good printed circuit board layout is necessary for optimum
performance. Lead lengths should be as short as possible.
The power supply pin must be well bypassed to reduce the
risk of oscillation. For normal single supply operation, where
the VS- pin is connected to the ground plane, a single 4.7µF
tantalum capacitor in parallel with a 0.1µF ceramic capacitor
from VS+ to GND will suffice. This same capacitor
combination should be placed at each supply pin to ground if
split-internal supplies are to be used. In this case, the VSpin becomes the negative supply rail.
For good AC performance, parasitic capacitance should be
kept to a minimum. Use of wire-wound resistors should be
avoided because of their additional series inductance. Use
of sockets should also be avoided if possible. Sockets add
parasitic inductance and capacitance can result in
compromised performance. Minimizing parasitic capacitance
at the amplifier's inverting input pin is also very important.
ΘJA = Thermal resistance of the package
The maximum power dissipation actually produced by an IC
is the total quiescent supply current times the total power
supply voltage, plus the power in the IC due to the load, or:
for sourcing:
V OUT i
PD MAX = V S × I SMAX + ( V S – V OUT i ) × ----------------RL i
for sinking:
PD MAX = V S × I SMAX + ( V OUT i – V S ) × I LOAD i
13
FN7489.6
May 4, 2006
ISL59830
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M16.15A
N
INDEX
AREA
H
0.25(0.010) M
16 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
B M
E
-B1
2
INCHES
GAUGE
PLANE
3
0.25
0.010
SEATING PLANE
-A-
A
D
h x 45°
-C-
e
α
A2
A1
B
0.17(0.007) M
L
C
0.10(0.004)
C A M
B S
NOTES:
SYMBOL
MIN
MAX
MIN
MAX
NOTES
A
0.061
0.068
1.55
1.73
-
A1
0.004
0.0098
0.102
0.249
-
A2
0.055
0.061
1.40
1.55
-
B
0.008
0.012
0.20
0.31
9
C
0.0075
0.0098
0.191
0.249
-
D
0.189
0.196
4.80
4.98
3
E
0.150
0.157
3.81
3.99
4
e
0.025 BSC
0.635 BSC
-
H
0.230
0.244
5.84
6.20
-
h
0.010
0.016
0.25
0.41
5
L
0.016
0.035
0.41
0.89
6
8°
0°
N
1. Symbols are defined in the “MO Series Symbol List” in Section
2.2 of Publication Number 95.
MILLIMETERS
α
16
0°
16
7
8°
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
Rev. 2 6/04
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions.
Interlead flash and protrusions shall not exceed 0.25mm (0.010
inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual
index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable
dambar protrusion shall be 0.10mm (0.004 inch) total in excess
of “B” dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions are not necessarily exact.
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
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14
FN7489.6
May 4, 2006