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October 22, 2004
Data Sheet
Microprocessor CORE Voltage Regulator
Two-Phase Buck PWM Controller
The ISL6560 two-phase current mode, PWM control IC
together with companion gate drivers, the HIP6601A,
HIP6602A, HIP6603A or HIP6604 and MOSFETs provides a
precision voltage regulation system for advanced
microprocessors. Two-phase power conversion is a marked
departure from earlier single phase converter configurations
previously employed to satisfy the ever increasing current
demands of modern microprocessors. Multi-phase
converters, by distributing the power and load current,
results in smaller and lower cost transistors with fewer input
and output capacitors. These reductions accrue from the
higher effective conversion frequency with higher frequency
ripple current due to the phase interleaving process of this
topology. For example, a two phase converter operating at
350kHz per phase will have a ripple frequency of 700kHz.
Higher converter bandwidth is also achievable, resulting in
faster response to load transients.
An outstanding feature of this controller IC includes highside current sensing with a single current sampling resistor
in the input line to the output MOSFET transistors. This
single current sampling resistor monitors each channels
input current assuring excellent current sharing. Current
mode control results in rapid response to changing load
demands.
Also featured are programmable VID codes with an
accuracy of 0.8% that range from 1.100–1.850V, and are
set by the microprocessor. Pull up currents on these VID
pins eliminates the need for external pull-up resistors.
Another feature of this controller IC is the PWRGD monitor
circuit and load protection circuits which provide overvoltage
protection, overcurrent protection and undervoltage
indication.
Ordering Information
PART NUMBER
ISL6560CB
ISL6560CB-T
ISL6560CBZ
(See Note)
TEMP. (°C)
0 to 70
PACKAGE
16 Ld SOIC
PKG.
DWG.#
M16.15
0 to 70
16 Ld SOIC
(Pb-free)
M16.15
ISL6560CBZ-T
(See Note)
16 Ld SOIC Tape and Reel (Pb-free)
ISL6560/62EVAL1
Evaluation Platform
FN9011.3
Features
• Two-phase power conversion
• Precision channel current sharing
• Precision CORE voltage regulation
- 0.8% accuracy
• Microprocessor voltage identification input
- VRM 9.0 compliant
- 5-bit VID input
- 1.100 to 1.850V in 25mV steps
- Programmable “droop” voltage
• Fast transient recovery time
• Overcurrent protection
• High output ripple frequency. . . . . . . . . . . . . 100kHz to 2MHz
• Pb-Free Available (RoHS Compliant)
Applications
• VRM9.X modules
• AMD Athlon™ processor voltage regulator
• Low output voltage, high current DC/DC converters
Related Literature
• Technical Brief TB363 Guidelines for Handling and
Processing Moisture Sensitive Surface Mount Devices
(SMDs)
Pinout
ISL6560 (SOIC)
TOP VIEW
VID4 1
16 VCC
VID3 2
15 REF
VID2 3
14 CS-
VID1 4
13 PWM1
VID0 5
12 PWM2
COMP 6
FB 7
16 Ld SOIC Tape and Reel
ISL6560
CT 8
11 CS+
10 PWRGD
9 GND
NOTE: Intersil Pb-free products employ special Pb-free material sets; molding
compounds/die attach materials and 100% matte tin plate termination finish,
which are RoHS compliant and compatible with both SnPb and Pb-free
soldering operations. Intersil Pb-free products are MSL classified at Pb-free
peak reflow temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020C.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 321-724-7143 | Intersil and Design is a trademark of Intersil Americas LLC.
Athlon™ is a trademark of Advanced Micro Devices, Inc. | Copyright © Intersil Americas LLC. 2003, 2004, All Rights Reserved
ISL6560
Block Diagram
VCC
REF
UVLO and
BIAS CIRCUITS
3V REFERENCE
PWRGD
OSCILLATOR
+
X 0.82
UV
-
VID4
CMP
+
-
-
VID1
PWM2
OVP
E/A
+
D/A
VID2
PWM1
CONTROL
LOGIC
+
X1.24
VID3
CT
-
CS+
CS-
VID0
FB
COMP
GND
Simplified Power System Diagram
FB
PWM 1
FB (Pin 7)
SYNCHRONOUS
RECTIFIED BUCK
CHANNEL
ISL6560
MICROPROCESSOR
PWM 2
SYNCHRONOUS
RECTIFIED BUCK
CHANNEL
A capacitor on this pin sets the frequency of the internal
oscillator.
Functional Pin Description
VID4 1
16 VCC
VID3 2
15 REF
VID2 3
14 CS-
VID1 4
13 PWM1
VID0 5
12 PWM2
11 CS+
FB 7
10 PWRGD
CT 8
9 GND
VID4 (Pin 1), VID3 (Pin 2), VID2 (Pin 3), VID1 (Pin 4)
and VID0 (Pin 5)
Voltage Identification inputs from microprocessor. These pins
respond to TTL and 3.3V logic signals. The ISL6560 decodes
VID bits to establish the output voltage. See Table 1.
COMP (Pin 6)
Output of the internal transconductance error amplifier.
Voltage at this pin sets the output current level of the current
2
Inverting input of the internal transconductance error
amplifier.
CT (Pin 8)
VID
COMP 6
sense comparator. Pulling this pin to ground disables the
oscillator and drives both PWM outputs low.
GND (Pin 9)
All signals are referenced to this bias and reference ground pin.
PWRGD (Pin 10)
This pin is an internal open drain connection. A high voltage
level at this pin with a resistor connected to this pin and VCC
indicates that CORE voltage is at the proper level,
CS+ (Pin 11) and CS- (Pin 14)
These inputs monitor the supply current to the upper
MOSFETs. CS+ is connected directly to the decoupled
supply voltage and current sensing resistor. CS- is
connected to the other end of the current sensing resistor
and the upper MOSFET drains.
PWM2 (Pin 12) and PWM1 (Pin 13)
PWM outputs that are connected to the gate driver ICs.
REF (Pin 15)
Three volt supply used to bias the output of the
transconductance amplifier.
VCC (Pin 16)
Connect this bias supply pin to a 12V supply.
FN9011.3
ISL6560
Absolute Maximum Ratings
Thermal Information
Supply Voltage (VCC) . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to 15V
CS+. CS- . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to VCC + 0.3V
PWRGD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to VCC
All Other Inputs and Outputs . . . . . . . . . . . . . . . . . . . . . . -0.3V to 5V
ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .3kV
Thermal Resistance (Note 1)
Operating Conditions
JA (oC/W)
SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
75
Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC
Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC
(SOIC - Lead Tips Only)
Ambient Temperature Range . . . . . . . . . . . . . . . . . . . . . 0oC to 70oC
Maximum Operating Junction Temperature . . . . . . . . . . . . . . 125oC
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12V 10%
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
VCC = 12V
-
5.8
9.0
mA
VCC  VUVLO, VCC Rising
-
5.7
8.9
mA
5.4
6.4
6.9
V
0.1
0.4
0.8
V
VCC SUPPLY CURRENT
Input Supply Current
ICC
Input Supply Current, UVLO Mode
Undervoltage Lock Out Voltage
ICC(UVLO)
VUVLO
Undervoltage Lock Out Hysteresis
DAC and REFERENCE VOLTAGES
Minimum DAC Programed Voltage
VFB
DAC Programmed to 1.100V
1.091
1.100
1.109
V
Middle DAC Programed Voltage
VFB
DAC Programmed to 1.475V
1.463
1.475
1.487
V
Maximum DAC Programed Voltage
VFB
DAC Programmed to 1.850V
1.835
1.850
1.865
V
-
0.05
-
%
124
134
%
VFB
Line Regulation
VCC = 10V to 14V
Crowbar Trip Point at FB Input
VCROWBAR
Percent of Nominal DAC Voltage
114
Crowbar Reset Point at FB Input
VCROWBAR
Percent of Nominal DAC Voltage
50
60
70
%
Crowbar Response Time
ICROWBAR
Overvoltage to PWM Going Low
-
300
-
ns
2.952
3.000
3.048
V
1
3
-
mA
Reference Voltage
VREF
Output Current
IREF
0mA  IREF  1mA
VID INPUTS
Input Low Voltage
VIL(VID)
-
-
0.6
V
Input High Voltage
VIH(VID)
2.2
-
-
V
10
20
40
A
4.5
5.0
5.5
V
VID Pull-Up
IVID
VIDx = 0V or VIDx = 3V
Internal Pull-Up Voltage
OSCILLATOR
Maximum Frequency
fCT(MAX)
Frequency Variation
fCT
CT Charging Current
ICT
CT Charging Current
ICT
2.0
-
-
MHz
TA = 25oC, CT = 91pF
430
500
570
kHz
TA = 25oC, VFB in Regulation
TA = 25oC, VFB = 0V
130
150
170
A
26
36
46
A
-
200
-
k
2.0
2.2
2.4
mS
-
1
-
mA
ERROR AMPLIFIER
Output Resistance
RO(ERR)
Transconductance
gm(ERR)
Output Current
Io(ERR)
3
FB Forced to VOUT - 3%
FN9011.3
ISL6560
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted
PARAMETER
SYMBOL
Input Bias Current
TEST CONDITIONS
IFB
Maximum Output Voltage
VCOMP(MAX) FB Forced to VOUT - 3%
Output Disable Threshold
VCOMP(OFF)
FB Low Foldback Threshold
VFB(LOW)
-3dB Bandwidth
BWERR
COMP = Open
VCS(TH)
CS+ = VCC, FB Forced to VOUT - 3%
MIN
TYP
MAX
UNITS
-
5
100
nA
-
3.0
-
V
560
720
800
mV
375
425
500
mV
-
500
-
kHz
142
157
172
mV
-
0
15
mV
CURRENT SENSE
Threshold Voltage
0.8  COMP  1V
Current Limit Foldback Voltage
Input Bias Current
Response Time
FB  375mV
75
95
115
mV
ni
1 V VCOMPV
-
12.5
-
V/V
ICS+, ICS-
CS+ = CS- = VCC
-
0.5
5.0
A
CS+ - (CS-) 172mV to PWM Going Low
-
50
-
ns
VCS(FOLD)
VCOMP/VCS
tCS
POWER GOOD COMPARATOR
Undervoltage Threshold
VPWRGD(UV) Percent of Nominal Output
76
82
88
%
Overvoltage Threshold
VPWRGD(OV) Percent of Nominal Output
114
124
134
%
Output Voltage Low
VOL(PWRGD) IPWRGD(SINK) = 100A
-
30
200
mV
Response Time
FB Going High
-
2
-
s
Response Time
FB Going Low
-
200
-
ns
PWM OUTPUTS
Output Voltage Low
VOL(PWM)
IPWM(SINK) = 400A
Output Voltage High
VOH(PWM)
IPWM(SOURCE) = 400A
Output Current
IPWM
Duty Cycle Limit, by Design
DMAX
Per Phase, Relative to fCT
TABLE 1. VOLTAGE IDENTIFICATION CODES
.
VID4
VID3
VID2
VID1
VID0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
-
100
500
mV
4.5
5.0
5.5
V
0.4
1
-
mA
-
-
50
%
TABLE 1. VOLTAGE IDENTIFICATION CODES (Continued)
VDAC
VID4
VID3
VID2
VID1
VID0
VDAC
1
Off
0
1
1
0
1
1.525
0
1.100
0
1
1
0
0
1.550
0
1
1.125
0
1
0
1
1
1.575
0
0
1.150
0
1
0
1
0
1.600
0
1
1
1.175
0
1
0
0
1
1.625
0
1
0
1.200
0
1
0
0
0
1.650
1
0
0
1
1.225
0
0
1
1
1
1.675
1
0
0
0
1.250
0
0
1
1
0
1.700
1
0
1
1
1
1.275
0
0
1
0
1
1.725
1
0
1
1
0
1.300
0
0
1
0
0
1.750
1
0
1
0
1
1.325
0
0
0
1
1
1.775
1
0
1
0
0
1.350
0
0
0
1
0
1.800
1
0
0
1
1
1.375
0
0
0
0
1
1.825
1
0
0
1
0
1.400
0
0
0
0
0
1.850
1
0
0
0
1
1.425
1
0
0
0
0
1.450
0
1
1
1
1
1.475
0
1
1
1
0
1.500
4
FN9011.3
ISL6560
General Circuit Description
RSENSE
INPUT
VOLTAGE
CS+
GATE DRIVER
and
OUTPUT FETs
CS-
ISL6560
+
PWM1
+Current Comparator
VCORE
OUTPUT
GATE
DRIVERS
and
Logic
GATE DRIVER
and
OUTPUT FETs
PWM2
RESET
SET
OSCILLATOR
FB
gm Error
Amplifier
COMP
VID4
+
RL
VID3
D/A
REF
CT
VID2
Reference
VID1
Voltage
VID0
GND
FIGURE 1. FUNCTIONAL SYSTEM BLOCK DIAGRAM SHOWING MAJOR COMPONENTS
Circuit Operation
Figure 1 will be used to describes operation of the controller.
A transconductance error amplifier provides the major
voltage control function. The error amplifier’s positive input is
connected to an internal DAC that is programmed via a 5-bit
code from the microprocessor. Regulation is accomplished
5
by the amplifier attempting to make both inputs equal. This
does not happen because of the limited loop gain and
provides the bases for droop compensation mentioned
earlier and described below.
3.0
2.5
VCOMP (Volts)
The ISL6560 is a two-power channel, current mode PWM
controller with input current sensing. A transconductance
error amplifier helps establish the desired droop voltage for
microprocessor power supplies and will be explained later.
Figure 1 is a functional system block diagram of the IC in a
power supply application. A single current sampling resistor,
RSENSE, on the input side of the supply monitors the current
for both channels via a comparator within the ISL6560. A
single comparator insures that both channels are monitored
by the same circuitry, helping to balance the operating
current of each channel. During normal operation the
comparator is tripped by the peak inductor current,
terminating the conduction cycle. As more current is needed
to supply the output load, the comparator threshold voltage
is increased, increasing the inductor current to
accommodate the increased load demands.
2.0
/V
5V
12.
1.5
1.0
0.5
{
Output
0
Disable
0
Threshold
20
40
60
80 100 120
VCS(CL) (mV)
140
160
FIGURE 2. CURRENT COMPARATOR THRESHOLD
VOLTAGE AS A FUNCTION OF VCOMP
Figure 2 shows a curve of the current comparator threshold
voltage as a function of the error amplifier output voltage,
VCOMP. From this curve, it can be seen that as VCOMP
FN9011.3
ISL6560
Droop Voltage
Gain of the error amplifier is gm x RL. If RL is 10k and the gm
is 2.2mS, the gain will be 2.2mS x 10k = 22. For example,
assume to satisfy the no-load to full load requirements the
VCOMP voltage must increase by 1.5V. The error amplifier
input voltage or droop voltage will be 1.5V / 22 = 68mV
below the DAC voltage. As will be shown later this is not the
sequence one uses when designing a supply, but is useful at
this point, to explain the operation.
Initial Voltage
The initial starting, or no-load voltage is set by the
programmed DAC voltage and the reference voltage set at
the output of the gm amplifier. The reference voltage is
connected to the upper end of the amplifier load resistor, RL,
shown in Figure 1. Assume that the voltage to the current
comparator is set to 1.2V to satisfy the inductor no-load
ripple current. This means that the gm amplifier does not
have to supply any output current. Under this condition, the
error amplifier input is the DAC voltage.
Now assume that the reference voltage to the gm amplifier is
set to 1.4V, instead of 1.2V. The gm amplifier must reduce its
output to 1.2V to set the comparator no-load threshold
voltage to the correct voltage to supply the inductor ripple
current. The error amplifier output must pull down or reduce
the voltage to the comparator by the added 200mV. This will
cause the gm amplifier input to go more positive to drive the
error amplifier output low. The initial no-load voltage, with a
gain of 22, will be 200mV/22 or 9mV high. If the reference
voltage is set low by the same amount the no-load starting
voltage will be low by that same amount.
A 3V reference is provided within the ISL6560. A voltage
divider is established by two external series resistors
connected between the reference voltage, REF and ground.
The center of the two resistors is connected to COMP and
sets the initial voltage. The parallel combination sets the
equivalent error amplifier load, RL. Determination of the
resistor values will be discussed later.
Oscillator Frequency
An external capacitor establishes the basic timing for the
sawtooth oscillator. An internal current source of 150A
ramps the timing capacitor from ground to approximately 3V
with low values of timing capacitors, (< 150pF). This
establishes the basic period for the oscillator. Approximately
150ns is fixed for the retrace. With increasing values of
timing capacitors the sawtooth amplitude is reduced
because the timing capacitor does not retrace to ground.
6
Figure 3 is a plot of the oscillator frequency versus timing
capacitor value.
During supply start-up, or when the error amplifier input is at
zero volts, the oscillator’s charging current is reduced from
its operating value of 150A to 36A, reducing its frequency.
Monitoring and Protection Systems
Power Good
Internal monitoring circuits verify, via a high open drain
PWRGD output signal, that the supply voltage is within
+124% to -82% of the programed DAC voltage. An external
pull-up resistor must be connected from this pin to a positive
supply. Load currents should be kept below 100A. Voltages
exceeding the above limits will drive the open drain PWRGD
pin low.
If the output voltage exceeds the 124% limit, the PWM
outputs will go low, turning OFF the upper gates and turning
ON the lower gates to protect the processor. When the
output voltage drops below the limit, normal operation is
restored.
Short Circuit Protection
When a short is placed on the supply, the input supply
current exceeds the current comparator maximum level. No
more current is available and the output voltage will fall.
When the regulator output voltage falls below approximately
375mv, the threshold voltage of the current comparator is
limited to 95mV. In addition, the oscillator frequency is
reduced. This effectively folds back the available output
current to limit load and regulator dissipation.
Supply Disable
The bracketed section on the left hand vertical axis of the
curve in of Figure 2 shows a range of voltages that will
initiate the disable function within the ISL6560. The PWM
outputs are driven low, opening the upper MOSFETs and
driving the lower MOSFETs ON. The oscillator is disabled
during this time. Connecting an open drain or open collector
device or a switch to pull VCOMP to ground will initiate the
disable function.
3000
FREQUENCY (kHz)
moves from approximately 1V to 3V, the current comparator
threshold voltage ranges from 0mV to 157mV. Also observe
as the output load demand increases, driving the inverting
input of the error amplifier lower, the output voltage of the
error amplifier increases. This voltage increase in VCOMP
increases the current comparator threshold voltage to satisfy
load demand.
1000
100
0
50 100 150 200 250 300 350 400
CAPACITOR CT (pF)
450 500
FIGURE 3. OSCILLATOR FREQUENCY vs. TIMING CAPACITOR
FN9011.3
ISL6560
the “K” term is fixed by the input output design criteria.
This section will highlight a 40A converter, providing the
design details for the entire supply. The hardware realization
of this design is the ISL6560/62 Evaluation Board. For this
example a 40A supply down converting from 12V will be
discussed. 5V operation is also viable as an input source.
Oscillator frequency is 350kHz, with a channel frequency of
175kHz. The ISL6560 has an internal DAC with VRM 9.0
VID codes. An output voltage of 1.8V, near the maximum
output voltage will be used to determine the selection of
inductors. Output voltage droop from no-load to full load
specification is ~65mv. This sets the effective DC output
resistance, (ROUT) to be 65mV/40A = 1.63m
3.5
L=
(VIN - VOUT) VOUT
fsw x IL
VIN
L=
K
fsw x IL
3.0
2.5
K (VOLTS)
Design Example
K=
(VIN - VOUT) VOUT
VIN
VIN = 12V
VIN = 10V
2.0
VIN = 8V
1.5
VIN = 5V
1.0
VIN = 3.3V
0.5
Inductor Selection
To assist in the selection of the output inductors, two curves
are provided. Figure 4 deals with the selection of the voltage
terms in the equation:
L=
(VIN - VOUT) VOUT
fsw x IL VIN
0
0
4
2
6
VOUT (VOLTS)
10
8
12
FIGURE 4. “K” AS A FUNCTION OF VOUT FOR
FAMILIES OF VIN
2.5
VIN = 12V
2.0
VIN = 8V
K (VOLTS)
Each channel handles half of the 40A output. An inductor
ripple current of 40% of the output current or 8A p-p/channel
was selected. There is always a compromise between ripple
current and regulator performance. Higher values of ripple
current, as expected, result in slightly greater dissipation in
series pass transistors and losses in other resistive elements
in the power path. These disadvantages are offset by
improved transient response, with lower values of output
capacitors and less output voltage overshoot when the
output current is step reduced from heavy load conditions.
This overshoot is primarily contributed by the energy stored
in the output filter network and is not highly influenced by the
control loop.
VIN = 10V
1.5
VIN = 5V
1.0
VIN = 3.3V
0.5
Where: L = inductor value
VIN = input voltage
VOUT = output voltage or CORE Voltage
fsw = oscillator frequency/2 (for each channel)
IL= inductor ripple current
0
0
0.5
1.0
1.5
2.0
VOUT (VOLTS)
2.5
3.0
3.5
FIGURE 5. EXPANSION OF FIGURE 4 FOR VOUT < 3.5V
The (VIN - VOUT) term is the voltage across the inductor and
the VOUT/VIN term is the converter duty cycle.
The curve of Figure 4 reduces the voltage terms to a single
voltage term, “K”. To further enhance readability of the curves,
the lower portion of Figure 4 was expanded in Figure 5 for output
voltages up to only 3.5V. The dotted lines show the selection of
an output voltage of 1.8V. With 12V input, K = 1.55V.
The curve of Figure 6 shows with the selection of the inductor
value. Initially a ripple current of 40% of the full load current was
established. Each channel contributes 20A, for a ripple current,
IL, of 8A. From this, the value entered into the left-hand axis of
Figure 6 is 1.55V/8A = 0.19. With a channel operating
frequency of 200kHz, the inductor value will be 900nH, as
shown by the dotted lines. This curve shows how you can
modify the inductor value by changing the ripple current since
7
FN9011.3
ISL6560
H
2M
1
z
(Amps)
Hz
1M
0
50
z
kH
20
K
IL
(VOLTS)
high peak currents can cause heating of these capacitors
and can result in premature failure if not properly designed.
The value of the RMS current that these capacitors must
share can be approximated with the aid of the curve of
Figure 7. The dotted lines show determination of the current
multiplier.
Frequency = Channel fsw
Hz
0k
0
10
For the 40A design with the 1.8V/12V = 0.15 duty cycle, the
RMS current is 0.24 x 40A = 9.6A. From this curve, it is
evident that the maximum current is only 10A. If the duty
cycle was 50%, each channel would be ON for its full cycle
and the ripple would go to zero.
z
kH
0.1
0.01
100nH
1H
Inductance
10H
FIGURE 6. INDUCTOR SELECTION CURVES
These curves help to visualize that in some cases, major
changes in some parameters only result in subtle changes in
other parameters. For example, going from 175kHz to the
200kHz channel frequency.
CURRENT MULTIPLIER
3
0.3
0.2
0.1
0
0
0.1
0.2
0.3
0.4
0.5
DUTY CYCLE (VO/VIN)
FIGURE 7. CURRENT MULTIPLIER vs. DUTY CYCLE
Output Capacitors
The combined series resistance and inductance of the
output capacitors is one of the limiting factors in the supply’s
response to transient loads. Most DC/DC converters do not
have the bandwidth or operating frequency to respond to
rapid load changes. Therefore, attention must be paid to the
filter network, for it must be the major source of energy
during step load changes. The output capacitors must
respond by supplying the initial load current, until the
regulator loop responds and the inductor current slews.
Bulk capacitors store energy, but are limited by the effective
series resistance and inductance path to their reservoir of
energy. Considering only the series resistance, the total
effective series resistance of the parallel connected
capacitors should be equal to or less than the effective DC
ROUT of the supply. As mentioned earlier ROUT is
approximately 1.63mFor this example, six 1500F, 4V
Sanyo OS-CON capacitors provide a maximum ESR of
1.66mroughly meeting the design target.
To a first order, output ripple voltage is the product of the
capacitor’s ESR and the ripple current. In this design it is
1.66m x 8A = 13.3mV.
Sixteen 22F ceramic capacitors help provide highfrequency bypassing by providing lower inductance and low
high-frequency impedance. It is essential that additional
ceramic capacitors also be place at the load to help stabilize
the load voltage and minimize additional droop at the load.
Rubycon ZA series capacitors were selected for the input
capacitors. Their 470F, 16V capacitors have a maximum
RMS current rating of 1.6A at 105oC ambient. For 9.6A, six
capacitors are required.
On any switching supply, high frequency decoupling may be
necessary on the supply input to keep the high peak current,
fast rise current pulses contained within the supply. Often a
small inductor is placed in series with the input line to help
reduce this potential source of EMI. Ceramic capacitors to
ground also help lower the high frequency impedance to
shunt the high frequency components to reduce and contain
the high speed current pulses.
RSENSE Selection
Each channel supplies a current of 20A. Add the 4A ripple
(half of the ripple current) component and the minimum
voltage across the current sense resistor that will trip the
comparator is the minimum limit of 142mV. The RSENSE
resistor value is then 142mV / 28A = 5.07m A 5m
resistor was used for this function to insure the minimum
current.
The maximum current is also important. The maximum
threshold voltage for the current comparator is 172mV. The
maximum current would be: 172mV / 5m = 34.4A per
channel. The 4A of ripple current per channel must be
subtracted to yield 30.4A per channel. The maximum output
current would be two times the channel current, or 60.8A.
Input Capacitors
The input capacitors are also critical to supply operation.
They must provide enough energy to prevent the input
voltage from dropping due to load transients. In addition, the
8
FN9011.3
ISL6560
To calculate the dissipation in the 5m resistor, we used
only half of the ripple current, 4A, to give a nominal
dissipation of:
1.8V
I RMS = Ip D = Ip -----------12V
2
2
·
Power = Ip  D  R SENSE = 3 0.4  0.15  5m
Power = 0.69W per channel or 1.38W for both channels
Where IP is the peak current and D is the duty cycle. Two
10m1W resistors in parallel were selected.
Once the value for RL is set, only the values of the resistors
that make up the voltage divider must be determined. Figure
8 shows the equations to determine the resistor network that
makes up RL.
VREF = 3V
RU
ni  R SENSE
12.5  5m
R L = ----------------------------------------- = -------------------------------------------------------- = 8.7k
gm  R OUT  2
2.2mS  1.63m  2
gm Amplifier Gain== gm  RL = 2.2mS  8.7k = 19.1
gmAmplifierGain
The ni term is the ratio of the VCOMP to the current
comparator threshold voltage; see Figure 2. RL is made up
of two resistors that form a voltage divider from the internal
3V reference supply.
As described earlier in the Circuit Description section, the
output voltage of the gm amplifier establishes the threshold
voltage of the current comparator. At approximately 1V, the
current comparator threshold voltage is near zero. With no
current demands, the regulator output voltage would be the
same as the programmed DAC voltage. However, an 8A
ripple current was selected for this design. This results in the
output of the gm amplifier moving upwards to supply the
ripple current. The voltage at the COMP pin, VSET, will be:
I RIPPLE  R SENSE  ni
V SET = 1V + ---------------------------------------------------------------2
8A  5m  12.5
= 1V + --------------------------------------------- = 1V + 250mV = 1.25V
2
The voltage divider establishes the reference voltage for
VCOMP that was set to 1.2V for this design, so the error
amplifier must drive the COMP pin 50mV more positive to
bring it to 1.25V from the 1.2V originally set. This additional
50mV output will result in an input voltage to the error
amplifier of: 50mv / 19.1 = 2.62mV below the programmed
DAC voltage of 1.8V. Neglected, is a negative term
associated with the 60ns delay of the current comparator.
This delay will cause the current ramp to be slightly greater
than predicted by the equation. This means that the initial
setting should be slightly reduced to account for the increase
in current.
9
V REF
R U = ---------------  R L
V SET
To COMP pin,
this voltage is VSET
RB
RL Selection
As discussed in the section under Droop Voltage and shown
in Figure 1, resistor RL establishes the gain of the
transconductance error amplifier. It is this resistor that sets
the droop voltage or regulation. Like any feedback system,
the higher the gain the better the regulation. The value of
this resistor may be determined from the following equation:
R U  R B = R L
I RIPPLE  R SENSE  ni
V SET = 1V + ---------------------------------------------------------------2
3V
R U = ----------------  8.7k = 20.9k
1.25V
V SET
R B = ------------------------------------  R U
V REF – V SET
1.25V
R B = -----------------------------  20.9k = 14.9k
3V – 1.25V
FIGURE 8. EQUATIONS TO DETERMINE RL DIVIDER
CC and RC Selection
Optimum transient response depends upon the selection of
the compensation capacitor network placed across the
output of the transconductance error amplifier.
To a first order, the selection of the capacitor, CC, placed
across the error amplifier may be determined by making the
product of the regulator output resistance and output
capacitors equal to the product of the RL and CC. This yields
the equation for the compensation capacitor:
R OUT  C OUT
1.63m  9mF
C C = --------------------------------------- = ----------------------------------------- = 1.68nF
RL
8.7k
A 1nF capacitor was selected from transient testing. To
prevent excessive phase shift due to the compensation
capacitor, it is usually necessary to place a resistor inseries
with the capacitor to prevent excessive phase shift beyond
the frequency of interest. This is pole cancellation and the
resistor is approximately 0.5 x RL. Figure 9 shows this
network and the equivalent circuit is approximately 0.5 x RL.
Many variables have been used in the selection of the
various gain and filter networks to this point. A broad range
of component tolerances range from 1% to 20% have
been used in the design. Therefore, it is important to
evaluate the entire system with dynamic pulse load testing.
This will verify optimum transient response and also indicate
poor response in terms of excessive overshoot, ringing or
oscillation if the compensation network is not optimum.
VREF = 3V
AC Equivalent
RU
CC
To COMP pin
CC
RC
RB
RC
RL
RC = 0.5 x RL
FIGURE 9. COMPENSATION CIRCUIT
FN9011.3
ISL6560
Ensure that both MOSFETs are within their maximum
junction temperature at high ambient temperature by
calculating the temperature rise according to package
thermal-resistance specifications. A separate heat sink may
be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
MOSFET Selection and Considerations
In high-current PWM applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes two
loss components; conduction loss and switching loss. These
losses are distributed between the upper and lower
MOSFETs according to duty factor (see the following
equations). The conduction losses are the main component
of power dissipation for the lower MOSFETs, Q2 and Q4 of
Figure 10. Only the upper MOSFETs, Q1 and Q3 have
significant switching losses, since the lower device turns on
and off into near-zero voltage.
2
I O  r DS  ON   V OUT I O  V IN  t SW  F S
P UPPER = ------------------------------------------------------------ + ---------------------------------------------------V IN
2
2
I O  r DS  ON    V IN – V OUT 
P LOWER = --------------------------------------------------------------------------------V IN
The equations assume linear voltage-current transitions and
do not model power loss due to the reverse-recovery of the
lower MOSFET’s body diode. The gate-charge losses are
dissipated by the driver IC and don't heat the MOSFETs.
However, large gate-charge increases the switching time,
tSW which increases the upper MOSFET switching losses.
Figure 10 shows a schematic of the circuit developed from
the proceeding computations. This circuit is implemented on
the ISL6560/62 Evaluation Board. The next section will
discuss PC board layout.
L3
12V
5m
1H
+VIN
6 - 470F
C15, C17-18, C29, C50-51
2 - 4.7F
C41-42
10
R6
4.7F
22k
R11
1nF C12
15k
R12
1 VID4
VCC 16
2 VID3
REF 15
3 VID2
CS- 14
4 VID1
PWM1 13
5 VID0
PWM2 12
6 COMP
20
R14
C20
R7
330
330pF
C11
CS+ 11
7 FB
PWRGD 10
8 CT
GND 9
Q1 HUF76139
15nF
C40
INPUT
VID CODES
from
PROCSSOR {
{
R13
0.1F
1 UGATE PHASE 8
2 BOOT
C1
PVCC 7
3 PWM
VCC 6
4 GND
LGATE 5
HIP6601ECB
1nF
C10
900nH
Q2 HUF76145
1F
C2
C21, C24-28
6 - 1200F
1k
R5
100pF
C13
+VCORE
Q3
HUF76139
150pF ISL6560
C14
4.3k
R27
L1
0.1F
1 UGATE PHASE 8
2 BOOT
C3
PVCC 7
3 PWM
VCC 6
4 GND
LGATE 5
HIP6601ECB
1F
L2
900nH
Q4
HUF76145
16 - 22F
C19, C30,
C34-37,
C39, C45-49,
C60-63
C4
FIGURE 10. SCHEMATIC DIAGRAM OF A 40A SUPPLY USING THE ISL6560 CONTROLLER AND HIP6601 GATE DRIVERS
PC Board Layout Considerations
Like all high-current supplies where low-voltage control
signals in the millivolt range must live with high voltage and
high-current switching signals, PC board layout becomes
crucial in obtaining a satisfactory supply.
Figure 11 shows a simplified diagram of the critical areas of
a PC board layout. This diagram and the following material
represent goals to work towards during the layout phase.
Goals will be compromised during the layout process due to
10
component placement and space constraints. The following
text reviews these layout considerations in more detail.
Current Sampling
1. Place the current sampling or sense resistor as close as
possible to the upper MOSFET drains. This is important
since the added inductance and resistance increase the
impedance and result in a reduction in drain voltage
during high peak pulse currents.
FN9011.3
ISL6560
2. Current sense is critical, especially at lower current levels
where the current comparator threshold voltage is lower.
A good Kelvin connection requires that the voltage
sample must be taken at the RSENSE resistor ends, and
not at the planes to which the resistor is connected.
3. The lines to the current sense resistor should be parallel
and run away from the PHASE or PWM signals to prevent
coupling of spikes to the current comparator input that
may delay or advance triggering of the comparator.
Parallel routing will work towards equal exposure for both
lines, so that the comparator common mode rejection
characteristic will reduce the influence of coupled noise.
4. Place the current sense filter network near the controller.
This will help reduce extraneous inputs to the
comparator.
5. Make sure the DC plus pulse voltage inputs to the current
sense comparator, CS+ and CS-, do not exceed the
voltage on the VCC pin by more than the specified limit of
VCC + 0.3V.
Voltage Sampling
1. To obtain optimum regulation use the Kelvin connection
for the input voltage sample as shown in Figure 11. The
ground connection, Pin 9 of the ISL6560 should be
connected to the system ground at the load.
2. The two voltage sampling lines described in item 1 above
should also be routed away from any high-current or
high- pulse voltages such as the phase lines or pads.
Doing this will reduce the possibility of coupling undesired
pulses into the feedback signal and either modifying the
output of the error amplifier or, if of sufficient amplitude,
spuriously triggering the current comparator by
readjusting the threshold voltage.
Other Considerations
1. Keep the leads to the timing capacitor connected to pin
CT short and return the ground directly to Pin 9.
2. When using a transistor to disable the converter by
pulling the CT pin to ground, place the transistor close to
the CT pin to minimize extraneous signal pickup.
3. As in all designs, keep decoupling networks near the pins
that must be decoupled. For example, the
decoupling/filter network on the FB input. The series
resistor should be located next to the FB pin.
4. Large power and ground planes are critical to keeping
performance and efficiency high. Consider a 1m
resistance in a 40A supply line. With 1.8V output, this
results in slightly over 2% power loss in the 72W supply.
12V
+VIN
Input
VID Codes
from
Processor {
{
Keep Leads Together
& Away from Output
1 VID4
VCC 16
2 VID3
REF 15
1 UGATE PHASE 8
3 VID2
CS- 14
2 BOOT
4 VID1
PWM1 13
5 VID0
PWM2 12
6 COMP
7 FB
CS+ 11
PWRGD 10
8 CT
Locate
Parts
Next
to IC
Place Near Drains of the
Output Transistors
PVCC 7
3 PWM
VCC 6
4 GND
LGATE 5
+VCORE
HIP6601ECB
GND 9
ISL6560
Locate
Parts
Next to IC
Try to return bypass
capacitors to ground
of lower MOSFETs
FIGURE 11. SCHEMATIC DIAGRAM SHOWING ONLY ONE CHANNEL OF ‘IDEAL’ COMPONENT PLACEMENT
11
FN9011.3
ISL6560
F. Output Capacitors:
ISL6560 Supply Design Sequence
Please note several changes from the computations in the
body of the data sheet. An operating frequency of 400kHz
was chosen. A 15mV offset voltage was added to the noload output voltage to show the design procedure.
Capacitor ESR  R OUT = 1.63m
Sanyo 1500F, 4V OS-CON Capacitors
have an ESR < 10m
Six capacitors < 1.66m
Total Capacitance = 9mF 
ISL6560 Supply Design Sequence
A. Specifications:
Output Current:
Input Voltage:
G. Input Capacitor’s RMS Current:
40A
12V
Use the curve of Figure B.
CURRENT MULTIPLIER
Output Voltage: VDAC + 15mV
Output Voltage for Calculations:
VDAC = 1.8V + 15mV
Droop Voltage: 65mV
Oscillator Frequency: 400kHz (fSW)
B. Calculate ROUT:
V DROOP
65mV
R OUT = ----------------------- = ---------------- = 1.63m
I OUT
40A
0.3
0.2
0.1
0
0.1
0
0.2
0.3
DUTY CYCLE (VO / VIN)
0.4
0.5
FIGURE B. CURRENT MULTIPLIER vs. DUTY CYCLE
G. Input Capacitor’s RMS Current: (continued)
C. Determine Frequency Setting Capacitor CT:
For 40A with a duty cycle (D) of:
V OUT
1.8V
D = --------------= ------------ = 0.15
V IN
12V
The multiplier from Figure B is 0.24.
From curve of Figure A, for 400kHz use 120pF.
FREQUENCY (kHz)
3000
I RMS = 0.24  40A = 9.6A
1000
Panasonic 470F, 16V Rubycon ZA series capacitors
have a RMS current rating of 1.6A.
Six capacitors were selected.
100
0
50 100 150 200 250 300 350 400
CAPACITOR CT (pF)
450 500
FIGURE A. OSCILLATOR FREQUENCY vs. TIMING
CAPACITOR
H. Current Sense Resistor (RSENSE):
V CS  TH MIN
142mV
R SENSE = ----------------------------------------- = ------------------------- = 5.07m
I OUT I RIPPLE
20A + 8A
------------- + ---------------------2
2
Use a 5m resistor
D. Select Inductor Ripple Current (IL):
I. RSENSE Dissipation:
Choose 40% of IOUT:
I RMS = I PEAK D
I L = 40A  0.4 = 16A
Power = I
Or 8A / Channel
2
P
 D  R SENSE
Where: IP = 34.4A -4A = 30.4A
(Using half the ripple current)
2
E. Determine the Inductors:
V IN – V OUT V OUT
12V – 1.8V
1.8V
L = ----------------------------------  ----------------- = -----------------------------------  -----------f SW
V IN
200kHz  8A 12V
------------  I L
2
Power = 30.4 A  0.15  5m
Power = 0.69W per channel or 1.38W for both channels
Used two 10m, 1W resistors in parallel.
J. RL Selection:
= 956nH
R
L
ni  R SENSE
12.5  5m
= ---------------------------------------= -------------------------------------------------------- = 8.7k
gm  R OUT  2
2.2mS  1.63m  2
gm Amplifier Gain = gm  RL = 2.2mS  8.7k = 19.1
12
FN9011.3
ISL6560
M16.15 (JEDEC MS-012-AC ISSUE C)
16 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE
Small Outline Plastic Packages (SOIC)
N
INCHES
INDEX
AREA
H
0.25(0.010) M
B M
SYMBOL
E
-B-
1
2
3
L
SEATING PLANE
-A-
h x 45o
A
D
-C-
e
B
0.25(0.010) M
C
0.10(0.004)
C A M
B S
MILLIMETERS
MAX
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
B
0.014
0.019
0.35
0.49
9
C
0.007
0.010
0.19
0.25
-
D
0.386
0.394
9.80
10.00
3
E
0.150
0.157
3.80
4.00
4
e
µ
A1
MIN
0.050 BSC
1.27 BSC
-
H
0.228
0.244
5.80
6.20
-
h
0.010
0.020
0.25
0.50
5
L
0.016
0.050
0.40
1.27
6
8o
0o
N

16
0o
16
7
8o
Rev. 1 02/02
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section
2.2 of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010
inch) per side.
5. The chamfer on the body is optional. If it is not present, a visual
index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch)
10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact.
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13
FN9011.3