HIP6014 ® Data Sheet March 2000 FN4420.2 Buck and Synchronous-Rectifier (PWM) Controller and Output Voltage Monitor Features The HIP6014 provides complete control and protection for a DC-DC converter optimized for high-performance microprocessor applications. It is designed to drive two N-Channel MOSFETs in a synchronous-rectified buck topology. The HIP6014 integrates all of the control, output adjustment, monitoring and protection functions into a single package. • Operates from +5V or +12V Input The output voltage of the converter is easily adjusted and precisely regulated. The HIP6014 includes a fully TTLcompatible 5-input digital-to-analog converter (DAC) that adjusts the output voltage from 2.1VDC to 3.5VDC in 0.1V increments and from 1.8VDC to 2.05VDC in 0.05V steps. The precision reference and voltage-mode regulator hold the selected output voltage to within ±1% over temperature and line voltage variations. The HIP6014 provides simple, single feedback loop, voltagemode control with fast transient response. It includes a 200kHz free-running triangle-wave oscillator that is adjustable from below 50kHz to over 1MHz. The error amplifier features a 15MHz gain-bandwidth product and 6V/ms slew rate which enables high converter bandwidth for fast transient performance. The resulting PWM duty ratio ranges from 0% to 100%. The HIP6014 monitors the output voltage with a window comparator that tracks the DAC output and issues a Power Good signal when the output is within ±10%. The HIP6014 protects against over-current and over-voltage conditions by inhibiting PWM operation. Additional built-in over-voltage protection triggers an external SCR to crowbar the input supply. The HIP6014 monitors the current by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor. • Drives Two N-Channel MOSFETs • Simple Single-Loop Control Design - Voltage-Mode PWM Control • Fast Transient Response - High-Bandwidth Error Amplifier - Full 0% to 100% Duty Ratio • Excellent Output Voltage Regulation - ±1% Over Line Voltage and Temperature • TTL-Compatible 5-Bit Digital-to-Analog Output Voltage Selection - Wide Range . . . . . . . . . . . . . . . . . . . 1.8VDC to 3.5VDC - 0.1V Binary Steps. . . . . . . . . . . . . . . 2.1VDC to 3.5VDC - 0.05V Binary Steps. . . . . . . . . . . . . 1.8VDC to 2.05VDC • Power-Good Output Voltage Monitor • Over-Voltage and Over-Current Fault Monitors - Does Not Require Extra Current Sensing Element, Uses MOSFET’s rDS(ON) • Small Converter Size - Constant Frequency Operation - 200kHz Free-Running Oscillator Programmable from 50kHz to over 1MHz Applications • Power Supply for Pentium®, Pentium Pro, Pentium II, PowerPC™, K6™, 6X86™ and Alpha™ Microprocessors • High-Power 5V to 3.xV DC-DC Regulators • Low-Voltage Distributed Power Supplies Ordering Information Pinout PART NUMBER HIP6014 (SOIC) TOP VIEW HIP6014CB VSEN 1 OCSET 2 19 OVP SS 3 18 VCC VID0 4 17 LGATE VID1 5 16 PGND VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD TEMP. RANGE (oC) 0 to 70 PACKAGE 20 Ld SOIC PKG. NO. M20.3 20 RT FB 10 11 GND 1 6X86TM is a trademark of Cyrix Corporation. AlphaTM is a trademark of Digital Equipment Corporation. K6TM is a trademark of Advanced Micro Devices, Inc. Pentium® is a registered trademark of Intel Corporation. PowerPCTM is a trademark of IBM. CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2002. All Rights Reserved PRISM® is a registered trademark of Intersil Americas Inc. PRISM and design is a trademark of Intersil Americas Inc. HIP6014 Typical Application 12V VIN = +5V OR +12V VCC PGOOD OCSET MONITOR AND PROTECTION SS EN OVP BOOT RT VID0 VID1 VID2 VID3 VID4 OSC UGATE PHASE HIP6014 +VOUT D/A FB LGATE - + + - COMP PGND GND VSEN Block Diagram VCC VSEN POWER-ON RESET (POR) 110% + - 90% PGOOD + - 115% + OVERVOLTAGE 10μA OVP - SOFTSTART + - OCSET REFERENCE 200μA OVERCURRENT SS BOOT UGATE 4V PHASE VID0 VID1 VID2 VID3 VID4 TTL D/A CONVERTER (DAC) DACOUT PWM COMPARATOR + - + - ERROR AMP FB GATE INHIBIT CONTROL LOGIC PWM LGATE PGND COMP GND OSCILLATOR RT 2 HIP6014 Absolute Maximum Ratings Thermal Information Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V Boot Voltage, VBOOT - VPHASE . . . . . . . . . . . . . . . . . . . . . . . .+15V Input, Output or I/O Voltage . . . . . . . . . . . . GND -0.3V to VCC +0.3V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 Thermal Resistance (Typical, Note 1) θJA (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118 Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC (SOIC - Lead Tips Only) Operating Conditions Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10% Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range . . . . . . . . . . . . . . . . . . . 0oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief 379 for details. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS - 5 - mA VCC SUPPLY CURRENT Nominal Supply ICC UGATE and LGATE Open POWER-ON RESET Rising VCC Threshold VOCSET = 4.5V - - 10.4 V Falling VCC Threshold VOCSET = 4.5V 8.2 - - V - 1.26 - V Rising VOCSET Threshold OSCILLATOR Free Running Frequency RT = OPEN 185 200 215 kHz Total Variation 6kΩ < RT to GND < 200kΩ -15 - +15 % - 1.9 - VP-P ΔVOSC Ramp Amplitude RT = Open REFERENCE AND DAC DAC(VID0-VID4) Input Low Voltage - - 0.8 V DAC(VID0-VID4) Input High Voltage 2.0 - - V DACOUT Voltage Accuracy -1.0 - +1.0 % - 88 - dB - 15 - MHz - 6 - V/μs 350 500 - mA - 5.5 10 Ω 300 450 - mA - 3.5 6.5 Ω - 115 120 % VOCSET = 4.5VDC 170 200 230 μA VSEN = 5.5V, VOVP = 0V 60 - - mA - 10 - μA ERROR AMPLIFIER DC Gain Gain-Bandwidth Product GBW Slew Rate SR COMP = 10pF GATE DRIVERS Upper Gate Source IUGATE VBOOT - VPHASE = 12V, VUGATE = 6V Upper Gate Sink RUGATE ILGATE = 0.3A Lower Gate Source ILGATE VCC = 12V, VLGATE = 6V Lower Gate Sink RLGATE ILGATE = 0.3A PROTECTION Over-Voltage Trip (VSEN/DACOUT) OCSET Current Source IOCSET OVP Sourcing Current IOVP Soft Start Current ISS POWER GOOD Upper Threshold (VSEN /DACOUT) VSEN Rising 106 - 111 % Lower Threshold (VSEN /DACOUT) VSEN Falling 89 - 94 % Hysteresis (VSEN /DACOUT) PGOOD Voltage Low VPGOOD 3 Upper and Lower Threshold - 2 - % IPGOOD = -5mA - 0.5 - V HIP6014 Typical Performance Curves 80 CGATE = 3300pF 70 60 RT PULLUP TO +12V ICC (mA) RESISTANCE (kΩ) 1000 100 50 CUPPER = CLOWER = CGATE 40 CGATE = 1000pF 30 10 20 RT PULLDOWN TO VSS CGATE = 10pF 10 10 100 1000 0 100 200 SWITCHING FREQUENCY (kHz) FIGURE 1. RT RESISTANCE vs FREQUENCY 300 400 500 600 700 800 SWITCHING FREQUENCY (kHz) 900 1000 FIGURE 2. BIAS SUPPLY CURRENT vs FREQUENCY Functional Pin Description VID0-4 (Pins 4-8) SS 3 18 VCC VID0 4 17 LGATE VID1 5 16 PGND VID0-4 are the input pins to the 5-bit DAC. The states of these five pins program the internal voltage reference (DACOUT). The level of DACOUT sets the converter output voltage. It also sets the PGOOD and OVP thresholds. Table 1 specifies DACOUT for the 32 combinations of DAC inputs. VID2 6 15 BOOT COMP (Pin 9) and FB (Pin 10) VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD COMP and FB are the available external pins of the error amplifier. The FB pin is the inverting input of the error amplifier and the COMP pin is the error amplifier output. These pins are used to compensate the voltage-control feedback loop of the converter. VSEN 1 20 RT OCSET 2 19 OVP FB 10 11 GND VSEN (Pin 1) GND (Pin 11) This pin is connected to the converters output voltage. The PGOOD and OVP comparator circuits use this signal to report output voltage status and for overvoltage protection. Signal ground for the IC. All voltage levels are measured with respect to this pin OCSET (Pin 2) PGOOD is an open collector output used to indicate the status of the converter output voltage. This pin is pulled low when the converter output is not within ±10% of the DACOUT reference voltage. Exception to this behavior are the cases where the VID pins combination yield a 0V converter output; in these cases PGOOD asserts a high level. Connect a resistor (ROCSET) from this pin to the drain of the upper MOSFET. ROCSET, an internal 200μA current source (IOCS), and the upper MOSFET on-resistance (rDS(ON)) set the converter over-current (OC) trip point according to the following equation: I OCS • R OCSET I PEAK = -------------------------------------------r DS ( ON ) PGOOD (Pin 12) PHASE (Pin 13) An over-current trip cycles the soft-start function. SS (Pin 3) Connect a capacitor from this pin to ground. This capacitor, along with an internal 10μA current source, sets the softstart interval of the converter. 4 Connect the PHASE pin to the upper MOSFET source. This pin is used to monitor the voltage drop across the MOSFET for over-current protection. This pin also provides the return path for the upper gate drive. UGATE (Pin 14) Connect UGATE to the upper MOSFET gate. This pin provides the gate drive for the upper MOSFET. HIP6014 BOOT (Pin 15) This pin provides bias voltage to the upper MOSFET driver. A bootstrap circuit may be used to create a BOOT voltage suitable to drive a standard N-Channel MOSFET. PGND (Pin 16) This is the power ground connection. Tie the lower MOSFET source to this pin. LGATE (Pin 17) Connect LGATE to the lower MOSFET gate. This pin provides the gate drive for the lower MOSFET. VCC (Pin 18) Provide a 12V bias supply for the chip to this pin. OVP (Pin 19) The OVP pin can be used to drive an external SCR in the event of an overvoltage condition. Output rising 15% more than the DAC-set voltage triggers a high output on this pin and disables PWM gate drive circuitry. (COMP pin) and reference input (+ terminal of error amp) to the SS pin voltage. Figure 3 shows the soft start interval with CSS = 0.1μF. Initially the clamp on the error amplifier (COMP pin) controls the converter’s output voltage. At t1 in Figure 3, the SS voltage reaches the valley of the oscillator’s triangle wave. The oscillator’s triangular waveform is compared to the ramping error amplifier voltage. This generates PHASE pulses of increasing width that charge the output capacitor(s). This interval of increasing pulse width continues to t2 . With sufficient output voltage, the clamp on the reference input controls the output voltage. This is the interval between t2 and t3 in Figure 3. At t3 the SS voltage exceeds the DACOUT voltage and the output voltage is in regulation. This method provides a rapid and controlled output voltage rise. The PGOOD signal toggles ‘high’ when the output voltage (VSEN pin) is within ±5% of DACOUT. The 2% hysteresis built into the power good comparators prevents PGOOD oscillation due to nominal output voltage ripple. RT (Pin 20) This pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from this pin to GND, the nominal 200kHz switching frequency is increased according to the following equation: PGOOD (2V/DIV.) 0V SOFT-START (1V/DIV.) 6 5 • 10 Fs ≈ 200kHz + --------------------R T ( kΩ ) (RT to GND) Conversely, connecting a pull-up resistor (RT) from this pin to VCC reduces the switching frequency according to the following equation: 0V 0V t1 t2 t3 TIME (5ms/DIV.) 7 4 • 10 Fs ≈ 200kHz – --------------------R T ( kΩ ) OUTPUT VOLTAGE (1V/DIV.) (RT to 12V) Functional Description Initialization The HIP6014 automatically initializes upon receipt of power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the input supply voltages. The POR monitors the bias voltage at the VCC pin and the input voltage (VIN) on the OCSET pin. The level on OCSET is equal to VIN less a fixed voltage drop (see over-current protection). The POR function initiates soft start operation after both input supply voltages exceed their POR thresholds. For operation with a single +12V power source, VIN and VCC are equivalent and the +12V power source must exceed the rising VCC threshold before POR initiates operation. Soft Start The POR function initiates the soft start sequence. An internal 10μA current source charges an external capacitor (CSS) on the SS pin to 4V. Soft start clamps the error amplifier output 5 FIGURE 3. SOFT START INTERVAL Over-Current Protection The over-current function protects the converter from a shorted output by using the upper MOSFET’s on-resistance, rDS(ON) to monitor the current. This method enhances the converter’s efficiency and reduces cost by eliminating a current sensing resistor. The over-current function cycles the soft-start function in a hiccup mode to provide fault protection. A resistor (ROCSET) programs the over-current trip level. An internal 200μA current sink develops a voltage across ROCSET that is referenced to VIN . When the voltage across the upper MOSFET (also referenced to VIN) exceeds the voltage across ROCSET , the over-current function initiates a soft-start sequence. The softstart function discharges CSS with a 10μA current sink and inhibits PWM operation. The soft-start function recharges CSS , and PWM operation resumes with the error amplifier clamped to the SS voltage. Should an overload occur while recharging CSS, the soft start function inhibits PWM operation while fully charging CSS to 4V to complete its cycle. Figure 4 shows this SOFT-START HIP6014 little power with this method. The measured input power for the conditions of Figure 4 is 2.5W. 4V The over-current function will trip at a peak inductor current (IPEAK) determined by: 2V I OCSET • R OCSET I PEAK = --------------------------------------------------r DS ( ON ) OUTPUT INDUCTOR 0V 15A where IOCSET is the internal OCSET current source (200μA typical). The OC trip point varies mainly due to the MOSFET’s rDS(ON) variations. To avoid over-current tripping in the normal operating load range, find the ROCSET resistor from the equation above with: 10A 5A 0A 1. The maximum rDS(ON) at the highest junction temperature. TIME (20ms/DIV.) 2. The minimum IOCSET from the specification table. 3. Determine IPEAK for I PEAK > I OUT ( MAX ) + ( ΔI ) ⁄ 2 , where ΔI is the output inductor ripple current. FIGURE 4. OVER-CURRENT OPERATION operation with an overload condition. Note that the inductor current increases to over 15A during the CSS charging interval and causes an over-current trip. The converter dissipates very For an equation for the ripple current see the section under component guidelines titled ‘Output Inductor Selection’. TABLE 1. OUTPUT VOLTAGE PROGRAM PIN NAME PIN NAME VID4 VID3 VID2 VID1 VID0 NOMINAL OUTPUT VOLTAGE DACOUT VID4 VID3 VID2 VID1 VID0 NOMINAL OUTPUT VOLTAGE DACOUT 0 1 1 1 1 0 1 1 1 1 1 0 0 1 1 1 0 0 1 1 1 1 0 2.1 0 1 1 0 1 0 1 1 1 0 1 2.2 0 1 1 0 0 0 1 1 1 0 0 2.3 0 1 0 1 1 0 1 1 0 1 1 2.4 0 1 0 1 0 0 1 1 0 1 0 2.5 0 1 0 0 1 0 1 1 0 0 1 2.6 0 1 0 0 0 0 1 1 0 0 0 2.7 0 0 1 1 1 0 1 0 1 1 1 2.8 0 0 1 1 0 0 1 0 1 1 0 2.9 0 0 1 0 1 1.80 1 0 1 0 1 3.0 0 0 1 0 0 1.85 1 0 1 0 0 3.1 0 0 0 1 1 1.90 1 0 0 1 1 3.2 0 0 0 1 0 1.95 1 0 0 1 0 3.3 0 0 0 0 1 2.00 1 0 0 0 1 3.4 0 0 0 0 0 2.05 1 0 0 0 0 3.5 NOTE: 0 = connected to GND or VSS, 1 = connected to VDD through pull-up resistors A small ceramic capacitor should be placed in parallel with ROCSET to smooth the voltage across ROCSET in the presence of switching noise on the input voltage. Output Voltage Program The output voltage of a HIP6014 converter is programmed to discrete levels between 1.8VDC and 3.5VDC . The 6 voltage identification (VID) pins program an internal voltage reference (DACOUT) with a TTL-compatible 5-bit digital-toanalog converter (DAC). The level of DACOUT also sets the PGOOD and OVP thresholds. Table 1 specifies the DACOUT voltage for the 32 different combinations of connections on the VID pins. The output voltage should not be adjusted while the converter is delivering power. HIP6014 All VID pin combinations resulting in a 0V output setting activate the Power-On Reset function and disable the gate drive circuitry. For these specific VID combinations, though, PGOOD asserts a high level. This unusual behavior has been implemented in order to allow for operation in dualmicroprocessor systems by AND-ing the PGOOD signals from the two individual power converters. Application Guidelines Layout Considerations As in any high frequency switching converter, layout is very important. Switching current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces. The critical components should be located as close together as possible, using ground plane construction or single point grounding. close to the SS pin because the internal current source is only 10μA. Provide local VCC decoupling between VCC and GND pins. Locate the capacitor, CBOOT as close as practical to the BOOT and PHASE pins. Feedback Compensation Figure 7 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage (VOUT) is regulated to the Reference voltage level. The error amplifier (Error Amp) output (VE/A) is compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated (PWM) wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (LO and CO). BOOT CBOOT VCC SS +12V Q2 CO CVCC GND HIP6014 FIGURE 6. PRINTED CIRCUIT BOARD SMALL SIGNAL LAYOUT GUIDELINES LO Q2 D2 CIN VOUT CO VIN DRIVER OSC LOAD Q1 PHASE LGATE LO VOUT CSS UGATE Q1 PHASE HIP6014 VIN +VIN D1 LOAD Remove input power before changing the output voltage. Adjusting the output voltage during operation could toggle the PGOOD signal and exercise the overvoltage protection. PWM COMPARATOR PGND LO - DRIVER + Δ VOSC PHASE RETURN FIGURE 5. PRINTED CIRCUIT BOARD POWER AND GROUND PLANES OR ISLANDS Figure 5 shows the critical power components of the converter. To minimize the voltage overshoot the interconnecting wires indicated by heavy lines should be part of ground or power plane in a printed circuit board. The components shown in Figure 5 should be located as close together as possible. Please note that the capacitors CIN and CO each represent numerous physical capacitors. Locate the HIP6014 within 3 inches of the MOSFETs, Q1 and Q2. The circuit traces for the MOSFETs’ gate and source connections from the HIP6014 must be sized to handle up to 1A peak current. Figure 6 shows the circuit traces that require additional layout consideration. Use single point and ground plane construction for the circuits shown. Minimize any leakage current paths on the SS PIN and locate the capacitor, CSS 7 CO ESR (PARASITIC) ZFB VE/A VOUT - ZIN + ERROR AMP REFERENCE DETAILED COMPENSATION COMPONENTS C2 C1 ZFB VOUT ZIN C3 R2 R3 R1 COMP - FB + HIP6014 DACOUT FIGURE 7. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN HIP6014 Modulator Break Frequency Equations 1 F LC = ---------------------------------------2π • L O • C O 1 F ESR = ----------------------------------------2π • ESR • C O The compensation network consists of the error amplifier (internal to the HIP6014) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network’s poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 7. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth 2. Place 1ST Zero Below Filter’s Double Pole (~75% FLC) 3. Place 2ND Zero at Filter’s Double Pole 4. Place 1ST Pole at the ESR Zero 5. Place 2ND Pole at Half the Switching Frequency 6. Check Gain against Error Amplifier’s Open-Loop Gain 7. Estimate Phase Margin - Repeat if Necessary Compensation Break Frequency Equations 1 F Z1 = --------------------------------2π • R 2 • C 1 1 F P1 = ----------------------------------------------------⎛ C1 • C2⎞ 2π • R 2 • ⎜ ---------------------⎟ ⎝ C 1 + C 2⎠ 1 F Z2 = ---------------------------------------------------2π • ( R 1 + R 3 ) • C 3 1 F P2 = --------------------------------2π • R 3 • C 3 Figure 8 shows an asymptotic plot of the DC-DC converter’s gain vs. frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 8. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the log-log graph of Figure 8 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. 8 100 FZ1 FZ2 FP1 FP2 80 OPEN LOOP ERROR AMP GAIN 60 GAIN (dB) The modulator transfer function is the small-signal transfer function of VOUT/VE/A. This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR. The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage ΔVOSC . 40 20 20LOG (R2/R1) 0 20LOG (VIN/ΔVOSC) MODULATOR GAIN -20 -40 -60 COMPENSATION GAIN CLOSED LOOP GAIN FLC 10 100 1K FESR 10K 100K 1M 10M FREQUENCY (Hz) FIGURE 8. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN Component Selection Guidelines Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern microprocessors produce transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. For example, Intel recommends that the high frequency decoupling for the Pentium Pro be composed of at least forty (40) 1μF ceramic capacitors in the 1206 surface-mount package. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a HIP6014 suitable component. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. Output Inductor Selection The output inductor is selected to meet the output voltage ripple requirements and minimize the converter’s response time to the load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: ΔI = VIN - VOUT Fs x L • VOUT ΔVOUT = ΔI x ESR VIN Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the HIP6014 will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time required to slew the inductor current from an initial current value to the transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitor. Minimizing the response time can minimize the output capacitance required. The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: tRISE = L x ITRAN VIN - VOUT tFALL = L x ITRAN VOUT where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the application or removal of load and dependent upon the DACOUT setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. With a +12V input, and output voltage level equal to DACOUT, tFALL is the longest response time. Input Capacitor Selection Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use small ceramic capacitors for high frequency decoupling and bulk capacitors to supply the current needed each time Q1 turns on. Place the small ceramic capacitors physically close to the MOSFETs and between the drain of Q1 and the source of Q2. 9 The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. The RMS current rating requirement for the input capacitor of a buck regulator is approximately 1/2 the DC load current. For a through hole design, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MVGX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. MOSFET Selection/Considerations The HIP6014 requires 2 N-Channel power MOSFETs. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. In high-current applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. The conduction losses are the largest component of power dissipation for both the upper and the lower MOSFETs. These losses are distributed between the two MOSFETs according to duty factor (see the equations below). Only the upper MOSFET has switching losses, since the Schottky rectifier clamps the switching node before the synchronous rectifier turns on. These equations assume linear voltagecurrent transitions and do not adequately model power loss due the reverse-recovery of the lower MOSFET’s body diode. The gate-charge losses are dissipated by the HIP6014 and don't heat the MOSFETs. However, large gatecharge increases the switching interval, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal-resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. PUPPER = Io2 x rDS(ON) x D + 1 Io x VIN x tSW x FS 2 PLOWER = Io2 x rDS(ON) x (1 - D) Where: D is the duty cycle = VOUT / VIN , tSW is the switch ON time, and FS is the switching frequency. HIP6014 Standard-gate MOSFETs are normally recommended for use with the HIP6014. However, logic-level gate MOSFETs can be used under special circumstances. The input voltage, upper gate drive level, and the MOSFET’s absolute gate-tosource voltage rating determine whether logic-level MOSFETs are appropriate. Figure 9 shows the upper gate drive (BOOT pin) supplied by a bootstrap circuit from VCC. The boot capacitor, CBOOT develops a floating supply voltage referenced to the PHASE pin. This supply is refreshed each cycle to a voltage of VCC less the boot diode drop (VD) when the lower MOSFET, Q2 turns on. Logic-level MOSFETs can only be used if the MOSFET’s absolute gate-to-source voltage rating exceeds the maximum voltage applied to VCC . +12V VCC HIP6014 DBOOT +5V or +12V + VD BOOT CBOOT UGATE - Schottky Selection Rectifier D2 is a clamp that catches the negative inductor swing during the dead time between turning off the lower MOSFET and turning on the upper MOSFET. The diode must be a Schottky type to prevent the lossy parasitic MOSFET body diode from conducting. It is acceptable to omit the diode and let the body diode of the lower MOSFET clamp the negative inductor swing, but efficiency will drop one or two percent as a result. The diode's rated reverse breakdown voltage must be greater than the maximum input voltage. Q1 PHASE + Figure 10 shows the upper gate drive supplied by a direct connection to VCC . This option should only be used in converter systems where the main input voltage is +5VDC or less. The peak upper gate-to-source voltage is approximately VCC less the input supply. For +5V main power and +12VDC for the bias, the gate-to-source voltage of Q1 is 7V. A logiclevel MOSFET is a good choice for Q1 and a logic-level MOSFET can be used for Q2 if its absolute gate-to-source voltage rating exceeds the maximum voltage applied to VCC. NOTE: VG-S ≈ VCC -VD Q2 LGATE D2 PGND NOTE: VG-S ≈ VCC GND FIGURE 9. UPPER GATE DRIVE - BOOTSTRAP OPTION +12V +5V OR LESS VCC HIP6014 BOOT UGATE Q1 PHASE - + LGATE PGND GND NOTE: VG-S ≈ VCC -5V Q2 D2 NOTE: VG-S ≈ VCC FIGURE 10. UPPER GATE DRIVE - DIRECT VCC DRIVE OPTION 10 HIP6014 HIP6014 DC-DC Converter Application Circuit Application Note AN9672. Although the Application Note details the HIP6004, the same evaluation platform can be used to evaluate the HIP6014. Figure 11 shows an application circuit of a DC-DC Converter for an Intel Pentium Pro microprocessor. Detailed information on the circuit, including a complete Bill-ofMaterials and circuit board description, can be found in VIN = +5V OR +12V F1 L1 - 1μH C1 5x 1000μF 2x 1μF 2N6394 +12V 2K D1 0.1μF VSEN 1 RT VID0 VID1 VID2 VID3 VID4 FB OVP 18 19 2 OCSET MONITOR AND PROTECTION SS 3 0.1μF 1000pF VCC 20 4 5 6 7 8 1K 12 PGOOD 15 BOOT OSC 14 UGATE 13 PHASE HIP6014 - - 17 LGATE + + 16 PGND 9 11 COMP 2.2nF GND 20K 8.2nF 0.1μF 15 Component Selection Notes: C0 - C9 - Each 1000μF 6.3W VDC, Sanyo MV-GX or Equivalent C1 - C5 - Each 330μF 25W VDC, Sanyo MV-GX or Equivalent L2 - Core: Micrometals T50-52B; Each Winding: 10 Turns of 16AWG L1 - Core: Micrometals T50-52; Winding: 5 Turns of 18AWG D1 - 1N4148 or Equivalent D2 - 3A, 40V Schottky, Motorola MBR340 or Equivalent Q1 , Q2 - Intersil MOSFET; RFP70N03 FIGURE 11. PENTIUM PRO DC-DC CONVERTER 11 L2 3μH D/A 10 1.33K 0.1μF Q1 Q2 D2 CO 9x 1000μF +VO HIP6014 Small Outline Plastic Packages (SOIC) M20.3 (JEDEC MS-013-AC ISSUE C) N 20 LEAD WIDE BODY SMALL OUTLINE PLASTIC PACKAGE INDEX AREA 0.25(0.010) M H B M INCHES E -B- 1 2 3 L SEATING PLANE -A- h x 45o A D -C- e A1 B C 0.10(0.004) 0.25(0.010) M C A M SYMBOL MIN MAX MIN MAX NOTES A 0.0926 0.1043 2.35 2.65 - A1 0.0040 0.0118 0.10 0.30 - B 0.013 0.0200 0.33 0.51 9 C 0.0091 0.0125 0.23 0.32 - D 0.4961 0.5118 12.60 13.00 3 E 0.2914 0.2992 7.40 7.60 4 e µα B S 0.050 BSC 1.27 BSC - H 0.394 0.419 10.00 10.65 - h 0.010 0.029 0.25 0.75 5 L 0.016 0.050 0.40 1.27 6 N α NOTES: MILLIMETERS 20 0o 20 8o 0o 7 8o 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. Rev. 0 12/93 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch) 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 12