AN960

AN960
New Components and Design Methods Bring Intelligence to
Battery Charger Applications
Author:
Terry Cleveland and
Catherine Vannicola,
Microchip Technology Inc.
INTRODUCTION
New design methods and components bring high
intelligence to battery charger and power-management
applications. When developing intelligent power
management systems, very complex and cumbersome
analog-only solutions are now old school. When combining low-cost microcontroller’s with analog-attach,
high-speed Pulse Width Modulators (PWMs), the benefits of a mixed signal design can be realized. In the
past, a complex power-management system was
developed using a high-speed analog PWM combined
with logic and specialty analog-only circuits. In some
cases, non-flexible, off-the-shelf solutions are available
for a price. For most applications, they have neither the
features, nor the capability, to meet any special
requirements. With the addition of a microcontroller, the
new design method can adapt to almost any system
command, environmental conditions or system-level
fault.
With the combination of a microcontroller and a highspeed analog PWM, the designer can enjoy the benefits of programmability with the peace of mind that the
power train is being controlled and protected by a
reliable high-speed analog loop. When external conditions warrant a change in output, the digital microcontroller can adjust the output of the supply, the switching
frequency of the supply, the minimum off time of the
power train switch, power-up soft start, take action in
the event of a system-level fault, etc. With the broad
range of Microchip’s PICmicro® microcontroller product
line, the microcontroller can be sized for the job. In
many applications, a microcontroller is already
resident. By adding the MCP1630 analog, high-speed
PWM, a power train can be easily added to the design.
This application note will describe a typical intelligent
battery charger power system application. As with most
real life applications, there are many demands made
on the power system designer to protect the system in
the case of battery removal, plugging the battery in
backwards, reverse polarity at the input, a battery
shorting and even more unimaginable situations. A
complete battery charger, fuel gauge system design
will be presented as an example of the mixed signal
 2004 Microchip Technology Inc.
design method. Battery reference material and basic
switchmode power supply converter trade-offs are
covered in the beginning of this application note.
BATTERY BACKGROUND
Definitions:
The anode of a cell during discharge is the negative
electrode. During charge, the anode is the positive
electrode. The anode supplies electrons to the load.
The C rate is the battery’s charge or discharge current
expressed as a multiple of the capacity. For example, if
a battery has a capacity of 500 mAhr, a charge rate (or
“C” rate) of 2C would imply a charge current of 1A.
Capacity is measured in units of Amp-hours and can
be described as the discharge current necessary to
reach the end voltage after one hour.
The cathode of a cell during discharge is the positive
electrode. During charge, the cathode is the negative
electrode. The cathode accepts electrons from the
load.
Charge acceptance is the ability of a battery to accept
charge by converting provided electrical energy into
stored chemical energy.
The electrolyte conducts ions inside the cell between
the anode and the cathode. The electrolyte must be a
good ionic conductor but not be electrically conductive,
since this would cause short-circuiting. Most
electrolytes are liquids, although some are solids.
Memory effect is a temporary failure of a battery due
to repeated incomplete discharge. This causes the
battery to lose capacity. Capacity can be restored by a
few repeated cycles of full discharge and charge.
Self-discharge is the loss of charge of an unloaded
cell due to internal chemical reactions.
The service life of a secondary battery is defined as
the length of useful performance in years, called float
life, or the number of times it can be usefully charged
and discharged, called cycle life.
Trickle charge is a low charge rate used to maintain a
battery in a fully charged condition.
DS00960A-page 1
AN960
Figure 1 shows the basic battery cell during charge and
discharge. The anode and cathode are electrically
isolated by a mechanical separator to prevent short
circuiting. The electrolyte surrounds the anode and the
cathode and can permeate the separator. During the
discharge cycle, positive ions flow from the anode to
the cathode, while negative ions flow from the cathode
to the anode. Electrons flow through the external load
from the anode to the cathode. During the charge
cycle, this process is reversed.
Discharge
Load
Electrolyte
Electrons
Negative Ions
(–)
Anode
(+)
Cathode
Overall reaction:
Pb + PbO2 + 2H2 SO 4 → 2PbSO 4 + 2H 2 O
The lead acid battery is low in cost and available in
large quantities. Lead acid batteries also come in a
variety of sizes and designs. They have good high-rate
performance, reasonably good low/high temperature
performance and have over 70% efficiency. They have
good charge retention and, since they are sealed, are
maintenance-free.
The lead acid battery’s main disadvantages are its lowcycle life (typically between 50 and 500 cycles), its
limited energy density (30-40 Wh/kg) and its weight.
Lead acid batteries should not be stored for long
periods of time in a discharged state that can lead to
irreversible polarization of the electrodes. The evolution of hydrogen in some designs can be an explosion
hazard, while the evolution of stibene and arsine in
other designs can be a health hazard.
Positive Ions
SEALED NICKEL CADMIUM
Separator
FIGURE 1:
Battery Cell.
The sealed nickel cadmium cell has a metallic cadmium negative electrode and a positive electrode made
from nickel oxyhydroxide, with potassium hydroxide as
the electrolyte. As the cell is discharged, the positive
plate reduces to nickel hydroxide and the negative
plate oxidizes to cadmium hydroxide.
Negative electrode:
Chemistries
There are a number of different battery chemistries
available to choose from when selecting a rechargeable cell. Three of these choices are: sealed lead acid,
nickel cadmium and nickel metal hydride. Each battery
chemistry has advantages and disadvantages, with
each having different charging requirements.
The sealed lead acid battery uses lead dioxide in the
positive electrode and metallic lead in the negative
electrode. The electrolyte is a sulfuric acid solution.
During discharge, both the positive and negative
electrodes convert to lead sulfate and water is
generated. During charge, the process is reversed. If
the cell is overcharged, hydrogen and oxygen gas are
produced and there is a loss of water, resulting in a loss
of capacity. The chemical processes during discharge
are shown below.
Negative electrode:
Pb → Pb
Pb
+
Positive electrode:
NiOOH + H2 O + e → Ni ( OH )2 + OH
-
Overall reaction:
Cd + 2NiOOH + 2H 2 O → Cd ( OH ) 2 + 2Ni ( OH ) 2
SEALED LEAD ACID
2+
Cd + 2OH → Cd ( OH )2 + 2e
SO 42-
2+
+ 2e
→ PbSO 4
The sealed nickel cadmium battery has a long cycle
life, good low-temperature and high-rate performance
capability, rapid recharge capability and a long shelf life
in any state of charge. The sealed design has features
to prevent a build-up of pressure if overcharge should
occur. It requires no maintenance, with the exception of
recharging.
Sealed nickel cadmium batteries are subject to the
memory effect. They have a higher cost than lead acid
batteries and are outperformed by the lead acid battery
at high temperatures. They have poor charge retention
and have a lower capacity than other competitive
batteries. There are also environmental concerns due
to the use of cadmium, a heavy metal.
Positive electrode:
+
PbO 2 + 4H + 2e → Pb
Pb
DS00960A-page 2
2+
+
SO 42-
2+
+ 2H 2 O
→ PbSO 4
 2004 Microchip Technology Inc.
AN960
SEALED NICKEL METAL HYDRIDE
The nickel metal hydride cell employs nickel oxyhydroxide in the positive electrode and a metal hydride rather
than cadmium in the negative electrode. The electrolyte
is potassium hydroxide. During discharge, the nickel
oxyhydroxide of the positive electrode is reduced to
nickel hydroxide and the metal hydride of the negative
electrode is oxidized to the metal alloy (M).
Negative electrode:
-
MH – OH → M + H 2 O + e
Positive electrode:
NiOOH + H2 O + e → Ni ( OH ) 2 + OH
Table 1 shows comparisons between
properties of different battery chemistries.
TABLE 1:
BATTERY TECHNOLOGY
COMPARISONS
Electrochemistry
PbAcid
Ni-Cd
NiMH
Nominal Cell
Voltage
V/cell
2.0
1.2
1.2
End of Life
Voltage
V/cell
1.7
1.0
1.0
Energy Density
W-hr/kg
W-hr/Itr
35
80
50
100
60
130
Self-Discharge
Rate
%/month
2
15
30
-
Overall reaction:
relevant
MH + NiOOH → M – Ni ( OH ) 2
Cycle Life
25% dischrg
100% dischrg
1200
300
2000
500
500
500
The metal alloy used in nickel metal hydride batteries
must be stable over a large number of charge/
discharge cycles. It also must be able to store hydrogen to obtain high energy density and battery capacity.
The metal alloy must have high electrochemical reactivity, good kinetic properties for high-rate performance,
high oxidation resistance and must be stable in an alkaline electrolyte. Two common types of metallic alloys
are: rare earth alloys based on lanthanum nickel and
alloys of titanium and zirconium. There are other possible substitutes that can be used to improve certain
aspects of the alloy’s performance.
Cost/W-hr
$/W-hr
0.50
1.00
1.50
Nickel metal hydride batteries have higher capacity
than the nickel cadmium batteries. There is minimal
environmental concern because they are cadmiumfree. They have a long shelf life in any state of charge
and are much smaller and lighter than the nickel
cadmium. They also have less susceptibility to the
memory effect.
The high-rate performance of the nickel metal hydride
battery is not as good as that of the nickel cadmium
battery. The negative electrodes of the nickel metal
hydride battery are more expensive. There is also less
capacity to deliver high peak currents, greater risk of
damage due to overcharging and they have a higher
self-discharge rate.
 2004 Microchip Technology Inc.
Recommended Charging for NiMH
Panasonic® recommends that its nickel metal hydride
batteries be rapidly charged between 0°C and +40°C
with a maximum current of 1 CmA. Rapid charge
should be between 0.5 CmA and 1 CmA. Charging
batteries above 1 CmA could cause the safety vents to
be activated due to a rise in the internal pressure of
batteries. If the battery is outside the recommended
temperature range, it must be trickle charged. Rapid
charge current may be applied when the voltage-percell is between 0.8V and 1.8V. Outside of this range,
trickle charge must be applied. If too high a charge
current is applied to deeply discharged batteries, the
full capacity of the batteries may not be achieved
during charge. Trickle charge should be between 0.033
and 0.05 CmA to avoid a harmful temperature rise.
After trickle charge, a transition current before rapid
charge of 0.2~0.3 CmA must be applied.
Rapid charge must end and trickle charge must be reapplied when the voltage drops 5 to 10 mV/cell during
charge. This drop in voltage means that the battery is
fully charged. Also, rapid charge must be stopped and
trickle charge applied, if the battery voltage exceeds
1.8V/cell, or if the temperature rises 1 to 2°C/min. The
temperature of a charging nickel metal hydride battery
must not exceed 55°C for A, AA and D-sized batteries,
50°C for QA, AAA and prismatic sizes, and 60°C for L-A,
L-fatA and SC sizes. Any time the temperature rises
above these prescribed values, the charge must be
switched to trickle to avoid impairing cycle life.
DS00960A-page 3
AN960
In order to measure the correct change in voltage,
there must be at least a 10 minute delay after the
initiation of rapid charge and the beginning of voltage
detection. This is done to avoid measuring voltage
swings when rapid charge is commenced.
Even extensive trickle charge can lead to harmful
overcharging, causing deterioration of the battery.
Therefore, the total time the battery spends being
charged should be limited to between 10 and 20 hours.
Stage 1:
Trickel Charge
Stage 2:
Fast Charge
VCELL
0.8V
VCELL(V)
ICH(A)
T(°C)
1.0C
Stage 3:
End Fast
Charge
Power Topology Options
BUCK CONVERTER
The buck converter or step-down converter can only
step the voltage down from a higher voltage level to a
lower voltage level. In this application, this may or may
not be acceptable. The minimum input voltage is 8V,
while the maximum output voltage is 6.4V, plus current
sense voltage and voltage drop across any protection
switch used to disconnect the battery in the event of an
over-discharge condition.
Stage 4:
Overcharge
Buck Converter
-∆V
Body Diode Path
L
+12V Input
ICH
+VBATT
C
0.2C
T (Cell Temperature)
0.05C
1hr
FIGURE 2:
4 NiMH Batteries
1.5hr
Charge Stages.
Chopped Input Current
Figure 2 shows the stages that a charging nickel metal
hydride battery goes through and what the battery
temperature, current and voltage characteristics are.
The charge termination method can be any one of the
above or a combination. Battery manufacturers recommend using this combination method of termination to
maximize the life and capacity of the NiMH battery.
FB
ISENSE
FIGURE 3:
SELECTING THE BEST POWER
TRAIN TOPOLOGY
The most important decision when developing a power
management system is selecting the “best” topology. In
most applications, there are several alternatives and
they should all be considered prior to making the final
selection. This decision is based on total system cost
and performance. Some topologies add complexity, but
this may result in savings that are beneficial to a
specific application.
DS00960A-page 4
Buck Converter.
When compared to other topologies, the buck
converter is considered low in complexity. It requires a
single switch, single inductor and diode (in addition to
input filter and output capacitor). When looking at the
buck converter for charging applications, there are a
couple of key drawbacks to consider.
• There is a path from the battery back to the input
through the buck switch body diode (when using a
MOSFET switch). This is resolved by either
adding a blocking diode at the input or, preferably,
at the output of the buck regulator.
• The input current is discontinuous or pulsing. This
can lead to another filtering stage at the input of
the converter to meet system-level EMI specifications. This filtering stage typically consists of an
inductor and capacitor to filter the pulsed input
current.
• In the event of a shorted buck switch (high-side),
there is no way to limit the current into the battery.
This situation is hazardous and should be avoided
by using a protection device (fuse).
 2004 Microchip Technology Inc.
AN960
BUCK-BOOST CONVERTER
FLYBACK CONVERTER
The buck-boost converter can be configured several
different ways. In order to keep the input polarity the
same as the output polarity, a minimum of two switches
and two diodes are used. One of the switches is in the
high-side (similar to the buck converter), while one
switch is in the low-side (similar to the boost regulator).
This increases the cost and complexity of the buckboost topology when compared to most other solutions.
A flyback converter utilizes a coupled or two winding
inductor. The flyback converter can provide true
galvanic isolation from input to output for applications
that have hazardous input voltages. This is commonly
used in off-line applications (for relatively low power
converters and battery chargers).
Flyback Converter
+VBATT
+12V Input
Buck - Boost Converter
CIN
1
COUT
1
+VBATT
+12V Input
COUT
CIN
L
Chopped Input Current
EXT
Blocking Diode
Blocking Diode
4 NiMH
Batteries
CS
0.15Ω
4 NiMH
Batteries
Chopped Input Current
FB
ISENSE
FB
ISENSE
FIGURE 5:
Flyback Converter.
Advantages of the Flyback Converter:
FIGURE 4:
Buck-Boost Regulator.
Advantages of the Buck-Boost Converter:
• Step-up and step-down capability.
• The output stage rectifier diode can be used as
the reverse blocking diode.
Disadvantages of the Buck-Boost Converter:
• Discontinuous or chopping input current typically
requires additional input filtering.
• In the event of a shorted high-side switch, there is
no way to limit the current into the battery (similar
to buck converter). This situation is hazardous
and should be avoided by using a protection
device (fuse).
• Two switches and two diodes necessary for buckboost capability without inverting output polarity.
 2004 Microchip Technology Inc.
• Input voltage can be greater or less than the
output voltage. The flyback has the same transfer
function as the buck-boost converter (1:1 turns
ratio coupled inductor).
• Low-side switch and low-side current sense
simplify the drive scheme and peak current sense
circuitry.
• The blocking diode necessary for battery charger
applications is inherent in the flyback topology.
• Additional output voltages are easily added using
additional winding on the coupled inductor. By
adjusting the turn ratio of the secondary windings,
a large step-up or step-down ratio can be
accomplished.
Disadvantages of the Flyback Converter:
• The main disadvantage of the flyback converter is
related to the leakage inductance of the coupled
inductor. Every magnetic device that has more
than one winding will have a leakage inductance
from one winding to the other. When the Flyback
low side switch turns off, this leakage inductance
is not clamped. This unclamped inductance
requires the addition of a snubber or damping
circuit. For low-power, low-cost applications, this
snubber tends to be dissipative and requires
additional components.
DS00960A-page 5
AN960
• The peak voltage observed on the drain of the
low-side switch (when off) is equal to:
VIN + (NP/NS) * VOUT + VLK
for a coupled inductor where NP = # of primary
turns and NS = # of secondary turns. The higher
the leakage inductance between the primary and
secondary windings, the more energy available
to develop voltage overshoot. To a certain
degree, this can be suppressed using a dissipative snubber which is detrimental to converter
efficiency.
• The flyback converter uses an inductor with two
windings. For some applications, this can lead to
a custom magnetic design. For applications
where a 1:1 turn ratio can be used, there are
many magnetic suppliers that offer standard off
the shelf products.
SEPIC CONVERTER
The Single-Ended Primary Inductive Converter
(SEPIC) was developed primarily to have step-up and
step-down capability without inverting the polarity of the
regulated output. The main difference between the
SEPIC converter and the previously mentioned topologies is the addition of another energy storage element.
The SEPIC uses a coupling capacitor to store and
transfer energy.
SEPIC Converter
C1
Coupling Capacitor
+12V Input
CIN
1
Blocking Diode
+VBATT
COUT
1
EXT
CS
Input Current
0.15Ω
4 NiMH
Batteries
FB
ISENSE
FIGURE 6:
SEPIC Converter.
Advantages of the SEPIC converter:
• Low-side switch and low-side current sense
simplify the drive scheme and peak current sense
circuitry.
• The blocking diode necessary for battery charger
applications is inherent in the SEPIC topology
(similar to the flyback).
• Input voltage can be greater or less than the
output voltage.
• The input current to the SEPIC converter is continuous, similar to a boost regulator. For topologies having continuous input current, the input
EMI filtering can be significantly reduced (or is not
even necessary). The amount of noise generated
at the input of the SEPIC is much lower than in
the case of the buck, buck-boost or even the flyback converter. In applications where noise
reflected back to the input from the switching
regulator is a concern, a SEPIC converter has a
distinct advantage.
DS00960A-page 6
 2004 Microchip Technology Inc.
AN960
• The SEPIC converter provides capacitive primaryto-secondary isolation. In the event of a main
switch short, the output voltage is not shorted to
the input voltage, similar to the case of the buck
converter. This provides a level of protection for
the load. In the case of battery chargers, even
with the reverse-current blocking diode, there is
no protection in the event of a short circuit.
• The coupling capacitor clamps the winding
leakage inductance energy; no snubber circuit is
necessary.
Selecting Topology
After careful consideration, the SEPIC converter was
chosen for this 12V nominal input and four NiMH cell
battery load (500 mA fast charge current). The technical advantages of the SEPIC converter outweigh the
disadvantages for this low-voltage, low-power
application. The SEPIC input is designed to have low
ripple current so an additional input filter may not be
necessary. The SEPIC rectifying diode is also used as
the battery reverse current blocking diode.
The main disadvantages of the SEPIC converter are
the higher switch current and the addition of the
coupling capacitor. For lower voltage applications (sub
100V switch), new MOSFET technology has lower onresistance while keeping switch capacitance low. There
have also been significant improvements in lower
voltage (sub 50V) ceramic capacitor technology. In this
example, a SOT23 85 milli-ohm switch rated for 30V
and 0805 1 µF 25V ceramic capacitor was used. For
high-voltage, high-power applications, the higher
switch current and additional energy storage coupling
capacitor is both a technical, as well as cost, concern.
 2004 Microchip Technology Inc.
DS00960A-page 7
AN960
4-CELL NIMH BATTERY CHARGER
APPLICATION EXAMPLE
Application Description
Complete system to charge four series NiMH
rechargeable batteries at 500 mA. Additional features
include:
• Charge batteries to manufacturer’s recommended
profile. (This can change from manufacturer to
manufacturer and should be adhered to for maximum battery performance. Many off-the-shelf
switching battery chargers set the trickle charge
current as a ratio of fast charge current not fully
optimizing the battery life.)
• Terminate charge current based off change in
voltage versus time and/or change in battery
temperature versus time or a maximum length of
time.
• Minimize size and cost of power system used to
charge batteries.
• Maintain high-power conversion efficiency.
• Minimize quiescent current draw on the battery
during power outages.
• Provide visual indication of state of charge.
• Protect the battery and circuitry from:
- battery disconnect
- battery short circuit
- overcharge
- overdischarge
• Soft start the power train during startup or
recovery from a fault mode.
• Integrate battery energy used to provide “fuel
gauge” to system for display.
• Provide I2C™ standard communications to host
system.
• Minimize EMI and noise on the input source of the
battery charger.
A list of requirements on the power management
system is not an uncommon situation. All of the
features and functions listed above are necessary to
provide an efficient and safe solution to charging NiMH
batteries.
DS00960A-page 8
Battery Charger Design Specifications
• Input Voltage Range:
- 8V to 15V
• Charge Current:
- Fast Charge = 0.5C to 1C or 500 mA
- Trickle Charge = 0.1C or 50 mA
- Top off Charge = 0.05C or 25 mA
• NiMH Cell Voltage Charge Range:
- 0V to 0.8V per Cell Trickle Charge Only
- 0.8V to 1.6V per cell Fast Charge
• Charge Termination Method:
- Based on termination voltage (-dV/dt)
- Based on temperature (+dT/dt)
- Based on charge cycle elapsed time
- Based on total charge cycle time
The charge termination method can be any one of the
above, or a combination. Battery manufacturer’s
recommend using this combination method of termination to maximize the life and capacity of the NiMH
battery.
• Low quiescent current draw on battery during
input power failure, typically 29 µA.
• Charge mode indication using LED.
• Failure mode indication using LED.
• Capability to sense discharge current and provide
I2C communication to host for remaining capacity
information.
• Minimum of 80% efficiency.
• Minimize board area and height.
• Overvoltage protection in the event of an open
battery.
• Overcharge protection to prevent the battery from
becoming dangerously overcharged.
• Overcurrent protection in the event of a shorted
battery.
• Overtemperature protection to prevent the battery
from reaching too high a temperature during
charge.
• Soft-start capability by holding the reference
voltage low during power-up.
• A simple fuel gauge that has a dual MCP6042
amplifier, a 1-channel sense voltage and a
1-channel sense current.
• Flexibility to optimize the charging algorithm for
new battery technology and add proprietary
features by coding the microcontroller.
• Ability to adapt to environmental effects, such as
ambient temperature.
• The output voltage range, input voltage range and
output power can be scaled using different
external component set.
 2004 Microchip Technology Inc.
AN960
Power Train Design
W1
A
IW1
+
C1
+
D
Q1
VIN
W2
IW2
W1
B
+
IW1
W2
IW2
W1
+
VIN
+
ICHARGE
+ C1
VIN
C
COUT
COUT
+
ICHARGE
C
+ 1
IW1
W2
IW2
COUT
+
ICHARGE
Sum of Winding Currents
IW2
Diode Current
Switch Current
IW1+IW2
IOUT (Average)
IOUT x (VOUT/VIN)
IW1
IIN (AVERAGE)
tON
FIGURE 7:
tOFF
tON
tOFF
tON
tOFF
SEPIC Converter Inductor, Switch and Diode Currents.
 2004 Microchip Technology Inc.
DS00960A-page 9
AN960
POWER TRAIN CALCULATIONS
SEPIC Converter
+12V Input
CIN
ICHARGE
+VBATT
L
IW1 W1
C1
W2
ICOUT
COUT
IC1
Q1
ISW
4 NiMH
Batteries
IW1+ IW2
FB
VC1/L2
(VOUT) / L2
ISENSE
IW2
VIN/L1
(+VC1 + VOUT - VIN)/L1
stant fixed frequency switching. This will eliminate the
variable pulse skipping that can occur while minimizing
the ripple current at high load. This reduces the
reflected input ripple current without adding extra filtering.
The SEPIC switch (Q1) is turned on at the start of the
cycle (Figure 7B). As shown in Figure 7B, the ISW1 current will ramp up at a rate of VIN / L1 (where L1 is the
inductance of the W1 winding of the coupled inductor).
The current in winding 2 (W2) ramps at VC1/L2. In the
case of the 1:1 coupled inductor, L1 = L2. It will be
shown that for the SEPIC converter topology, the DC
voltage across VC1 is equal to VIN. For the case of the
coupled inductor, ∆IW1 = ∆IW2. The switch current (Q1)
is equal to the sum of IW1 and IW2 during the switch on
time. The input current is continuous (and equal to IW1),
minimizing the input ripple current when compared to
other topology choices for the constant current battery
charger.
When Q1 turns off, the path for current flow changes
(Figure 7C).
IW1
SEPIC Converter
ICOUT
tON tOFF
CIN
FIGURE 8:
C1
The power train switching frequency is adapted to the
charge current using the PICmicro® microcontroller
and MCP1630 architecture. This adaptation provides
the best switching frequency for both operating conditions. For example, when the charging current is
500 mA, the SEPIC converter operates at 1 MHz and
minimizes the ripple current seen at the input of the
converter. This will reduce, or even eliminate the need
for additional input filtering, depending on the application. When the converter changes charge current to
trickle charge, the input sensed current used to terminate the duty cycle is very small, making it difficult to
maintain a fixed frequency. Under these conditions,
pulse-skipping can occur. By adapting the switching
frequency to 400 kHz, the sensed current ramp is
increased and the SEPIC converter will maintain a con-
+VBATT
IW1
ICOUT
W2
COUT
IW2
4 NiMH
Batteries
IW1 + IW2
FB
Switching Frequency:
- Fast Charge FSW = 1 MHz
- Trickle Charge FSW = 400 kHz
ICHARGE
L
W1
SEPIC Switch ON.
To design a SEPIC converter or any switching power
converter, it is best to visualize the switching waveforms. For the SEPIC converter, there can be many
modes of operation depending on the continuity of the
inductor current. For this application, a coupled inductor is used, eliminating several of the discontinuous
modes of inductor current operation. Since the
windings are coupled, they are either operating in the
Continuous Inductor Current mode or the
Discontinuous Inductor Current mode.
DS00960A-page 10
IW1 + IW2
+12V Input
VC1/L2
(VOUT) / L2
ISENSE
IW2
VIN/L1
(+VC1 + VOUT - VIN)/L1
IW1
ICOUT
tON tOFF
FIGURE 9:
SEPIC Switch OFF.
With Q1 off, the path for current is now from the input,
through winding 1 (W1) and the coupling cap (C1) to the
output. Another path for current flow exists through
winding 2 (W2) to the output. The sum of W1 and W2
currents flow to the output. During the switch off time,
the sum of W1 and W2 current must also replenish
COUT (output capacitor). Looking at the ICOUT
waveform, it can be seen that the output current of the
SEPIC converter is discontinuous and pulsed. This
may require additional filtering for low output ripple
 2004 Microchip Technology Inc.
AN960
voltage applications. Typically the output ripple is not a
concern for battery charger or constant current lighting
applications.
The transfer function for the SEPIC converter is derived
in a similar fashion as other switching power topologies. Like all switching systems operating in the steady
state, the volt-time product of the magnetic device(s)
must be balanced and the charge-time product on all
capacitor(s) must be balanced (what goes in must
equal what comes out over a switching cycle during
steady-state operation).
Looking closer at the SEPIC topology, the input components (W1, Q1) look like a standard boost converter
input stage. The output components, (W2, D and COUT)
look like an inverted buck-boost converter. The goal is
to determine the volt-time product on the magnetic
devices during the switch on time and set that equal to
the volt-time product on the magnetics during the
switch off time. Using this premise, the transfer function
can be derived.
T SW = 1 ⁄ FSW
TSW = Switching Period
FSW = Switching Frequency
Switch on time = tON
Switch off time = tOFF
Duty Cycle (D) = tON / TSW
Balance the inductor volt-time product in the boost
stage (W1).
Q1 Turned on (+ Slope).
∆I W1 ⁄ tON = V IN ⁄ L W1
Q1 Turned off (- Slope).
∆I W1 ⁄ t OFF
Q1 Turned on (+ Slope).
∆IW2
V C1
------------ = --------t ON
L W2
Q1 Turned off (- Slope).
∆I W2
V OUT
------------ = ------------t OFF
L W2
Inductor slope’s must me equal for volt-time balance.
VC1
V OUT
t ON × ---------- = t OFF × ------------L W2
L W2
Multiply both sides by 1/(tON + tOFF).
VC1 × D = VOUT × ( 1 – D )
Solving for VC1.
1–D
VC1 = VOUT ×  -------------
 D 
Basic Definitions:
-
For the second stage, the inductor slopes must also be
equal.
VC1 + VOUT – VIN
= -------------------------------------------L W1
Inductor slope’s must be equal for volt-time balance.
V IN
VC1 + V OUT – V IN
t ON × ---------- = tOFF × -------------------------------------------L W1
L W1
Set VC1 = VC1 for both the Boost stage and the BuckBoost stage.
1
1–D
VC1 = V IN ×  ------------- – V OUT = VOUT ×  -------------
 1 – D
 D 
Solving for VOUT/VIN.
V OUT
D
------------- =  -------------
 1 – D
VIN
Looking back, if D/(1-D) X VIN is substituted for VOUT, it
is shown that VC1 = VIN. This is true if C1 is sufficiency
large enough that the ripple voltage on C1 is low.
Now that the duty cycle is known as a VOUT/VIN
relationship, the duty cycle can be calculated for any
input output condition. Remember, this transfer function is dependent upon the fact that inductor current is
continuous or never reached zero. If it does reach zero,
this transfer function is no longer true and there is
another state added to the operation.
Multiply both sides by 1/(tON + tOFF).
VIN × D = ( V C1 + VOUT – VIN ) × ( 1 – D )
Solve for VC1.
1
V C1 = VIN ×  ------------- – V OUT
 1 – D
 2004 Microchip Technology Inc.
DS00960A-page 11
AN960
Inductor Winding Current Calculation
The first step to calculating the inductor winding current
is to know the maximum output power. For this constant current battery charger application, the output
power is simply the maximum output voltage times the
charge current.
P OUT = V OUT × I CHARGE
Maximum output voltage is equal to 6.4V (4 batteries @
1.5V + VSENSE + margin).
POUT =6.4V x 500 mA or 3.2 Watts.
Since energy is conserved, the input power is equal to
the output power (assuming 100% efficiency). An
efficiency estimate can be used to closer approximate
the input current.
PIN = P OUT ⁄ ( Efficiency )
PIN = 3.2 Watts / 85%; 85% used as a typical
efficiency estimate.
PIN = 3.77 Watts
I IN ( AVG ) ) = P IN ⁄ V IN
IINAVG = 3.77 Watts/12V (Nominal)
IINAVG = 314 mA. (Typical average input
current)
The peak-to-peak W1 inductor current ripple calculation was shown earlier. Given the derived transfer function and the maximum voltage on the output of the
converter to be 6.4V, the switch on time is estimated.
Switch On Time
V OUT ⁄ ( V OUT + V IN )
t ON = ---------------------------------------------------FSW
For the 12V input and 6.4V output case, the on time of
the switch is estimated to be approximately 348 ns.
(1 MHz switching frequency).
The input peak to peak ripple current can be calculated.
LW1 = LW2 = 20 µH
Input Peak-to-Peak Ripple Current (W1)
∆IL(W1)
=
(12V / 20 µH) x tON = 208 mA
IL(W1)PK
=
IINAVG +1/2 x ∆IL(W1)
IL(W1)PK
=
418 mA for winding 1 (W1)
IL(W1)MIN
=
IINAVG -1/2 x ∆IL(W1)
IL(W1)MIN
=
210 mA for winding 1 (W1)
DS00960A-page 12
W2 Peak-to-Peak Ripple Current
∆IL(W2) = (12V / 20 µH) x tON = 208 mA
IL(W2)PK = IOUTAVG +1/2 x ∆IL(W2)
IL(W2)PK = 604 mA for winding 2 (W2)
IL(W2)MIN = IOUTAVG -1/2 x ∆IL(W2)
IL(W2)MIN = 396 mA for winding 1 (W2)
Note: In the case of VIN = VOUT, the current in
W1 = W2 (ripple and average).
The coupled inductor winding currents calculated
above are used to determine the size of the inductor
necessary. High switching frequency has several
advantages, smaller ripple current, lower peak and
RMS current and lower volt-time product on the
inductor core. This leads to a small, low-cost solution.
SEPIC Switch Current and Voltage Calculations
The average input current is equal to the input power
divided by the input voltage.
GIVEN:
The ripple current in winding (W2) is calculated in a
similar fashion. The main difference is that the average
current in W2 is equal to IOUT or 500 mA in this
application.
The switch current (IQ1) is equal to the combination of
the winding currents during the switch on time. When
the switch is turned on, it conducts the current in W1
and W2.
ISW = IW1 + IW2 = 814 mA (Average)
ISWPK = 418 mA + 604 mA = 1.12 A
The minimum switch current is equal to:
ISWMIN = 210 mA + 396 mA = 606 mA
RMS of a Trapezoidal waveform
2
I SWRMS =
2
 IA + IA × IB + IB
D ×  ---------------------------------------
3


IA = 606 mA = Minimum,
IB = 1.12 A = Maximum
The RMS value of the switch current is approximately
516 mA.
The peak switch voltage is equal to VIN + VOUT for the
SEPIC converter. Any leakage inductance voltage
spike is clamped through the output diode by the output
capacitor. A switch voltage rating for this application
should be a minimum of VIN(MAX) + VOUT(MAX).
VSW = 15V +6.4V
VSW = 21.4V
A 30V, 80 milli-ohm, logic-level switch is selected.
MOSFET switching losses should also be considered
when selecting the MOSFET switch. Low on resistance
switches tend to have high capacitance and will switch
slower, increasing switching losses. The lowest RDSON
MOSFET is not necessarily the best choice. When
using the SOT23 package for a 30V MOSFET, there
are many choices available.
 2004 Microchip Technology Inc.
AN960
SEPIC Diode Voltage and Current Calculations
A schottky diode is recommended for low-voltage
applications. For battery charger applications, the
SEPIC diode will block current flow from the battery
back to the input. The reverse leakage current of the
selected schottky diode can be a critical parameter, if
low battery drain is desired. Low schottky diode forward
drop is also a key parameter; the low drop improves
converter efficiency.
The maximum reverse voltage across the SEPIC diode
occurs during the switch on time. The cathode of the
schottky diode is connected to VOUT, the anode of the
schottky diode is connected to the SEPIC coupling
capacitor. The voltage across the coupling capacitor
voltage is equal to VIN; the voltage across the diode is
equal to VOUT - (-VIN) or VOUT + VIN.
The peak SEPIC diode current occurs when the switch
is turned off. The peak diode current is equal to the
peak current in W2, plus the peak current in W1 or
1.12A. The average diode current is equal to the output
current (IOUT), typical of all topologies with a series
diode in the path of the output.
SEPIC Coupling Capacitor (C1) RMS Current
Calculations and Voltage Rating
The RMS current in the SEPIC coupling capacitor is
mainly dependant upon output power with some
influence by inductor ripple current. As output power
increases, the capacitor ripple current will increase as
well. As shown in Figure 7 (during the switch on time),
the current in winding 2 (output current) is flowing
through the coupling capacitor C1. During the switch off
time, the C1 current is equal to the current in winding
number 1 (W1). As previously discussed, the W1
current is equal to the average input current. Therefore,
the worst case or maximum RMS current in the
coupling capacitor will occur at maximum output power
and minimum input voltage. To estimate size for the
coupling capacitor, the capacitor derivative equation
can be used.
dV
I C = C × ------dt
In this example, there is an average of 500 mA flowing
through the capacitor during the off time of the switch.
The switch off time is approximately 65% or 0.65 x 1 µs
or 650 ns. For a 5% capacitor voltage change at 8V,
minimum input (400 mV), the capacitance necessary
for C1 is equal to (500 mA)/(400 mV/650 ns) or 677 nF.
For this application, a standard 1 µF X7R 25V ceramic
capacitor was used.
+ IW2
IW2
IW1
0
IC1
- IW1
Note: Area above 0 equals the area below zero.
FIGURE 10:
C1 Ripple Current.
As shown in Figure 10, the coupling capacitor ripple
current is largely dependent upon output power and
input voltage. As the input voltage decreases, the current in W1 increases. During the switch on time, the
current flowing in W2 is equal to the current flowing in
C1. When the switch turns off, the current quickly
changes magnitude and direction so that the current
flowing in C1 is equal to the current in W1, magnitude
and direction.
As an approximation, the RMS current in C1:
IC1 ( RMS ) = IOUT × VOUT ⁄ VIN
For worst-case situations, the RMS current in the C1
coupling capacitor is equal to 500 mA x (6.4V / 8V)1/2
or 447 mA. The current rating for small multi-layer
ceramic capacitors is typically much higher than
500 mA. For higher power applications, it may be
necessary to use multiple capacitors in parallel to keep
the RMS current within ratings.
The rate of change of voltage across the capacitor is
related to the amount of current through the capacitor
and the size or energy storage capability of the
capacitor.
For the SEPIC converter coupling capacitor, the voltage is approximated to be a DC value when deriving
the duty cycle. The ripple voltage should be no more
than a few percent of the voltage across the capacitor
or the input voltage. In this example, the minimum input
voltage and C1 DC voltage is 8V, so there should be no
more than 5% or 40 mV of ripple on the coupling
capacitor.
 2004 Microchip Technology Inc.
DS00960A-page 13
AN960
Features
This section describes how several of the NiMH battery
charger board features were implemented using the
combination of the MCP1630 and the PIC16LF818.
INPUT POWER AND BIAS
The input power to the NiMH charger is a single +12V
nominal input. For normal operation, this input source
can vary from +8V to +15V.
Any devices powered when the +12V nominal input is
removed must have low quiescent current. When the
+12V nominal source is provided, a +5V bias is generated using a standard high voltage linear regulator
(U1). There is nothing special about U1, it can be a lowcost bipolar or CMOS. If the +12V input is removed, the
input source to this regulator is removed. This +5V output is used to power components that are only necessary for charging. In this diagram, the +5V powers U2
(MCP1630 high-speed PWM) and U4 Overvoltage
(OV) comparator. The +5V rail also is used to provide
bias power to U5, U6 and U7. U5 is a low IQ low dropout
CMOS LDO (MCP1700). U6 is a PICmicro MCU and
U7 is a dual amplifier (MCP6042) used to condition the
A/D input signals. All of these devices are powered
even when the input source is removed by the battery.
Low quiescent current operation is necessary to minimize the discharge of the battery during power outage.
This architecture allows the partitioning of the high
speed charging circuit bias and the low IQ battery monitoring and fuel gauge bias. U3 is used as a hardware
disconnect to prevent deeply discharging the NiMH
batteries. If the battery voltage ever dips below the
TC54 threshold (2.9V) the output of the TC54 disconnects the battery from the load. Normal charging will
resume when the input power is applied. The total IQ
drain on the battery was measured at 29 µA for this
architecture.
Input Power Architecture
Total I Q = 29 µA
+VBATT
+12V Input
10 µA
+VBATT_SENSE
+5V Bias
U1
U2
U3
Linear
Reg
1 µA
VIN
OV
U4
OSC IN
U5
MCP1700 VDD
+VBATT
1 µA
TC54
–
+
MCP1630
PWM
15 µA
VDD
VDD
A/D
A/D
+VBATT_SENSE
1 µA
+
–
PIC16F818
U6
1 µA
–
+
1/2 MCP6042
U7
Fuel Gauge
1/2 MCP6042
U7
FIGURE 11:
DS00960A-page 14
Input Power Diagram.
 2004 Microchip Technology Inc.
AN960
CHARGING PROFILE
Battery Charger Architecture
+VBATT
+12V Input
U2
U4
+
VREF
MCP
1630
–
Comparator
+VBATT_SENSE
VBATT
Disconnect
Pulse ON and
OFF (50% = Trickle)
U6
VREF = 625 mV
16 kΩ
1 kΩ
61.9 kΩ
22 pF
OSC
Temp Sensor Bias
TC1047A U8
107 kΩ
0.39Ω
61.9 kΩ
+5V Bias
PIC16F818
Battery Temperature Sensor
Trickle Charge
L = Fast Charge
High-Impedance = Top Off Charge
FIGURE 12:
Charging Diagram.
The MCP1630 is used to generate a cycle-by-cycle
1 MHz pulse width that is used to regulate the battery
current. The MCP1630 receives the fundamental
switching frequency from the microcontroller. This
signal not only sets the SEPIC converter switching
frequency, it also sets the maximum duty cycle for the
converter. The SEPIC switching frequency is 1 MHz
during fast charge and 400 kHz for slow charge. Before
connecting to the MCP1630, this clock signal is connected to the input of a general purpose comparator
(U4) (faster than 1 µs). If the non-inverting input of U4
is high (indicating an overvoltage condition on the
charger output (open battery)), the oscillator signal
generated by the PICmicro microcontroller does not get
to the MCP1630, providing fast OV protection. If a bat-
 2004 Microchip Technology Inc.
tery is disconnected or open, the converter output voltage will rise to the maximum OV setting. The MCP1630
will act like a hysteretic ripple regulator until the microcontroller shuts the oscillator off and automatically
attempts to restart the converter. If the OV condition
persists (9 attempts), the microcontroller declares an
OV and provides a slow blinking red LED as indication.
By using the general purpose I/O output of the microcontroller, the charging current can be controlled using
the firmware. A single I/O can adjust the charge current
from a typical 500 mA fast charge to a typical trickle
charge current of 50 mA. To generate a lower top-off
charge current, the microcontroller can turn the 50 mA
trickle charge on and off at a 50% duty cycle to provide
DS00960A-page 15
AN960
an average charge current of 25 mA. In addition to controlling the magnitude of the charge current, the
amount of time can be set and adjusted using the
microcontroller firmware. In the case of the NiMH
charger, the charge cycle always begins with a 2
minute, 50 mA trickle charge. The fast charge begins
(and is terminated) using any one of three conditions.
A negative change in the battery voltage over time, a
positive change in battery temperature over time or by
timing out a maximum fast charge timer. After fast
charge, the NiMH charger will trickle charge for a fixed
amount of time and finish with a top-off charge of
25 mA for a fixed amount of time. Once the charge is
removed, the battery voltage will begin to drop as a
result of self-discharge. Once the battery voltage
reaches a predetermined voltage, the charger will automatically initiate a new charge cycle. The typical battery charge cycle is shown below.
NiMH Charge Cycle @ 0.5C Rate
Battery Voltage
Temperature
Output Current
6.5
600
6
500
5.5
400
5
300
4.5
200
4
100
Current / Temperature
(mA / °C)
Voltage (V)
Output Voltage
Pulsed Charge
25 mA Average
3.5
0
25
50
75
100
125
150
175
200
225
0
250
Time (Minutes)
FIGURE 13:
DS00960A-page 16
4-Cell Charge Profile.
 2004 Microchip Technology Inc.
AN960
CHARGER EFFICIENCY
By using the MCP1630, the battery charger switching
frequency was set to 1 MHz. In addition to small size,
high converter efficiency is desired to minimize power
dissipation during the charging process. The typical
charger efficiency with +12V nominal applied is
approximately 85% at maximum output power (6.4V).
86.0%
ICharge = 500 mA
VBATT = 6.4V
Efficiency (%)
85.0%
84.0%
VBATT = 5.7V
83.0%
82.0%
81.0%
VBATT = 3.8V
80.0%
79.0%
8
FIGURE 14:
9
10
11
12
VIN (Volts)
13
14
15
4-Cell NiMH (Typical) Charger Efficiency.
CONCLUSION
For applications that require intelligent power management solutions like battery chargers, the combination of
a microcontroller and MCP1630 high-speed PWM is
very powerful. It brings the programmability benefits of
the microcontroller and adds the performance of a
high-speed analog PWM. The analog PWM will
respond to changes in input voltage and output current
very quickly. No code or execution time is necessary to
regulate or protect the circuit. The microcontroller is
used for programmability, establishing charge profile
conditions and monitoring the circuit for fault conditions
and taking the appropriate action, in the event of a
specific fault.
 2004 Microchip Technology Inc.
DS00960A-page 17
AN960
NOTES:
DS00960A-page 18
 2004 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED,
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© 2004, Microchip Technology Incorporated, Printed in the
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DS00960A-page 19
 2004 Microchip Technology Inc.
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DS00960A-page 20
 2004 Microchip Technology Inc.