LT3471 Dual 1.3A, 1.2MHz Boost/Inverter in 3mm × 3mm DFN DESCRIPTION FEATURES n n n n n n n n n n n The LT®3471 dual switching regulator combines two 42V, 1.3A switches with error amplifiers that can sense to ground providing boost and inverting capability. The low VCESAT bipolar switches enable the device to deliver high current outputs in a small footprint. The LT3471 switches at 1.2MHz, allowing the use of tiny, low cost and low profile inductors and capacitors. High inrush current at start-up is eliminated using the programmable soft-start function, where an external RC sets the current ramp rate. A constant frequency current mode PWM architecture results in low, predictable output noise that is easy to filter. 1.2MHz Switching Frequency Low VCESAT Switches: 330mV at 1.3A High Output Voltage: Up to 40V Wide Input Range: 2.4V to 16V Inverting Capability 5V at 630mA from 3.3V Input 12V at 320mA from 5V Input –12V at 200mA from 5V Input Uses Tiny Surface Mount Components Low Shutdown Current: < 1μA Low Profile (0.75mm) 10-Lead 3mm × 3mm DFN Package The LT3471 switches are rated at 42V, making the device ideal for boost converters up to ±40V as well as SEPIC and flyback designs. Each channel can generate 5V at up to 630mA from a 3.3V supply, or 5V at 510mA from four alkaline cells in a SEPIC design. The device can be configured as two boosts, a boost and inverter or two inverters. APPLICATIONS n n n n n n n n Organic LED Power Supply Digital Cameras White LED Power Supply Cellular Phones Medical Diagnostic Equipment Local ±5V or ±12V Supply TFT-LCD Bias Supply xDSL Power Supply The LT3471 is available in a low profile (0.75mm) 10-lead 3mm × 3mm DFN package. L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION OLED Driver 2.2μH VIN 3.3V 90.9k 4.7k SHDN/SS1 4.7μF SW1 15k VREF 0.1μF LT3471 10μF CONTROL 2 15k FB2N 4.7k 0.33μF SHDN/SS2 GND VOUT1 = –7V 75 70 65 55 75pF 105k 3471 TA01 10μH 80 60 FB2P SW2 1μF VOUT1 = 7V 85 FB1P VIN 90 FB1N 0.33μF VIN 95 EFFICIENCY (%) CONTROL 1 OLED Driver Efficiency VOUT1 7V 350mA 15μH VIN 10μF VOUT2 –7V 250mA 50 0 100 200 300 400 IOUT (mA) 3471 TA01b 3471fb 1 LT3471 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) VIN Voltage ................................................................16V SW1, SW2 Voltage ..................................... –0.4V to 42V FB1N, FB1P, FB2N, FB2P Voltage......... 12V or VIN – 1.5V SHDN/SS1, SHDN/SS2 Voltage ............................... 16V VREF Voltage.............................................................1.5V Maximum Junction Temperature ........................ 125°C Operating Temperature Range (Note 2) ...– 40°C to 85°C Storage Temperature Range...................– 65°C to 125°C TOP VIEW 10 SW1 FB1N 1 FB1P 2 VREF 3 FB2P 4 7 SHDN/SS2 FB2N 5 6 SW2 9 SHDN/SS1 11 8 VIN DD PACKAGE 10-LEAD (3mm × 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 43°C/ W, θJC = 3°C/W EXPOSED PAD (PIN 11) IS GND MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LT3471EDD#PBF LT3471EDD#TRPBF LBHM 10-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C LEAD BASED FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LT3471EDD LT3471EDD#TR LBHM 10-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ This product is only offered in trays. For more information go to: http://www.linear.com/packaging/ ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = VSHDN = 3V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX 2.1 2.4 V 0.991 0.987 1.000 1.009 1.013 V V 1 1.4 Minimum Operating Voltage Reference Voltage l UNITS Reference Voltage Current Limit (Note 3) Reference Voltage Load Regulation 0mA ≤ IREF ≤ 100μA (Note 3) 0.1 0.2 %/100μA Reference Voltage Line Regulation 2.6V ≤ VIN ≤ 16V 0.03 0.08 %/V Error Amplifier Offset Transition from Not Switching to Switching, VFBP = VFBN = 1V ±2 ±3 mV l mA FB Pin Bias Current V FB = 1V (Note 3) 60 100 nA Quiescent Current VSHDN = 1.8V, Not Switching 2.5 4 mA Quiescent Current in Shutdown VSHDN = 0.3V, VIN = 3V 0.01 1 μA 1 1.2 1.4 90 86 94 % % 15 % Switching Frequency Maximum Duty Cycle l Minimum Duty Cycle Switch Current Limit At Minimum Duty Cycle At Maximum Duty Cycle (Note 4) Switch VCESAT ISW = 0.5A (Note 5) Switch Leakage Current VSW = 5V SHDN/SS Input Voltage High 1.5 0.9 1.8 MHz 2.05 1.45 2.6 2.0 A A 150 250 mV 0.01 1 μA V 3471fb 2 LT3471 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = VSHDN = 3V unless otherwise noted. PARAMETER CONDITIONS SHDN Input Voltage Low Quiescent Current ≤ 1μA MIN SHDN Pin Bias Current VSHDN = 3V, VIN = 4V VSHDN = 0V TYP 22 0 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3471E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating MAX UNITS 0.3 V 36 0.1 μA μA temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Current flows out of the pin. Note 4: See Typical Performance Characteristics for guaranteed current limit vs duty cycle. Note 5: VCESAT is 100% tested at wafer level only. TYPICAL PERFORMANCE CHARACTERISTICS Quiescent Current vs Temperature 1.010 2.4 1.005 2.2 VREF VOLTAGE 100mV/DIV VREF (V) 1.000 2.0 0.995 1.8 1.6 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 VREF CURRENT 200μA/DIV 0.990 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 3471 G01 3471 G02 SHDN/SS Current vs SHDN/SS Voltage Switch Saturation Voltage vs Switch Current Current Limit vs Duty Cycle 2.2 SHDN/SS VOLTAGE 1V/DIV 3471 G04 CURRENT LIMIT (A) 700 TYPICAL 1.8 VIN > VSHDN/SS 800 TA = 25°C 2.0 VIN = 3.3V SHDN/SS CURRENT 20μV/DIV 3471 G03 125 600 1.6 90°C GUARANTEED 1.4 VCESAT (mV) QUIESCENT CURRENT (mA) VREF Voltage vs VREF Current VREF Voltage vs Temperature 2.6 1.2 1.0 0.8 0.6 25°C 400 300 200 0.4 100 0.2 0 500 0 0 20 60 40 DUTY CYCLE (%) 80 100 3471 G05 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 SW CURRENT (A) 3471 G06 3471fb 3 LT3471 TYPICAL PERFORMANCE CHARACTERISTICS Peak Switch Current vs SHDN/SS Voltage 1.50 2.0 1.45 1.8 1.40 1.6 SWITCH CURRENT (A) FREQUENCY (MHz) Oscillator Frequency vs Temperature 1.35 1.30 1.25 1.20 1.15 125 VOUT2 5V/DIV CONTROL 1 AND 2 5V/DIV 0.8 0.6 0.2 100 VOUT1 2V/DIV 1.0 1.05 50 25 0 75 TEMPERATURE (°C) ISUPPLY 1A/DIV 1.2 0.4 –25 TA = 25°C 1.4 1.10 1.00 –50 Start-Up Waveform (Figure 2 Circuit) 0 0.5ms/DIV 0 3471 G09 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2.0 VSHDN/SS (V) 3471 G07 3471 G08 PIN FUNCTIONS FB1N (Pin 1): Negative Feedback Pin for Switcher 1. Connect resistive divider tap here. Minimize trace area at FB1N. Set VOUT = VFB1P(1 + R1/R2), or connect to ground for inverting topologies. FB1P (Pin 2): Positive Feedback Pin for Switcher 1. Connect either to VREF or a divided down version of VREF, or connect to a resistive divider tap for inverting topologies. and minimize the metal trace area connected to this pin to minimize EMI. SHDN/SS2 (Pin 7): Shutdown and Soft-Start Pin. Tie to 1.8V or more to enable device. Ground to shut down. Softstart function is provided when the voltage at this pin is ramped slowly to 1.8V with an external RC circuit. VIN (Pin 8): Input Supply. Must be locally bypassed. VREF (Pin 3): 1.00V Reference Pin. Can supply up to 1mA of current. Do not pull this pin high. Must be locally bypassed with no less than 0.01μF and no more than 1μF. A 0.1μF ceramic capacitor is recommended. Use this pin as the positive feedback reference or connect a resistor divider here for a smaller reference voltage. SHDN/SS1 (Pin 9): Same as SHDN/SS2 but for Switcher 1. Note: taking either SHDN/SS pin high will enable the part. Each switcher is individually enabled with its respective SHDN/SS pin. FB2P (Pin 4): Same as FB1P but for Switcher 2. Exposed Pad (Pin 11): Ground. Connect directly to local ground plane. This ground plane also serves as a heat sink for optimal thermal performance. FB2N (Pin 5): Same as FB1N but for Switcher 2. SW1 (Pin 10): Same as SW2 but for Switcher 1. SW2 (Pin 6): Switch Pin for Switcher 2 (Collector of internal NPN power switch). Connect inductor/diode here 3471fb 4 LT3471 BLOCK DIAGRAM 2 FB1P 10 SW1 + – A1 1 FB1N – RC + CC 8 VIN 1.00V REFERENCE VREF DRIVER A2 R S Q1 Q + 3 ∑ 0.01Ω – 9 4 SHDN/SS1 FB2P RAMP GENERATOR LEVEL SHIFTER 6 SW2 + – A3 5 FB2N – RC + CC 7 SHDN/SS2 11 GND LEVEL SHIFTER DRIVER A4 R S Q2 Q + ∑ 0.01Ω – RAMP GENERATOR GND 1.2MHz OSCILLATOR 3471 F01 Figure 1. Block Diagram OPERATION The LT3471 uses a constant frequency, current mode control scheme to provide excellent line and load regulation. Refer to the Block Diagram. At the start of each oscillator cycle, the SR latch is set, which turns on the power switch, Q1 (Q2). A voltage proportional to the switch current is added to a stabilizing ramp and the resulting sum is fed into the positive terminal of the PWM comparator A2 (A4). When this voltage exceeds the level at the negative input of A2 (A4), the SR latch is reset, turning off the power switch Q1 (Q2). The level at the negative input of A2 (A4) is set by the error amplifier A1 (A3) and is simply an amplified version of the difference between the negative feedback voltage and the positive feedback voltage, usually tied to the reference voltage VREG. In this manner, the error amplifier sets the correct peak current level to keep the output in regulation. If the error amplifier’s output increases, more current is delivered to the output. Similarly, if the error decreases, less current is delivered. Each switcher functions independently but they share the same oscillator and thus the switchers are always in phase. Enabling the part is done by taking either SHDN/SS pin above 1.8V. Disabling the part is done by grounding both SHDN/SS pins. The soft-start feature of the LT3471 allows for clean start-up conditions by limiting the amount of voltage rise at the output of comparator A1 and A2, which in turn limits the peak switching current. The soft-start feature for each switcher is enabled by slowly ramping that switcher’s SHDN/SS pin, using an RC network, for example. Typical resistor and capacitor values are 0.33μF and 4.7k, allowing for a start-up time on the order of milliseconds. The LT3471 has a current limit circuit not shown in the Block Diagram. The switch current is constantly monitored and not allowed to exceed the maximum switch current (typically 1.6A). If the switch 3471fb 5 LT3471 OPERATION current reaches this value, the SR latch is reset regardless of the state of the comparator A2 (A4). Also not shown in the Block Diagram is the thermal shutdown circuit. If the temperature of the part exceeds approximately 160°C, both latches are reset regardless of the state of comparators A2 and A4. The current limit and thermal shutdown circuits protect the power switch as well as the external components connected to the LT3471. APPLICATIONS INFORMATION Duty Cycle The typical maximum duty cycle of the LT3471 is 94%. The duty cycle for a given application is given by: DC = | VOUT |+| VD | – | VIN | | VOUT |+| VD | – | VCESAT | Where VD is the diode forward voltage drop and VCESAT is in the worst case 330mV (at 1.3A) The LT3471 can be used at higher duty cycles, but it must be operated in the discontinuous conduction mode so that the actual duty cycle is reduced. Setting Output Voltage Setting the output voltage depends on the topology used. For normal noninverting boost regulator topologies: R1 VOUT = VFBP 1+ R2 where VFBN is connected between R1 and R2 (see the Typical Applications section for examples). Select values of R1 and R2 according to the following equation: V R1= R2 OUT – 1 VREF A good value for R2 is 15k which sets the current in the resistor divider chain to 1.00V/15k = 67μA. VFBP is usually just tied to VREF = 1.00V, but VFBP can also be tied to a divided down version of VREF or some other voltage as long as the absolute maximum ratings for the feedback pins are not exceeded (see Absolute Maximum Ratings). For inverting topologies, VFBN is tied to ground and VFBP is connected between R1 and R2. R2 is between VFBP and VREF and R1 is between VFBP and VOUT (see the Applications section for examples). In this case: R1 VOUT = VREF R2 Select values of R1 and R2 according to the following equation: V R1=R2 OUT VREF A good value for R2 is 15k, which sets the current in the resistor divider chain to 1.00V/15k = 67μA. Switching Frequency and Inductor Selection The LT3471 switches at 1.2 MHz, allowing for small valued inductors to be used. 4.7μH or 10μH will usually suffice. Choose an inductor that can handle at least 1.4A without saturating, and ensure that the inductor has a low DCR (copper-wire resistance) to minimize I2R power losses. Note that in some applications, the current handling requirements of the inductor can be lower, such as in the SEPIC topology where each inductor only carries one half of the total switch current. For better efficiency, use similar valued inductors with a larger volume. Many different sizes and shapes are available from various manufacturers. Choose a core material that has low losses at 1.2 MHz, such as ferrite core. Table 1. Inductor Manufacturers Sumida (847) 956-0666 www.sumida.com TDK (847) 803-6100 www.tdk.com Murata (714) 852-2001 www.murata.com 3471fb 6 LT3471 APPLICATIONS INFORMATION Soft-Start and Shutdown Features CAPACITOR SELECTION To shut down the part, ground both SHDN/SS pins. To shut down one switcher but not the other one, ground that switcher’s SHDN/SS pin. The soft-start feature provides a way to limit the inrush current drawn from the supply upon start-up. To use the soft-start feature for either switcher, slowly ramp up that switcher’s SHDN/SS pin. The rate of voltage rise at the output of the switcher’s comparator (A1 or A3 for switcher 1 or switcher 2 respectively) tracks the rate of voltage rise at the SHDN/SS pin once the SHDN/SS pin has reached about 1.1V. The soft-start function will go away once the voltage at the SHDN/SS pin exceeds 1.8V. See the Peak Switch Current vs SHDN/SS Voltage graph in the Typical Performance Characteristics section. The rate of voltage rise at the SHDN/SS pin can easily be controlled with a simple RC network connected between the control signal and the SHDN/SS pin. Typical values for the RC network are 4.7kΩ and 0.33μF, giving start-up times on the order of milliseconds. This RC time constant can be adjusted to give different start-up times. If different values of resistance are to be used, keep in mind the SHDN/SS Current vs SHDN/SS voltage graph along with the Peak Switch Current vs SHDN/SS Voltage graph, both found in the Typical Performance Characteristics section. The impedance looking into the SHDN/SS pin depends on whether the SHDN/SS is above or below VIN. Normally SHDN/SS will not be driven above VIN, and thus the impedance looks like 100kΩ in series with a diode. If the voltage of the SHDN/SS pin is above VIN, the impedance looks more like 50kΩ in series with a diode. This 100kΩ or 50kΩ impedance can have a slight effect on the start-up time if you choose the R in the RC soft-start network too large. Another consideration is selecting the soft-start time so that the soft-start feature is dominated by the RC network and not the capacitor on VREF. (See VREF voltage reference section of the Applications Information for details.) Low ESR (equivalent series resistance) capacitors should be used at the output to minimize the output ripple voltage. Multi-layer ceramic capacitors are an excellent choice, as they have extremely low ESR and are available in very small packages. X5R dielectrics are preferred, followed by X7R, as these materials retain the capacitance over wide voltage and temperature ranges. A 4.7μF to 15μF output capacitor is sufficient for most applications, but systems with very low output currents may need only a 1μF or 2.2μF output capacitor. Solid tantalum or OS-CON capacitors can be used, but they will occupy more board area than a ceramic and will have a higher ESR. Always use a capacitor with a sufficient voltage rating. The soft-start feature is of particular importance in applications where the switch will see voltage levels of 30V or higher. In these applications, the simultaneous presence of high current and voltage during startup may cause an overstress condition to the switch. Therefore, depending on input and output voltage conditions, higher RC time constant values may be necessary to improve the ruggedness of the design. Ceramic capacitors also make a good choice for the input decoupling capacitor, which should be placed as close as possible to the LT3471. A 4.7μF to 10μF input capacitor is sufficient for most applications. Table 2 shows a list of several ceramic capacitor manufacturers. Consult the manufacturers for detailed information on their entire selection of ceramic parts. Table 2. Ceramic Capacitor Manufacturers Taiyo Yuden (408) 573-4150 www.t-yuden.com AVX (803) 448-9411 www.avxcorp.com Murata (714) 852-2001 www.murata.com The decision to use either low ESR (ceramic) capacitors or the higher ESR (tantalum or OS-CON) capacitors can affect the stability of the overall system. The ESR of any capacitor, along with the capacitance itself, contributes a zero to the system. For the tantalum and OS-CON capacitors, this zero is located at a lower frequency due to the higher value of the ESR, while the zero of a ceramic capacitor is at a much higher frequency and can generally be ignored. A phase lead zero can be intentionally introduced by placing a capacitor (CPL) in parallel with the resistor (R3) between VOUT and VFB as shown in Figure 2. The frequency of the zero is determined by the following equation. ƒZ = 1 2π • R3 • CPL 3471fb 7 LT3471 APPLICATIONS INFORMATION L1 2.2μH D1 VIN RSS1 4.7k CONTROL 1 1.8V 0V 10 9 SHDN/SS1 SW1 FB1N CSS1 0.33μF VIN 2.6V TO 4.2V Li-Ion FB1P 8 10μF RSS2 4.7k CONTROL 2 1.8V 0V CSS2 0.33μF VREF VIN 1 2 VOUT1 7V C3 4.7μF C2 0.1μF SHDN/SS2 GND SW2 11 6 FB2P L2 10μH R3 90.9k R4 15k 3 LT3471 FB2N 7 CPL 33pF 5 R2 15k 4 3471 F02 C5 1μF L3 15μH R1 105k C6 75pF VOUT2 –7V VIN C4 10μF D2 C1, C2: X5R OR X7R 6.3V C3, C4: X5R OR X7R 10V C5: XR5 OR X7R 16V CPL: OPTIONAL D1, D2: ON SEMICONDUCTOR MBRM-120 L1: SUMIDA CR43-2R2 L2: SUMIDA CDRH4D18-100 L3: SUMIDA CDRH4D18-150 Supply Current of Figure 2 During Start-Up without Soft-Start RC Network ISUPPLY 0.5A/DIV VIN = 3.3V ISUPPLY 0.5A/DIV VOUT1 2V/DIV VIN > VSHDN/SS 0.1ms/DIV Supply Current of Figure 2 During Start-Up with Soft-Start RC Network VOUT1 2V/DIV VIN = 3.3V VIN > VSHDN/SS 3471 F02b 0.2ms/DIV 3471 F02c Figure 2. Li-Ion OLED Driver 3471fb 8 LT3471 APPLICATIONS INFORMATION By choosing the appropriate values for the resistor and capacitor, the zero frequency can be designed to improve the phase margin of the overall converter. The typical target value for the zero frequency is between 35kHz to 55kHz. Figure 3 shows the transient response of the step-up converter from Figure 2 without the phase lead capacitor CPL. Although adequate for many applications, phase margin is not ideal as evidenced by 2-3 “bumps” in both the output voltage and inductor current. A 33pF capacitor for CPL results in ideal phase margin, which is revealed in Figure 4 as a more damped response and less overshoot. VOUT 200mV/DIV AC COUPLED VIN = 3.3V IL1 0.5A/DIV AC/COUPLED LOAD CURRENT 100mA/DIV AC/COUPLED VIN > VSHDN/SS 50μs/DIV Figure 3. Transient Response of Figure 2’s Step-Up Converter without Phase Lead Capacitor VOUT 200mV/DIV AC COUPLED VIN = 3.3V IL1 0.5A/DIV AC/COUPLED LOAD CURRENT 100mA/DIV AC/COUPLED VIN > VSHDN/SS VREG VOLTAGE REFERENCE Pin 3 of the LT3471 is a bandgap voltage reference that has been divided down to 1.00V and buffered for external use. This pin must be bypassed with at least 0.01μF and no more than 1μF. This will ensure stability as well as reduce the noise on this pin. The buffer has a built-in current limit of at least 1mA (typically 1.4mA). This not only means that you can use this pin as an external reference for supplemental circuitry, but it also means that it is possible to provide a soft-start feature if this pin is used as one of the feedback pins for the error amplifier. Normally the soft-start time will be dominated by the RC time constant discussed in the soft-start and shutdown section. However, because of the finite current limit of the buffer for the VREG pin, it will take some time to charge up the bypass capacitor. During this time, the voltage at the VREG pin will ramp up, and this action provides an alternate means for soft-starting the circuit. If the largest recommended bypass capacitor is used, 1μF, the worst-case (longest) soft-start function that would be provided from the VREF pin is: 1μF • 1.00V =1.0ms 1.0mA Choose the RC network such that the soft-start time is longer than this time, or choose a smaller bypass capacitor for the VREF pin (but always larger than 0.01μF) so that the RC network dominates the soft-starting of the LT3471. The voltage at the VREF pin can also be divided down and used for one of the feedback pins for the error amplifier. This is especially useful in LED driver applications, where the current through the LEDs is set using the voltage reference across a sense resistor in the LED chain. Using a smaller or divided down reference leads to less wasted power in the sense resistor. See the Typical Applications section for an example of LED driving applications. 50μs/DIV Figure 4. Transient Response of Figure 2’s Step-Up Converter with 33pF Phase Lead Capacitor 3471fb 9 LT3471 APPLICATIONS INFORMATION DIODE SELECTION Compensation—Theory A Schottky diode is recommended for use with the LT3471. For high efficiency, a diode with good thermal characteristics at high currents should be used such as the On Semiconductor MBRM120. This is a 20V diode. Where the switch voltage exceeds 20V, use the MBRM140, a 40V diode. These diodes are rated to handle an average forward current of 1.0A. In applications where the average forward current of the diode is less than 0.5A, use the Philips PMEG 2005, 3005, or 4005 (a 20V, 30V or 40V diode, respectively). Like all other current mode switching regulators, the LT3471 needs to be compensated for stable and efficient operation. Two feedback loops are used in the LT3471: a fast current loop which does not require compensation, and a slower voltage loop which does. Standard Bode plot analysis can be used to understand and adjust the voltage feedback loop. LAYOUT HINTS The high speed operation of the LT3471 demands careful attention to board layout. You will not get advertised performance with careless layout. Figure 5 shows the recommended component placement. As with any feedback loop, identifying the gain and phase contribution of the various elements in the loop is critical. Figure 6 shows the key equivalent elements of a boost converter. Because of the fast current control loop, the power stage of the IC, inductor and diode have been replaced by the equivalent transconductance amplifier gmp. gmp acts as a current source where the output current is proportional to the VC voltage. Note that the maximum output current of gmp is finite due to the current limit in the IC. CONTROL 2 CONTROL 1 CSS1 RSS1 GND – CSS2 RSS2 GND gmp GND VOUT + CPL RESR RL C4 C1 VOUT2 VC gma L2 L1 VOUT1 D1 SW1 10 9 L3 VCC • 8 SW2 6 SHDN/SS1 C3 7 SHDN/SS2 C5 RC • FB1P VREF FB2P FB2N 2 3 4 5 R3 R2 R1 – R2 CC: COMPENSATION CAPACITOR COUT: OUTPUT CAPACITOR CPL: PHASE LEAD CAPACITOR gma: TRANSCONDUCTANCE AMPLIFIER INSIDE IC gmp: POWER STAGE TRANSCONDUCTANCE AMPLIFIER RC: COMPENSATION RESISTOR RL: OUTPUT RESISTANCE DEFINED AS VOUT DIVIDED BY ILOAD(MAX) RO: OUTPUT RESISTANCE OF gma R1, R2: FEEDBACK RESISTOR DIVIDER NETWORK RESR: OUTPUT CAPACITOR ESR D2 LT3471 1 COUT 1.00V REFERENCE 3471 F06 PIN 11 GND FB1N RO CC GND GND R4 + Figure 6. Boost Converter Equivalent Model R1 VOUT2 VOUT1 C2 3471 F05 Figure 5. Suggested Layout Showing a Boost on SW1 and an Inverter on SW2. Note the Separate Ground Returns for All High Current Paths (Using a Multilayer Board) 3471fb 10 LT3471 APPLICATIONS INFORMATION From Figure 6, the DC gain, poles and zeroes can be calculated as follows: Output Pole: P1= 2 2 • π • RL • COUT Value Units RL 20 Ω Application Specific COUT 4.7 μF Application Specific RESR 10 mΩ Application Specific RO 0.9 MΩ Not Adjustable CC 90 pF Not Adjustable CPL 33 pF Adjustable RC 55 kΩ Not Adjustable 1 R1 90.9 kΩ Adjustable 2 • π • RESR • COUT R2 15 kΩ Adjustable VOUT 7 V VIN 3.3 V gma 50 μmho Not Adjustable gmp 9.3 mho Not Adjustable L 2.2 μH fS 1.2 MHz Error Amp Zero: Z1= 1 2 • π • RC • CC V 1 DC GAIN: A= REF • gma • RO • gmp • RL • VOUT 2 RHP Zero: Z3= Table 3. Bode Plot Parameters Parameter 1 Error Amp Pole: P2= 2 • π • RO • CC ESR Zero: Z2 = Using the circuit of Figure 2 as an example, Table 3 shows the parameters used to generate the Bode plot shown in Figure 7. VIN 2 • RL 2 • π • VOUT 2 • L High Frequency Pole: P3> fS 3 1 2 • π • R1• CPL 1 Phase Lead Pole: P4 = R1• R2 2 • π • CPL • R1+R2 Comment Application Specific Application Specific Application Specific Not Adjustable Phase Lead Zero: Z4 = 70 0 60 –50 50 –100 GAIN (dB) 40 30 –150 20 –200 10 –250 0 –300 –10 –20 –30 100 PHASE (DEG) The Current Mode zero is a right half plane zero which can be an issue in feedback control design, but is manageable with proper external component selection. From Figure 7, the phase is –115° when the gain reaches 0dB giving a phase margin of 65°. This is more than adequate. The crossover frequency is 50kHz. –350 GAIN PHASE 1k –400 10k 100k FREQUENCY (Hz) 1M 3471 F07 Figure 7. Bode Plot of 3.3V to 7V Application 3471fb 11 LT3471 TYPICAL APPLICATIONS Li-Ion OLED Driver L1 2.2μH D1 VIN CONTROL 1 1.8V 0V VIN 2.6V TO 4.2V Li-Ion CONTROL 2 1.8V 0V RSS1 4.7k 10 9 SHDN/SS1 SW1 FB1N CSS1 0.33μF FB1P 8 C1 10μF RSS2 4.7k CSS2 0.33μF VREF VIN 7 1 2 SHDN/SS2 GND FB2P VCONTROL C2 0.1μF 0V TO 1V R5 20k 5 4 6 L2 15μH VOUT1 7V 500mA WHEN VIN = 4.2V 350mA WHEN VIN = 3.3V 250mA WHEN VIN = 2.6V R2 15k R6 10k SW2 11 C3 4.7μF R4 15k 3 LT3471 FB2N R3 90.9k C6 33pF 3471 TA02 C5 1μF L3 15μH R1 105k VIN C4 10μF D2 C1, C2: X5R OR X7R 6.3V C3, C4: X5R OR X7R 10V C5: XR5 OR X7R 16V C6: OPTIONAL C6 75pF D1, D2: ON SEMICONDUCTOR MBRM-120 L1: SUMIDA CR43-2R2 L2: SUMIDA CDRH4D18-100 L3: SUMIDA CDRH4D18-150 VOUT2 –7V TO –4V –7V WHEN VCONTROL = 0V –4V WHEN VCONTROL = 1 –7V, 300mA WHEN VIN = 4.2V –7V, 250mA WHEN VIN = 3.3V –7V, 200mA WHEN VIN = 2.6V Li-Ion OLED Driver Efficiency 95 VOUT = 7V 90 VIN = 4.2V EFFICIENCY (%) 85 VIN = 3.3V VIN = 2.6V 80 VIN = 4.2V VIN = 3.3V 75 70 VIN = 2.6V 65 60 VOUT = –7V 55 50 0 100 300 200 IOUT (mA) 400 500 3471 TA02b 3471fb 12 LT3471 TYPICAL APPLICATIONS Single Li-Ion Cell to 5V, 12V Boost Converter L1 3.3μH C3 10μF VOUT1 5V 900mA IF VIN = 4.2V 630mA IF VIN = 3.3V 425mA IF VIN = 2.6V C4 10μF VOUT2 12V 300mA IF VIN = 4.2V 210mA IF VIN = 3.3V 145mA IF VIN = 2.6V D1 VIN CONTROL 1 1.8V OV 9 SHDN/SS1 SW1 FB1N CSS1 0.33μF FB1P 8 VIN 2.6V TO 4.2V CONTROL 2 1.8V 0V 10 RSS1 4.7k RSS2 4.7k VREF VIN CSS2 0.33μF 2 C2 0.1μF FB2P SHDN/SS2 GND 11 R1 20k R2 4.99k 3 LT3471 C1 4.7μF 7 C5 100pF 1 FB2N 4 5 SW2 L2 6.8μH 6 3471 TA03 VIN D2 C6 220pF R3 54.9k R4 4.99k C1-C3: X5R OR X7R 6.3V C4: X5R OR X7R 16V D1, D2: ON SEMICONDUCTOR MBRM-120 L1: SUMIDA CR43-3R3 L2: SUMIDA CR43-6R8 3471fb 13 LT3471 TYPICAL APPLICATIONS Li-Ion 20 White LED Driver L1 2.2μH D1 VIN CONTROL 1 1.8V OV 9 SHDN/SS1 SW1 FB1N CSS1 0.33μF FB1P 8 VIN 2.6V TO 4.2V CONTROL 2 1.8V OV C3 0.22μF 10 RSS1 4.7k RSS2 4.7k VREF VIN CSS2 0.33μF 2 3 C2 0.1μF FB2P 7 1 LT3471 C1 4.7μF SHDN/SS2 GND FB2N IOUT1 20mA 4 R1 90.9k 10 WHITE LEDs R2 10k 5 SW2 11 6 3471 TA04 4.99Ω L2 2.2μH VIN D2 C4 0.22μF IOUT2 20mA C1, C2: X5R OR X7R 6.3V C3, C4: X5R OR X7R 50V D1, D2: ON SEMICONDUCTOR MBRM-140 L1, L2: SUMIDA CDRH2D-2R2 10 WHITE LEDs 4.99Ω 3471fb 14 LT3471 TYPICAL APPLICATIONS Li-Ion or 4-Cell Alkaline to 3.3V and 5V SEPIC C3 4.7μF L1 10μH D1 VIN CONTROL 1 1.8V OV 10 RSS1 4.7k 9 SW1 SHDN/SS1 RSS2 4.7k SHDN/SS2 GND R1 34.8k C2 0.1μF LT3471 4 FB2P 7 C4 15μF R2 15k 3 VREF VIN C1 4.7μF CSS2 0.33μF 2 FB1P 8 C7 56pF 1 FB1N CSS1 0.33μF VIN 2.6V TO 6.5V CONTROL 2 1.8V OV L2 10μH VOUT1 3.3V 640mA AT VIN = 6.5V 550mA AT VIN = 5V 470mA AT VIN = 4V 410mA AT VIN = 3.3V 340mA AT VIN = 2.6V 5 FB2N SW2 11 6 3471 TA05 C5 10μF L3 10μH D2 VIN C1, C3, C5: X5R OR X7R 10V C4, C6: X5R OR X7R 6.3V D1, D2: ON SEMICONDUCTOR MBRM-120 L1-L4: MURATA LQH43CN100K032 L4 10μH C6 15μF C8 R3 56pF 60.4k R4 15k VOUT2 5V 500mA AT VIN = 6.5V 420mA AT VIN = 5V 360mA AT VIN = 4V 300mA AT VIN = 3.3V 250mA AT VIN = 2.6V PACKAGE DESCRIPTION DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1698) R = 0.115 TYP 6 0.38 ± 0.10 10 0.675 ±0.05 3.50 ±0.05 1.65 ±0.05 2.15 ±0.05 (2 SIDES) 3.00 ±0.10 (4 SIDES) PACKAGE OUTLINE 1.65 ± 0.10 (2 SIDES) PIN 1 TOP MARK (SEE NOTE 6) (DD) DFN 1103 5 0.200 REF 0.25 ± 0.05 0.50 BSC 2.38 ±0.05 (2 SIDES) 1 0.25 ± 0.05 0.50 BSC 0.75 ±0.05 0.00 – 0.05 2.38 ±0.10 (2 SIDES) BOTTOM VIEW—EXPOSED PAD RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3471fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT3471 TYPICAL APPLICATIONS 5V to ±12V Dual Supply Boost/Inverting Converter L1 10μH D1 VIN 10 CONTROL 1 1.8V OV 4.7k 9 SHDN/SS1 SW1 FB1N FB1P 0.33μF 8 VIN 5V CONTROL 2 1.8V OV VREF VIN 0.33μF SHDN/SS2 GND FB2N 4 5 C7 56pF 6 3471 TA06 • VIN R3 15k SW2 11 L2 10μH C3 4.7μF R2 4.99k 3 C2 0.1μF FB2P 7 2 LT3471 C1 4.7μF 4.7k R1 C6 56pF 54.9k 1 VOUT1 12V 320mA R4 182k • C5 1μF D2 C1, C2: X5R OR X7R 6.3V C3, C4: X5R OR X7R 16V C5: X5R OR X7R 25V D1, D2: ON SEMICONDUCTOR MBRM-120 L3 10μH C4 4.7μF VOUT2 –12V 200mA L1: SUMIDA CR43-10 L2, L3: SUMIDA CLS63-10 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1611 550mA (ISW ), 1.4MHz, High Efficiency Micropower Inverting DC/DC Converter VIN: 1.1V to 10V, VOUT(MAX) = –34V, IQ = 3mA, ISD < 1μA, ThinSOT Package LT1613 550mA (ISW ), 1.4MHz, High Efficiency Step-Up DC/DC Converter VIN: 0.9V to 10V, VOUT(MAX) = 34V, IQ = 3mA, ISD < 1μA, ThinSOT Package LT1614 750mA (ISW ), 600kHz, High Efficiency Micropower Inverting DC/DC Converter VIN: 1V to 12V, VOUT(MAX) = –24V, IQ = 1mA, ISD < 10μA, MS8, S8 Packages LT1615/LT1615-1 300mA/80mA (ISW ), High Efficiency Step-Up DC/DC Converters VIN = 1V to 15V, VOUT(MAX) = 34V, IQ = 20μA, ISD < 1μA, ThinSOT Package LT1617/LT1617-1 350mA/100mA (ISW ), High Efficiency Micropower Inverting DC/DC Converters VIN = 1.2V to 15V, VOUT(MAX) = –34V, IQ = 20μA, ISD < 1μA, ThinSOT Package LT1930/LT1930A 1A (ISW ), 1.2MHz/2.2MHz, High Efficiency Step-Up DC/DC Converters VIN: 2.6V to 16V, VOUT(MAX) = 34V, IQ = 4.2mA/5.5mA, ISD < 1μA, ThinSOT Package LT1931/LT1931A 1A (ISW ), 1.2MHz/2.2MHz High Efficiency Micropower Inverting DC/DC Converters VIN = 2.6V to 16V, VOUT(MAX) = –34V, IQ = 5.8mA, ISD < 1μA, ThinSOT Package LT1943 (Quad) Quad Boost, 2.6A Buck, 2.6A Boost, 0.3A Boost, 0.4A Inverter 1.2MHz TFT DC/DC Converter VIN = 4.5V to 22V, VOUT(MAX) = 40V, IQ = 10μA, ISD < 35μA, TSSOP28E Package LT1945 (Dual) Dual Output, Boost/Inverter, 350mA (ISW ), Constant Off-Time, High Efficiency Step-Up DC/DC Converter VIN = 1.2V to 15V, VOUT(MAX) = ±34V, IQ = 40μA, ISD < 1μA, 10-Lead MS Package LT1946/LT1946A 1.5A (ISW ), 1.2MHz/2.7MHz, High Efficiency Step-Up DC/DC Converters VIN: 2.45V to 16V, VOUT(MAX) = 34V, IQ = 3.2mA, ISD < 1μA, MS8 Package LT3436 3A (ISW ), 1MHz, 34V Step-Up DC/DC Converter VIN: 3V to 25V, VOUT(MAX) = 34V, IQ = 0.9mA, ISD < 6μA, TSSOP16E Package LT3462/LT3462A 300mA (ISW ), 1.2MHz/2.7MHz, High Efficiency Inverting DC/DC Converters with Integrated Schottkys VIN = 2.5V to 16V, VOUT(MAX) = –38V, IQ = 2.9mA, ISD < 1μA, ThinSOT Package LT3463/LT3463A VIN = 2.3V to 15V, VOUT(MAX) = ±40V, IQ = 40μA, ISD < 1μA, Dual Output, Boost/Inverter, 250mA (ISW ), Constant Off-Time, High Efficiency Step-Up DC/DC Converters with Integrated Schottkys DFN Package LT3464 85mA (ISW ), High Efficiency Step-Up DC/DC Converter with Integrated Schottky and PNP Disconnect VIN = 2.3V to 10V, VOUT(MAX) = 34V, IQ = 25μA, ISD < 1μA, ThinSOT Package 3471fb 16 Linear Technology Corporation LT 1008 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2004