LINER LT3471

LT3471
Dual 1.3A, 1.2MHz
Boost/Inverter in
3mm × 3mm DFN
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FEATURES
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DESCRIPTIO
The LT®3471 dual switching regulator combines two 42V,
1.3A switches with error amplifiers that can sense to
ground providing boost and inverting capability. The low
VCESAT bipolar switches enable the device to deliver high
current outputs in a small footprint. The LT3471 switches
at 1.2MHz, allowing the use of tiny, low cost and low
profile inductors and capacitors. High inrush current at
start-up is eliminated using the programmable soft-start
function, where an external RC sets the current ramp rate.
A constant frequency current mode PWM architecture
results in low, predictable output noise that is easy to filter.
1.2MHz Switching Frequency
Low VCESAT Switches: 330mV at 1.3A
High Output Voltage: Up to 40V
Wide Input Range: 2.4V to 16V
Inverting Capability
5V at 630mA from 3.3V Input
12V at 320mA from 5V Input
–12V at 200mA from 5V Input
Uses Tiny Surface Mount Components
Low Shutdown Current: < 1µA
Low Profile (0.75mm) 10-Lead 3mm × 3mm
DFN Package
The LT3471 switches are rated at 42V, making the device
ideal for boost converters up to ±40V as well as SEPIC and
flyback designs. Each channel can generate 5V at up to
630mA from a 3.3V supply, or 5V at 510mA from four
alkaline cells in a SEPIC design. The device can be configured as two boosts, a boost and inverter or two inverters.
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APPLICATIO S
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Organic LED Power Supply
Digital Cameras
White LED Power Supply
Cellular Phones
Medical Diagnostic Equipment
Local ±5V or ±12V Supply
TFT-LCD Bias Supply
xDSL Power Supply
The LT3471 is available in a low profile (0.75mm) 10-lead
3mm × 3mm DFN package.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
OLED Driver
2.2µH
VIN
3.3V
90.9k
4.7µF
SW1
4.7k
SHDN/SS1
90
FB1N
0.33µF
FB1P
VIN
15k
0.1µF
LT3471
10µF
CONTROL 2
15k
FB2N
4.7k
0.33µF
SHDN/SS2
GND
VOUT1 = 7V
85
VREF
VIN
95
EFFICIENCY (%)
CONTROL 1
OLED Driver Efficiency
VOUT1
7V
350mA
80
VOUT1 = –7V
75
70
65
FB2P
60
SW2
75pF
105k
10µH
1µF
15µH
VIN
10µF
3471 TA01
55
VOUT2
–7V
250mA
50
0
100
200
300
400
IOUT (mA)
3471 TA01b
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LT3471
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
VIN Voltage .............................................................. 16V
SW1, SW2 Voltage ....................................– 0.4V to 42V
FB1N, FB1P, FB2N, FB2P Voltage ....... 12V or VIN – 1.5V
SHDN/SS1, SHDN/SS2 Voltage .............................. 16V
VREF Voltage ........................................................... 1.5V
Maximum Junction Temperature ......................... 125°C
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Storage Temperature Range ................. – 65°C to 125°C
ORDER PART
NUMBER
TOP VIEW
10 SW1
FB1N
1
FB1P
2
VREF
3
FB2P
4
7 SHDN/SS2
FB2N
5
6 SW2
9 SHDN/SS1
11
LT3471EDD
8 VIN
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/ W, θJC = 3°C/ W
EXPOSED PAD (PIN 11) IS GND
MUST BE SOLDERED TO PCB
DD PART MARKING
LBHM
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating
temperature range, otherwise specifications are TA = 25°C. VIN = VSHDN = 3V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
2.1
2.4
V
0.991
0.987
1.000
1.009
1.013
V
V
1
1.4
Minimum Operating Voltage
Reference Voltage
●
Reference Voltage Current Limit
(Note 3)
Reference Voltage Load Regulation
0mA ≤ IREF ≤ 100µA (Note 3)
0.1
0.2
%/100µA
Reference Voltage Line Regulation
2.6V ≤ VIN ≤ 16V
0.03
0.08
%/V
Error Amplifier Offset
Transition from Not Switching to Switching, VFBP = VFBN = 1V
±2
±3
mV
FB Pin Bias Current
(Note 3)
60
100
nA
Quiescent Current
VSHDN = 1.8V, Not Switching
2.5
4
mA
Quiescent Current in Shutdown
VSHDN = 0.3V, VIN = 3V
0.01
1
µA
1
1.2
1.4
90
86
94
%
%
15
%
●
Switching Frequency
Maximum Duty Cycle
●
Minimum Duty Cycle
Switch Current Limit
At Minimum Duty Cycle
At Maximum Duty Cycle (Note 4)
Switch VCESAT
ISW = 1.3A (Note 5)
Switch Leakage Current
VSW = 5V
SHDN/SS Input Voltage High
1.5
0.9
mA
2.05
1.45
2.6
2.0
A
A
330
440
mV
0.01
1
µA
1.8
SHDN Input Voltage Low
Quiescent Current ≤ 1µA
SHDN Pin Bias Current
VSHDN = 3V, VIN = 4V
VSHDN = 0V
Note 1: Absolute Maximum Ratings are those values beyond which the life of
a device may be impaired.
Note 2: The LT3471E is guaranteed to meet performance specifications from
0°C to 70°C. Specifications over the –40°C to 85°C operating temperature
range are assured by design, characterization and correlation with statistical
process controls.
MHz
V
22
0
0.3
V
36
0.1
µA
µA
Note 3: Current flows out of the pin.
Note 4: See Typical Performance Characteristics for guaranteed current
limit vs duty cycle.
Note 5: VCESAT is 100% tested at wafer level.
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LT3471
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TYPICAL PERFOR A CE CHARACTERISTICS
Quiescent Current
vs Temperature
2.4
1.005
2.2
VREF (V)
1.000
2.0
VREF
VOLTAGE
100mV/DIV
0.995
1.8
1.6
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
0.990
–50
125
50
25
0
75
TEMPERATURE (°C)
–25
125
100
VREF CURRENT 200µA/DIV
3471 G01
3741 G03
3471 G02
SHDN/SS Current
vs SHDN/SS Voltage
Switch Saturation Voltage
vs Switch Current
Current Limit vs Duty Cycle
2.2
VIN = 3.3V
600
1.6
CURRENT LIMIT (A)
VIN > VSHDN/SS
700
TYPICAL
1.8
SHDN/SS
CURRENT
20µV/DIV
800
TA = 25°C
2.0
90°C
GUARANTEED
1.4
VCESAT (mV)
QUIESCENT CURRENT (mA)
VREF Voltage vs VREF Current
VREF Voltage vs Temperature
1.010
2.6
1.2
1.0
0.8
0.6
500
25°C
400
300
200
0.4
100
0.2
SHDN/SS VOLTAGE 1V/DIV
3741 G04
0
0
0
20
60
40
DUTY CYCLE (%)
80
100
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
SW CURRENT (A)
3471 G05
Peak Switch Current
vs SHDN/SS Voltage
1.50
2.0
1.45
1.8
1.40
1.6
SWITCH CURRENT (A)
FREQUENCY (MHz)
Oscillator Frequency
vs Temperature
1.35
1.30
1.25
1.20
1.15
3471 G07
VOUT1
2V/DIV
VOUT2
5V/DIV
CONTROL 1 AND 2
5V/DIV
0.8
0.6
0.2
125
ISUPPLY
1A/DIV
1.0
0.4
100
TA = 25°C
1.2
1.05
50
25
0
75
TEMPERATURE (°C)
Start-Up Waveform
(Figure 2 Circuit)
1.4
1.10
1.00
–50 –25
3471 G06
0
0.5ms/DIV
0
3471 G09
0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2.0
VSHDN/SS (V)
3471 G08
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LT3471
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FB1N (Pin 1): Negative Feedback Pin for Switcher 1.
Connect resistive divider tap here. Minimize trace area at
FB1N. Set VOUT = VFB1P(1 + R1/R2), or connect to ground
for inverting topologies.
and minimize the metal trace area connected to this pin to
minimize EMI.
SHDN/SS2 (Pin 7): Shutdown and Soft-Start Pin. Tie to
1.8V or more to enable device. Ground to shut down. Softstart function is provided when the voltage at this pin is
ramped slowly to 1.8V with an external RC circuit.
FB1P (Pin 2): Positive Feedback Pin for Switcher 1. Connect either to VREG or a divided down version of VREG, or
connect to a resistive divider tap for inverting topologies.
VIN (Pin 8): Input Supply. Must be locally bypassed.
VREF (Pin 3): 1.00V Reference Pin. Can supply up to 1mA
of current. Do not pull this pin high. Must be locally
bypassed with no less than 0.01µF and no more than 1µF.
A 0.1µF ceramic capacitor is recommended. Use this pin
as the positive feedback reference or connect a resistor
divider here for a smaller reference voltage.
SHDN/SS1 (Pin 9): Same as SHDN/SS2 but for Switcher 1.
Note: taking either SHDN/SS pin high will enable the part.
Each switcher is individually enabled with its respective
SHDN/SS pin.
SW1 (Pin 10): Same as SW2 but for Switcher 1.
Exposed Pad (Pin 11): Ground. Connect directly to local
ground plane. This ground plane also serves as a heat sink
for optimal thermal performance.
FB2P (Pin 4): Same as FB1P but for Switcher 2.
FB2N (Pin 5): Same as FB1N but for Switcher 2.
SW2 (Pin 6): Switch Pin for Switcher 2 (Collector of
internal NPN power switch). Connect inductor/diode here
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BLOCK DIAGRA
2
FB1P
10 SW1
+
–
A1
1
FB1N
–
RC
+
CC
8
VIN
1.00V
REFERENCE
VREF
DRIVER
A2
R
S
Q1
Q
+
0.01Ω
Σ
3
–
9
4
SHDN/SS1
FB2P
RAMP
GENERATOR
LEVEL
SHIFTER
6 SW2
+
–
A3
5
FB2N
–
RC
+
CC
7
SHDN/SS2
11 GND
LEVEL
SHIFTER
DRIVER
A4
R
S
Q2
Q
+
0.01Ω
Σ
–
RAMP
GENERATOR
1.2MHz
OSCILLATOR
GND
3471 F01
Figure 1. Block Diagram
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LT3471
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OPERATIO
The LT3471 uses a constant frequency, current mode
control scheme to provide excellent line and load regulation. Refer to the Block Diagram. At the start of each
oscillator cycle, the SR latch is set, which turns on the
power switch, Q1 (Q2). A voltage proportional to the
switch current is added to a stabilizing ramp and the
resulting sum is fed into the positive terminal of the PWM
comparator A2 (A4). When this voltage exceeds the level
at the negative input of A2 (A4), the SR latch is reset,
turning off the power switch Q1 (Q2). The level at the
negative input of A2 (A4) is set by the error amplifier A1
(A3) and is simply an amplified version of the difference
between the negative feedback voltage and the positive
feedback voltage, usually tied to the reference voltage
VREG. In this manner, the error amplifier sets the correct
peak current level to keep the output in regulation. If the
error amplifier’s output increases, more current is delivered to the output. Similarly, if the error decreases, less
current is delivered. Each switcher functions independently but they share the same oscillator and thus the
switchers are always in phase. Enabling the part is done by
taking either SHDN/SS pin above 1.8V. Disabling the part
is done by grounding both SHDN/SS pins. The soft-start
feature of the LT3471 allows for clean start-up conditions
by limiting the amount of voltage rise at the output of
comparator A1 and A2, which in turn limits the peak
switching current. The soft-start feature for each switcher
is enabled by slowly ramping that switcher’s SHDN/SS pin,
using an RC network, for example. Typical resistor and
capacitor values are 0.33µF and 4.7kΩ, allowing for a
start-up time on the order of milliseconds. The LT3471 has
a current limit circuit not shown in the Block Diagram. The
switch current is constantly monitored and not allowed to
exceed the maximum switch current (typically 1.6A). If the
switch current reaches this value, the SR latch is reset
regardless of the state of the comparator A2 (A4). Also not
shown in the Block Diagram is the thermal shutdown
circuit. If the temperature of the part exceeds approximately 160°C, both latches are reset regardless of the state
of comparators A2 and A4. The current limit and thermal
shutdown circuits protect the power switch as well as the
external components connected to the LT3471.
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APPLICATIONS INFORMATION
Duty Cycle
The typical maximum duty cycle of the LT3471 is 94%.
The duty cycle for a given application is given by:
DC =
| VOUT | + | VD | – | VIN |
| VOUT | + | VD | – | VCESAT |
Where VD is the diode forward voltage drop and VCESAT is
in the worst case 330mV (at 1.3A)
The LT3471 can be used at higher duty cycles, but it must
be operated in the discontinuous conduction mode so that
the actual duty cycle is reduced.
Setting Output Voltage
Setting the output voltage depends on the topology used.
For normal noninverting boost regulator topologies:
⎛ R1⎞
VOUT = VFBP ⎜ 1 + ⎟
⎝ R2 ⎠
where VFBN is connected between R1 and R2 (see the
Typical Applications section for examples).
Select values of R1 and R2 according to the following
equation:
⎛ V
⎞
R1 = R2 ⎜ OUT ⎟
⎝ VREF – 1⎠
A good value for R2 is 15k which sets the current in the
resistor divider chain to 1.00V/15k = 67µA.
VFBP is usually just tied to VREF = 1.00V, but VFBP can also
be tied to a divided down version of VREF or some other
voltage as long as the absolute maximum ratings for the
feedback pins are not exceeded (see Absolute Maximum
Ratings).
For inverting topologies, VFBN is tied to ground and VFBP
is connected between R1 and R2. R2 is between VFBP and
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LT3471
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APPLICATIONS INFORMATION
VREF and R1 is between VFBP and VOUT (see the Applications section for examples). In this case:
⎛ R1⎞
VOUT = VREF ⎜ ⎟
⎝ R2 ⎠
Select values of R1 and R2 according to the following
equation:
⎛V ⎞
R1 = R2 ⎜ OUT ⎟
⎝ VREF ⎠
A good value for R2 is 15k, which sets the current in the
resistor divider chain to 1.00V/15k = 67µA.
Switching Frequency and Inductor Selection
The LT3471 switches at 1.2 MHz, allowing for small valued
inductors to be used. 4.7µH or 10µH will usually suffice.
Choose an inductor that can handle at least 1.4A without
saturating, and ensure that the inductor has a low DCR
(copper-wire resistance) to minimize I2R power losses.
Note that in some applications, the current handling
requirements of the inductor can be lower, such as in the
SEPIC topology where each inductor only carries one half
of the total switch current. For better efficiency, use similar
valued inductors with a larger volume. Many different
sizes and shapes are available from various manufacturers. Choose a core material that has low losses at 1.2 MHz,
such as ferrite core.
Table 1. Inductor Manufacturers
Sumida
(847) 956-0666
www.sumida.com
TDK
(847) 803-6100
www.tdk.com
Murata
(714) 852-2001
www.murata.com
Soft-Start and Shutdown Features
To shut down the part, ground both SHDN/SS pins. To
shut down one switcher but not the other one, ground that
switcher’s SHDN/SS pin. The soft-start feature provides a
way to limit the inrush current drawn from the supply upon
start-up. To use the soft-start feature for either switcher,
slowly ramp up that switcher’s SHDN/SS pin. The rate of
voltage rise at the output of the switcher’s comparator (A1
or A3 for switcher 1 or switcher 2 respectively) tracks the
rate of voltage rise at the SHDN/SS pin once the SHDN/SS
pin has reached about 1.1V. The soft-start function will go
away once the voltage at the SHDN/SS pin exceeds 1.8V.
See the Peak Switch Current vs SHDN/SS Voltage graph in
the Typical Performance Characteristics section. The rate
of voltage rise at the SHDN/SS pin can easily be controlled
with a simple RC network connected between the control
signal and the SHDN/SS pin. Typical values for the RC
network are 4.7kΩ and 0.33µF, giving start-up times on
the order of milliseconds. This RC time constant can be
adjusted to give different start-up times. If different values
of resistance are to be used, keep in mind the SHDN/SS
Current vs SHDN/SS voltage graph along with the Peak
Switch Current vs SHDN/SS Voltage graph, both found in
the Typical Performance Characteristics section. The impedance looking into the SHDN/SS pin depends on whether
the SHDN/SS is above or below VIN. Normally SHDN/SS
will not be driven above VIN, and thus the impedance looks
like 100kΩ in series with a diode. If the voltage of the
SHDN/SS pin is above VIN, the impedance looks more like
50kΩ in series with a diode. This 100kΩ or 50kΩ impedance can have a slight effect on the start-up time if you
choose the R in the RC soft-start network too large.
Another consideration is selecting the soft-start time so
that the soft-start feature is dominated by the RC network
and not the capacitor on VREF. (See VREF voltage reference
section of the Applications Information for details.)
CAPACITOR SELECTION
Low ESR (equivalent series resistance) capacitors should
be used at the output to minimize the output ripple voltage.
Multi-layer ceramic capacitors are an excellent choice, as
they have extremely low ESR and are available in very
small packages. X5R dielectrics are preferred, followed by
X7R, as these materials retain the capacitance over wide
voltage and temperature ranges. A 4.7µF to 15µF output
capacitor is sufficient for most applications, but systems
with very low output currents may need only a 1µF or 2.2µF
output capacitor. Solid tantalum or OS-CON capacitors
can be used, but they will occupy more board area than a
ceramic and will have a higher ESR. Always use a capacitor
with a sufficient voltage rating.
Ceramic capacitors also make a good choice for the input
decoupling capacitor, which should be placed as close as
possible to the LT3471. A 4.7µF to 10µF input capacitor is
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LT3471
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APPLICATIONS INFORMATION
L1
2.2µH
D1
VIN
CONTROL 1
1.8V
0V
RSS1
4.7k
9
SW1
SHDN/SS1
FB1N
CSS1
0.33µF
VIN
2.6V TO 4.2V
Li-Ion
CONTROL 2
1.8V
0V
10
FB1P
8
10µF
RSS2
4.7k
CSS2
0.33µF
VREF
VIN
1
2
SHDN/SS2
GND
R3
90.9k
VOUT1
7V
C3
4.7µF
R4
15k
3
C2
0.1µF
LT3471
FB2N
7
CPL
33pF
FB2P
5
R2
15k
4
SW2
11
6
L2
10µH
C5
1µF
L3
15µH
R1
105k
C6
75pF
VIN
C4
10µF
D2
VOUT2
–7V
3471 F02
C1, C2: X5R OR X7R 6.3V
C3, C4: X5R OR X7R 10V
C5: XR5 OR X7R 16V
CPL: OPTIONAL
D1, D2: ON SEMICONDUCTOR MBRM-120
L1: SUMIDA CR43-2R2
L2: SUMIDA CDRH4D18-100
L3: SUMIDA CDRH4D18-150
Figure 2. Li-Ion OLED Driver
Supply Current of Figure 2 During
Start-Up without Soft-Start RC Network
Supply Current of Figure 2 During
Start-Up with Soft-Start RC Network
ISUPPLY
0.5A/DIV
ISUPPLY
0.5A/DIV
VOUT1
2V/DIV
VOUT1
2V/DIV
0.1ms/DIV
3471 F02b
sufficient for most applications. Table 2 shows a list of
several ceramic capacitor manufacturers. Consult the
manufacturers for detailed information on their entire
selection of ceramic parts.
Table 2. Ceramic Capacitor Manufacturers
Taiyo Yuden
(408) 573-4150
www.t-yuden.com
AVX
(803) 448-9411
www.avxcorp.com
Murata
(714) 852-2001
www.murata.com
The decision to use either low ESR (ceramic) capacitors or
the higher ESR (tantalum or OS-CON) capacitors can
0.2ms/DIV
3471 F02c
affect the stability of the overall system. The ESR of any
capacitor, along with the capacitance itself, contributes a
zero to the system. For the tantalum and OS-CON capacitors, this zero is located at a lower frequency due to the
higher value of the ESR, while the zero of a ceramic capacitor is at a much higher frequency and can generally be
ignored.
A phase lead zero can be intentionally introduced by
placing a capacitor (CPL) in parallel with the resistor (R3)
between VOUT and VFB as shown in Figure 2. The frequency
of the zero is determined by the following equation.
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LT3471
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APPLICATIONS INFORMATION
ƒZ =
VREG VOLTAGE REFERENCE
1
2π • R3 • CPL
By choosing the appropriate values for the resistor and
capacitor, the zero frequency can be designed to improve
the phase margin of the overall converter. The typical
target value for the zero frequency is between 35kHz to
55kHz. Figure 3 shows the transient response of the stepup converter from Figure 2 without the phase lead capacitor CPL. Although adequate for many applications, phase
margin is not ideal as evidenced by 2-3 “bumps” in both
the output voltage and inductor current. A 33pF capacitor
for CPL results in ideal phase margin, which is revealed in
Figure 4 as a more damped response and less overshoot.
VOUT
200mV/DIV
AC COUPLED
IL1
0.5A/DIV
AC COUPLED
Pin 3 of the LT3471 is a bandgap voltage reference that has
been divided down to 1.00V and buffered for external use.
This pin must be bypassed with at least 0.01µF and no
more than 1µF. This will ensure stability as well as reduce
the noise on this pin. The buffer has a built-in current limit
of at least 1mA (typically 1.4mA). This not only means that
you can use this pin as an external reference for supplemental circuitry, but it also means that it is possible to
provide a soft-start feature if this pin is used as one of the
feedback pins for the error amplifier. Normally the softstart time will be dominated by the RC time constant
discussed in the soft-start and shutdown section. However, because of the finite current limit of the buffer for the
VREG pin, it will take some time to charge up the bypass
capacitor. During this time, the voltage at the VREG pin will
ramp up, and this action provides an alternate means for
soft-starting the circuit. If the largest recommended bypass capacitor is used, 1µF, the worst-case (longest) softstart function that would be provided from the VREF pin is:
1µF • 1.00 V
= 1.0ms
1.0mA
LOAD CURRENT
100mA/DIV
AC COUPLED
50µs/DIV
3471 F03
Figure 3. Transient Response of Figure 2’s Step-Up
Converter without Phase Lead Capacitor
VOUT
200mV/DIV
AC COUPLED
IL1
0.5A/DIV
AC COUPLED
LOAD CURRENT
100mA/DIV
AC COUPLED
50µs/DIV
Choose the RC network such that the soft-start time is
longer than this time, or choose a smaller bypass capacitor for the VREF pin (but always larger than 0.01µF) so that
the RC network dominates the soft-starting of the LT3471.
The voltage at the VREF pin can also be divided down and
used for one of the feedback pins for the error amplifier.
This is especially useful in LED driver applications, where
the current through the LEDs is set using the voltage
reference across a sense resistor in the LED chain. Using
a smaller or divided down reference leads to less wasted
power in the sense resistor. See the Typical Applications
section for an example of LED driving applications.
DIODE SELECTION
3471 F04
Figure 4. Transient Response of Figure 2’s Step-Up
Converter with 33pF Phase Lead Capacitor
A Schottky diode is recommended for use with the LT3471.
For high efficiency, a diode with good thermal characteristics at high currents should be used such as the On
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LT3471
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APPLICATIONS INFORMATION
Semiconductor MBRM120. This is a 20V diode. Where the
switch voltage exceeds 20V, use the MBRM140, a 40V
diode. These diodes are rated to handle an average forward current of 1.0A. In applications where the average
forward current of the diode is less than 0.5A, use the
Philips PMEG 2005, 3005, or 4005 (a 20V, 30V or 40V
diode, respectively).
LAYOUT HINTS
The high speed operation of the LT3471 demands careful
attention to board layout. You will not get advertised
performance with careless layout. Figure 5 shows the
recommended component placement.
CONTROL 2
CONTROL 1
CSS1
RSS1
GND
CSS2
RSS2
GND
GND
C4
C1
VOUT2
L2
L1
VOUT1 D1
SW1
10
8
9
C3
•
SHDN/SS1
SW2
6
7
SHDN/SS2
GND
C5
D2
LT3471
PIN 11 GND
R4
FB1P
VREF
FB2P
FB2N
1
2
3
4
5
R3
R2
From Figure 6, the DC gain, poles and zeroes can be
calculated as follows:
RHP Zero: Z3 =
R1
VOUT2
VOUT1
As with any feedback loop, identifying the gain and phase
contribution of the various elements in the loop is critical.
Figure 6 shows the key equivalent elements of a boost
converter. Because of the fast current control loop, the
power stage of the IC, inductor and diode have been
replaced by the equivalent transconductance amplifier
gmp. gmp acts as a current source where the output current
is proportional to the VC voltage. Note that the maximum
output current of gmp is finite due to the current limit in the
IC.
Output Pole: P1=
•
GND
FB1N
Like all other current mode switching regulators, the
LT3471 needs to be compensated for stable and efficient
operation. Two feedback loops are used in the LT3471: a
fast current loop which does not require compensation,
and a slower voltage loop which does. Standard Bode plot
analysis can be used to understand and adjust the voltage
feedback loop.
2
2 • π • RL • COUT
1
Error Amp Pole: P2 =
2 • π • RO • CC
1
Error Amp Zero: Z1=
2 • π • RC • CC
1
V
DC GAIN: A = REF • gma • RO • gmp • RL •
2
VOUT
1
ESR Zero: Z2 =
2 • π • RESR • COUT
L3
VCC
Compensation—Theory
C2
3471 F05
Figure 5. Suggested Layout Showing a Boost on SW1 and an
Inverter on SW2. Note the Separate Ground Returns for All High
Current Paths (Using a Multilayer Board)
VIN2 • RL
2 • π • VOUT2 • L
f
High Frequency Pole: P3 > S
3
1
Phase Lead Zero: Z 4 =
2 • π • R1 • CPL
1
Phase Lead Pole: P4 =
R1 • R2
2 • π • CPL •
R1 + R2
3471f
9
LT3471
U
U
W
U
APPLICATIONS INFORMATION
Table 3. Bode Plot Parameters
–
Parameter
gmp
VOUT
+
CPL
+
VC
RESR
COUT
1.00V
REFERENCE
gma
RC
RO
RL
R1
–
R2
CC
3471 F06
CC: COMPENSATION CAPACITOR
COUT: OUTPUT CAPACITOR
CPL: PHASE LEAD CAPACITOR
gma: TRANSCONDUCTANCE AMPLIFIER INSIDE IC
gmp: POWER STAGE TRANSCONDUCTANCE AMPLIFIER
RC: COMPENSATION RESISTOR
RL: OUTPUT RESISTANCE DEFINED AS VOUT DIVIDED BY ILOAD(MAX)
RO: OUTPUT RESISTANCE OF gma
R1, R2: FEEDBACK RESISTOR DIVIDER NETWORK
RESR: OUTPUT CAPACITOR ESR
Figure 6. Boost Converter Equivalent Model
The Current Mode zero is a right half plane zero which can
be an issue in feedback control design, but is manageable
with proper external component selection.
Using the circuit of Figure 2 as an example, Table 3 shows
the parameters used to generate the Bode plot shown in
Figure 7.
Value
RL
20
Ω
Application Specific
4.7
µF
Application Specific
RESR
10
mΩ
Application Specific
RO
0.9
MΩ
Not Adjustable
CC
90
pF
Not Adjustable
CPL
33
pF
Adjustable
RC
55
kΩ
Not Adjustable
R1
90.9
kΩ
Adjustable
R2
15
kΩ
Adjustable
VOUT
7
V
Application Specific
VIN
3.3
V
Application Specific
gma
50
µmho
Not Adjustable
gmp
9.3
mho
Not Adjustable
L
2.2
µH
fS
1.2
MHz
0
–50
50
Not Adjustable
–100
40
30
–150
20
–200
10
–250
0
PHASE (DEG)
GAIN (dB)
Application Specific
From Figure 7, the phase is –115° when the gain reaches
0dB giving a phase margin of 65°. This is more than
adequate. The crossover frequency is 50kHz.
60
–300
–10
–30
100
Comment
COUT
70
–20
Units
–350
GAIN
PHASE
1k
–400
10k
100k
FREQUENCY (Hz)
1M
3471 F07
Figure 7. Bode Plot of 3.3V to 7V Application
3471f
10
LT3471
U
TYPICAL APPLICATIO S
Li-Ion OLED Driver
L1
2.2µH
D1
VIN
VIN
2.6V TO 4.2V
Li-Ion
CONTROL 2
1.8V
0V
10
9
SW1
SHDN/SS1
FB1N
CSS1
0.33µF
FB1P
8
C1
10µF
RSS2 4.7k
CSS2
0.33µF
VREF
VIN
FB2N
7
SHDN/SS2
GND
FB2P
1
2
C3
4.7µF
VOUT1
7V
500mA WHEN VIN = 4.2V
350mA WHEN VIN = 3.3V
250mA WHEN VIN = 2.6V
R4
15k
3
VCONTROL
C2
0.1µF 0V TO 1V
R5
20k
5
R2
15k
4
R6
10k
SW2
11
R3
90.9k
C6
33pF
LT3471
6
L2
15µH
C5
1µF
L3
15µH
R1
105k
VIN
C6
75pF
C4
10µF
D2
3471 TA02
C1, C2: X5R OR X7R 6.3V
C3, C4: X5R OR X7R 10V
C5: XR5 OR X7R 16V
C6: OPTIONAL
D1, D2: ON SEMICONDUCTOR MBRM-120
L1: SUMIDA CR43-2R2
L2: SUMIDA CDRH4D18-100
L3: SUMIDA CDRH4D18-150
VOUT2
–7V TO –4V
–7V WHEN VCONTROL = 0V
–4V WHEN VCONTROL = 1
–7V, 300mA WHEN VIN = 4.2V
–7V, 250mA WHEN VIN = 3.3V
–7V, 200mA WHEN VIN = 2.6V
Li-Ion OLED Driver Efficiency
95
VOUT = 7V
90
VIN = 4.2V
85
EFFICIENCY (%)
CONTROL 1
1.8V
0V
RSS1
4.7k
VIN = 3.3V
VIN = 2.6V
80
VIN = 4.2V
VIN = 3.3V
75
70
VIN = 2.6V
65
60
VOUT = –7V
55
50
0
100
300
200
IOUT (mA)
400
500
3471 TA02b
3471f
11
LT3471
U
TYPICAL APPLICATIO S
Single Li-Ion Cell to 5V, 12V Boost Converter
L1
3.3µH
C3
10µF
VOUT1
5V
900mA IF VIN = 4.2V
630mA IF VIN = 3.3V
425mA IF VIN = 2.6V
C4
10µF
VOUT2
12V
300mA IF VIN = 4.2V
210mA IF VIN = 3.3V
145mA IF VIN = 2.6V
D1
VIN
CONTROL 1
1.8V
OV
9
SW1
SHDN/SS1
FB1N
CSS1
0.33µF
FB1P
8
VIN
2.6V TO 4.2V
CONTROL 2
1.8V
0V
10
RSS1
4.7k
RSS2
4.7k
VREF
VIN
CSS2
0.33µF
2
C2
0.1µF
FB2P
SHDN/SS2
GND
11
R1
20k
R2
4.99k
3
LT3471
C1
4.7µF
7
C5
100pF
1
FB2N
4
5
SW2
L2
6.8µH
6
VIN
D2
C6
220pF
R3
54.9k
R4
4.99k
3471 TA03
C1-C3: X5R OR X7R 6.3V
C4: X5R OR X7R 16V
D1, D2: ON SEMICONDUCTOR MBRM-120
L1: SUMIDA CR43-3R3
L2: SUMIDA CR43-6R8
3471f
12
LT3471
U
TYPICAL APPLICATIO S
Li-Ion 20 White LED Driver
L1
2.2µH
D1
VIN
CONTROL 1
1.8V
OV
9
SW1
SHDN/SS1
FB1N
CSS1
0.33µF
FB1P
8
VIN
2.6V TO 4.2V
CONTROL 2
1.8V
OV
C3
0.22µF
10
RSS1
4.7k
RSS2
4.7k
VREF
VIN
CSS2
0.33µF
2
3
C2
0.1µF
FB2P
7
1
LT3471
C1
4.7µF
SHDN/SS2
GND
FB2N
IOUT1
20mA
4
R1
90.9k
10 WHITE LEDs
R2
10k
5
SW2
11
6
4.99Ω
L2
2.2µH
VIN
D2
C4
0.22µF
IOUT2
20mA
C1, C2: X5R OR X7R 6.3V
C3, C4: X5R OR X7R 50V
D1, D2: ON SEMICONDUCTOR MBRM-140
L1, L2: SUMIDA CDRH2D-2R2
10 WHITE LEDs
4.99Ω
3471 TA04
3471f
13
LT3471
U
TYPICAL APPLICATIO S
Li-Ion or 4-Cell Alkaline to 3.3V and 5V SEPIC
C3
4.7µF
L1
10µH
D1
VIN
CONTROL 1
1.8V
OV
10
RSS1
4.7k
9
SW1
SHDN/SS1
FB1N
CSS1
0.33µF
FB1P
8
VIN
2.6V TO 6.5V
CONTROL 2
1.8V
OV
L2
10µH
RSS2
4.7k
VREF
VIN
CSS2
0.33µF
1
2
R1
34.8k
C2
0.1µF
FB2P
SHDN/SS2
GND
C4
15µF
R2
15k
3
LT3471
C1
4.7µF
7
C7
56pF
VOUT1
3.3V
640mA AT VIN = 6.5V
550mA AT VIN = 5V
470mA AT VIN = 4V
410mA AT VIN = 3.3V
340mA AT VIN = 2.6V
FB2N
4
5
SW2
11
6
L3
10µH
C5
10µF
D2
VIN
C1, C3, C5: X5R OR X7R 10V
C4, C6: X5R OR X7R 6.3V
D1, D2: ON SEMICONDUCTOR MBRM-120
L1-L4: MURATA LQH43CN100K032
L4
10µH
C6
15µF
C8
R3
56pF 60.4k
R4
15k
VOUT2
5V
500mA AT VIN = 6.5V
420mA AT VIN = 5V
360mA AT VIN = 4V
300mA AT VIN = 3.3V
250mA AT VIN = 2.6V
3471 TA05
3471f
14
LT3471
U
PACKAGE DESCRIPTIO
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
0.675 ±0.05
3.50 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
R = 0.115
TYP
6
3.00 ±0.10
(4 SIDES)
0.38 ± 0.10
10
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
(DD10) DFN 1103
5
0.200 REF
1
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
0.00 – 0.05
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3471f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT3471
U
TYPICAL APPLICATIO
5V to ±12V Dual Supply Boost/Inverting Converter
L1
10µH
D1
VIN
10
CONTROL 1
1.8V
OV
4.7k
9
SW1
SHDN/SS1
FB1N
FB1P
0.33µF
8
VIN
5V
CONTROL 2
1.8V
OV
VREF
VIN
0.33µF
SHDN/SS2
GND
FB2N
4
5
C7
56pF
6
•
VIN
R3
15k
SW2
11
L2
10µH
C3
4.7µF
R2
4.99k
3
C2
0.1µF
FB2P
7
2
LT3471
C1
4.7µF
4.7k
R1
C6
56pF 54.9k
1
VOUT1
12V
320mA
R4
182k
•
C5
1µF
D2
L3
10µH
C4
4.7µF
VOUT2
–12V
200mA
3471 TA06
C1, C2: X5R OR X7R 6.3V
C3, C4: X5R OR X7R 16V
C5: X5R OR X7R 25V
D1, D2: ON SEMICONDUCTOR MBRM-120
L1: SUMIDA CR43-10
L2, L3: SUMIDA CLS63-10
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1611
550mA (ISW), 1.4MHz, High Efficiency Micropower Inverting
DC/DC Converter
VIN: 1.1V to 10V, VOUT(MAX) = –34V, IQ = 3mA, ISD < 1µA,
ThinSOT Package
LT1613
550mA (ISW), 1.4MHz, High Efficiency Step-Up
DC/DC Converter
VIN: 0.9V to 10V, VOUT(MAX) = 34V, IQ = 3mA, ISD < 1µA,
ThinSOT Package
LT1614
750mA (ISW), 600kHz, High Efficiency Micropower Inverting
DC/DC Converter
VIN: 1V to 12V, VOUT(MAX) = –24V, IQ = 1mA, ISD < 10µA,
MS8, S8 Packages
LT1615/LT1615-1
300mA/80mA (ISW), High Efficiency Step-Up DC/DC Converters VIN = 1V to 15V, VOUT(MAX) = 34V, IQ = 20µA, ISD < 1µA,
ThinSOT Package
LT1617/LT1617-1
350mA/100mA (ISW), High Efficiency Micropower Inverting
DC/DC Converters
VIN = 1.2V to 15V, VOUT(MAX) = –34V, IQ = 20µA, ISD < 1µA,
ThinSOT Package
LT1930/LT1930A
1A (ISW), 1.2MHz/2.2MHz, High Efficiency
Step-Up DC/DC Converters
VIN: 2.6V to 16V, VOUT(MAX) = 34V, IQ = 4.2mA/5.5mA,
ISD < 1µA, ThinSOT Package
LT1931/LT1931A
1A (ISW), 1.2MHz/2.2MHz High Efficiency Micropower Inverting VIN = 2.6V to 16V, VOUT(MAX) = –34V, IQ = 5.8mA, ISD < 1µA,
DC/DC Converters
ThinSOT Package
LT1943 (Quad)
Quad Boost, 2.6A Buck, 2.6A Boost, 0.3A Boost, 0.4A Inverter
1.2MHz TFT DC/DC Converter
VIN = 4.5V to 22V, VOUT(MAX) = 40V, IQ = 10µA, ISD < 35µA,
TSSOP28E Package
LT1945 (Dual)
Dual Output, Boost/Inverter, 350mA (ISW), Constant Off-Time,
High Efficiency Step-Up DC/DC Converter
VIN = 1.2V to 15V, VOUT(MAX) = ±34V, IQ = 40µA, ISD < 1µA,
10-Lead MS Package
LT1946/LT1946A
1.5A (ISW), 1.2MHz/2.7MHz, High Efficiency
Step-Up DC/DC Converters
VIN: 2.45V to 16V, VOUT(MAX) = 34V, IQ = 3.2mA, ISD < 1µA,
MS8 Package
LT3436
3A (ISW), 1MHz, 34V Step-Up DC/DC Converter
VIN: 3V to 25V, VOUT(MAX) = 34V, IQ = 0.9mA, ISD < 6µA,
TSSOP16E Package
LT3462/LT3462A
300mA (ISW), 1.2MHz/2.7MHz, High Efficiency Inverting
DC/DC Converters with Integrated Schottkys
VIN = 2.5V to 16V, VOUT(MAX) = –38V, IQ = 2.9mA, ISD < 1µA,
ThinSOT Package
LT3463/LT3463A
Dual Output, Boost/Inverter, 250mA (ISW), Constant Off-Time,
High Efficiency Step-Up DC/DC Converters with Integrated
Schottkys
VIN = 2.3V to 15V, VOUT(MAX) = ±40V, IQ = 40µA, ISD < 1µA,
DFN Package
LT3464
85mA (ISW), High Efficiency Step-Up DC/DC Converter with
Integrated Schottky and PNP Disconnect
VIN = 2.3V to 10V, VOUT(MAX) = 34V, IQ = 25µA, ISD < 1µA,
ThinSOT Package
3471f
16
Linear Technology Corporation
LT/TP 0804 1K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
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