DC275A - Demo Manual

DEMO MANUAL DC275
DC/DC CONVERTER
DESCRIPTIO
LTC1702
Dual 550kHz Synchronous
2-Phase 15A DC/DC Converter
U
Demonstration circuit DC275 is a dual, high efficiency
regulator using the LTC®1702 switching regulator controller. The LTC1702 is optimized for high efficiency with
low input voltages. Typical applications are power for a
digital signal processor (DSP), microprocessor and/or
an application specific integrated circuit (ASIC). The
input voltage of the LTC1702 can range from 3V to 7V.
One of the output voltages (VOUT2) is fixed at 3.3V and the
other (VOUT1) is programmable from 1.6V to 2.5V by
means of a jumper. The LTC1702 includes two complete,
on-chip, independent switching regulator controllers,
each designed to drive a pair of external N-channel
MOSFET devices in a voltage mode control, synchronous
buck configuration. The LTC1702 also provides opendrain logic outputs (PGOOD1 and PGOOD2) that indicate
whether either output has risen to within 5% of the final
output voltage. An optional latching fault mode protects
the load if the output rises 15% above the intended
voltage. The LTC1702 uses a constant 550kHz switching
frequency, minimizing external component size and maximizing load transient performance. Operating efficiencies exceeding 90% are obtained for load current currents
from 1A to 14A. Additionally, the supply current in
shutdown is less than 100µA. Gerber files for this circuit
board are available. Call the LTC factory.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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PERFOR A CE SU
ARY
PARAMETER
CONDITIONS
VALUE
VIN
Input Voltage Range
4.75V to 7V
VOUT2
Fixed Output Voltage
3.3V
IOUT2
Maximum Output Load Current
Typical Output Ripple
IOUT = 15A
VOUT1
Jumper Selectable Output Voltage
IOUT1
Maximum Output Load Current
Typical Output Ripple
IOUT = 15A
17mV
IQ
Supply Current in Shutdown
100µA
15A
18mV
1.6V, 1.8V, 2V or 2.5V
15A
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TYPICAL PERFOR A CE CHARACTERISTICS A D BOARD PHOTO
LTC1702 Efficiency
EFFICIENCY (%)
100
VIN = 5V
VOUT = 3.3V
VOUT = 2.5V
90
VOUT = 1.6V
80
70
0
5
10
LOAD CURRENT (A)
15
1702 G01
1
2
E2
PWRGD1
E3
SD1
E8
GND
+
+
+
R8
10k
VIN
C9
180µF
C8
C10
1µF 180µF
VOUT1
1.6V, 1.8V,
2V OR 2.5V
AT 15A
E1
C11
180µF
C12
180µF
+
R7
20k
2V
1%
JP2
JP1
C13
820pF
Q3
R6
40.2k
1.8V
1%
D3
Q1
1.6V
1.8V
2.0V
2.5V
JP1
OUT
IN
OUT
OUT
JP2
OUT
OUT
IN
OUT
JP3
R16
8.87k
2.5V
1%
Q4
Q2
JP3
OUT
OUT
OUT
IN
VOUT1 JUMPER SELECT
R5
10k
1.6V
1%
R4
10k
1%
R3
1.2k
L1
1µH
C23
1µF
C14
680pF
R9 47k
C15
27pF
C7 1µF
R2 27k
C5
1µF
12
11
10
9
8
7
6
5
4
3
2
1
FB1
SGND
COMP1
FAULT
PGOOD2
PGND
SW2
TG2
BG2
BOOST2
IMAX2
VCC
FB2
COMP2
RUN/SS2
LTC1702
RUN/SS1
FCB
PGOOD1
IMAX1
SW1
TG1
BG1
BOOST1
PVCC
13
14
15
16
17
18
19
20
21
22
23
24
C17 1µF
C16 1µF
R10 27k
D1
C2
330µF
22 BG2
21 TG2
20 SW2
19 PGND
18 PGOOD2
17 FAULT
16 RUN/SS2
4
5
6
7
8
9
TG1
SW1
IMAX1
PGOOD1
FCB
RUN/SS1
13 VCC
FB1 12
LTC1702CGN
GN PACKAGE
24-LEAD NARROW PLASTIC SSOP
14 FB2
SGND 11
15 COMP2
23 BOOST2
3
BG1
COMP1 10
24 IMAX2
2
BOOST1
TOP VIEW
1
PVCC
+
C1
330µF
C21
27pF
+
Figure 1. Dual 550kHz Synchronous 2-Phase 15A DC/DC Converter
BURST
JP4
CONT
VIN
C6
1µF
D2
C4
10µF
C22
3300pF
C18
680pF
Q8
Q6
C3
330µF
R15 68k
Q7
Q5
+
D4
R13
4.99k
1%
R12
15.8k
1%
R11
1.6k
L2
1µH
C24
1µF
+
+
R14
10k
VIN
+
C26
C19
180µF 180µF
+ C25
180µF
DC175 F01
E5
PWRGD2
E6
FAULT
E7
SD2
E11
GND
C20
1µF
C27
180µF
E4
VOUT2
3.3V/15A
E9
GND
E10
VIN
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R1
10Ω
5%
PACKAGE A D SCHE ATIC DIAGRA SM
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NOTES: UNLESS OTHERWISE SPECIFIED
D1, D2: ON SEMICONDUCTOR MBR0520LT1
D3, D4: ON SEMICONDUCTOR MBRS340T3
Q1 TO Q8: FAIRCHILD FDS6670A
DEMO MANUAL DC275
DC/DC CONVERTER
DEMO MANUAL DC275
DC/DC CONVERTER
PARTS LIST
REFERENCE
DESIGNATOR
QUANTITY
PART NUMBER
DESCRIPTION
VENDOR
TELEPHONE
330µF 10V 10% Tantalum Capacitor
Kemet
(408) 986-0424
C1 to C3
3
T510X337K010AS
C4
1
1206ZG106ZAT1A
10µF 10V Y5V Capacitor
AVX
(843) 946-0362
C5 to C8, C16,
C17, C20, C23
C24
9
06036D105MAT1A
1µF 6V X5R Capacitor
AVX
(843) 946-0362
C9 to C12,
C19, C25 to
C27
8
EEFUEOG181R
180µF 4V SP Capacitor
Panasonic
(714) 373-7334
C13
1
06035C821MAT1A
820pF 50V X7R Capacitor
AVX
(843) 946-0362
C14, C18
2
06035C681MAT1A
680pF 50V X7R Capacitor
AVX
(843) 946-0362
C15, C21
2
06035A270MAT1A
27pF 50V NPO Capacitor
AVX
(843) 946-0362
C22
1
06035C332MAT1A
3300pF 50V X7R Capacitor
AVX
(843) 946-0362
D1, D2
2
MBR0520LT1
Schottky Diode
ON Semiconductor
(602) 244-6600
D3, D4
2
MBRS340T3
Schottky Diode
ON Semiconductor
(602) 244-6600
E1, E4, E8 to E11
6
2501-2
1-Pin Terminal
Mill-Max
(516) 922-6000
E2, E3, E5 to E7
5
2308-2
1-Pin Terminal
Mill-Max
(516) 922-6000
JP1 to JP3
3
3801S-02G2
0.100"CC 2-Pin Jumper
Comm Con
(626) 301-4200
JP4
1
3801S-03G2
0.100"CC 3-Pin Jumper
Comm Con
(626) 301-4200
JP1, JP4
2
CCIJ230-G
0.100"CC Shunt
Comm Con
(626) 301-4200
L1, L2
2
CEP125-1R0MC-H or
ETQP6F1R0SSP
1µH 20A SMT Inductor
Sumida
Panasonic
(847) 956-0667
(714) 373-7334
Q1 to Q8
8
FDS6670A
SO-8 N-Channel MOSFET
Fairchild
(408) 822-2126
R1
1
CR16-100JM
10Ω 1/16W 5% Chip Resistor
Tad
(714) 255-9123
R2, R10
2
CR16-273JM
27k 1/16W 5% Chip Resistor
Tad
(714) 255-9123
R3
1
CR16-122JM
1.2k 1/16W 5% Chip Resistor
Tad
(714) 255-9123
R4, R5
2
CR16-1002FM
10k 1/16W 1% Chip Resistor
Tad
(714) 255-9123
R6
1
CR16-4022FM
40.2k 1/16W 1% Chip Resistor
Tad
(714) 255-9123
R7
1
CR16-2002FM
20k 1/16W 1% Chip Resistor
Tad
(714) 255-9123
R8, R14
2
CR16-103JM
10k 1/16W 5% Chip Resistor
Tad
(714) 255-9123
R9
1
CR16-473JM
47k 1/16W 5% Chip Resistor
Tad
(800) 508-1521
R11
1
CR16-162JM
1.6k 1/16W 5% Chip Resistor
Tad
(800) 508-1521
R12
1
CR16-1582FM
15.8k 1/16W 1% Chip Resistor
Tad
(800) 508-1521
R13
1
CR16-4991FM
4.99k 1/16W 1% Chip Resistor
Tad
(800) 508-1521
R15
1
CR16-683JM
68k 1/16W 5% Chip Resistor
Tad
(800) 508-1521
R16
1
CR16-8871FM
8.87k 1/16W 1% Chip Resistor
Tad
(800) 508-1521
U1
1
LTC1702CGN
24-Lead SSOP IC
LTC
(408) 432-1900
3
DEMO MANUAL DC275
DC/DC CONVERTER
QUICK START GUIDE
Refer to Figure 2 for proper measurement equipment
setup and follow the procedure outlined below:
6. The SD1 and SD2 pins should be left floating for
normal operation and tied to GND for shutdown.
1. Connect the input power to the VIN and GND terminals
on the board using 12-gauge or heavier wire soldered
to the terminals. The input voltage is limited to
between 4.75V and 7V.
7. Connect a voltmeter across the VIN and GND terminals to measure input voltage.
8. Connect a voltmeter across the VOUT1 and GND
terminals and another across the VOUT2 and GND
terminals to measure the output voltages.
2. Connect an ammeter in series with the input supply to
measure input current.
9. For applications where the minimum load current is
greater than 1A, set jumper JP4 to the “Continuous”
position.
3. Since this demo board operates from a low input
voltage and supplies high output current, it is essential that the input supply voltage be well regulated. If
the input power supply is equipped with remote
sense lines, connect SENSE + to the VIN terminal and
SENSE – to GND terminal on the board.
10. Set the desired output voltage (VOUT1) with jumpers
JP1 to JP3, as shown in Table 1.
11. After all connections are made, turn on the power and
verify that VOUT1 and VOUT2 are correct.
4. Connect either power resistors or an electronic load
to the VOUT1, VOUT2 and GND terminals using
12-gauge or heavier wire, soldered to the terminals.
Table 1
5. Connect an ammeter in series with each of the output
loads to measure output currents.
POSITION
OUTPUT VOLTAGE
No Jumper
1.6V
JP1
1.8V
JP2
2.0V
JP3
2.5V
A
INPUT
SUPPLY
V
PWRGD2
FAULT
PWRGD1
GND
VIN
SD1
SD2
BURST CONT
GND
JP2
JP1
JP3
LOAD
GND
LOAD
V
V
A
VOUT2
DC275A-LTC1702CGN
DUAL OUTPUT BUCK REGULATORS
Figure 2. Proper Measurement Setup
4
A
VOUT1
DEMO MANUAL DC275
DC/DC CONVERTER
U
OPERATIO
The circuit in Figure 1 highlights the capabilities of the
LTC1702. This design provides one fixed 3.3V output
(VOUT2) and one output (VOUT1) that is jumper selectable
from 1.6V to 2.5V. The LTC1702 is a voltage mode
controller, designed to drive a pair of external N-channel
MOSFETs using a fixed 550kHz switching frequency. The
synchronous buck architecture automatically shifts to
discontinuous operation and then to Burst ModeTM operation as the output load decreases, ensuring maximum
efficiency over a wide range of load currents. This mode is
recommended for load currents less than 1A and can be
implemented on the demo board by moving jumper JP4 to
the “Burst” position.
Theory of Operation
The LTC1702 has two independent switching regulators.
For the sake of simplicity and to minimize repetition, only
side “1” will be discussed. The divided output (VOUT1) is
compared to the 0.8V reference. The difference voltage is
multiplied by the error amplifier’s (FB) gain. The resulting
error signal is then compared to an internally generated,
fixed frequency sawtooth waveform by the PWM comparator, which generates a pulse width modulated signal.
This PWM signal drives the external MOSFETs through
TG1 and BG1. The output of this chopper circuit is then
filtered by L1 and C9 to C12 to produce the desired DC
output voltage.
2-Phase Operation
The LTC1702 dual switching regulator controller also
features the considerable benefits of 2-phase operation.
The LTC1702 includes a single master clock that drives the
two sides such that side 1 is 180° out of phase with side
2. This technique, known as 2-phase switching, has the
effect of doubling the frequency of the switching pulses
seen by the input capacitor and significantly reduces their
RMS value. With 2-phase switching, the input capacitor is
sized as required to support the larger of the two sides at
maximum load current. As the load current increases on
the lower current side, it tends to cancel, rather than add
to, the RMS current seen by the input capacitor; thus no
additional capacitance is needed.
Capacitor Considerations
The input capacitors are Kemet T510X337K010AS, 330µF,
10V tantalums. The input capacitors must be rated for the
RMS input ripple. A good rule of thumb is that the input
ripple current will be 50% of the output current. Since the
LTC1702 uses 2-phase switching, the input bulk capacitors should be able to fully handle the RMS ripple current
of just one load. As the load current increases on the other
side, it tends to cancel, rather than to add to, the ripple
current requirements for the input capacitors. For a continuous output current of 15A, the ripple current rating of
the input capacitors should be 7.5A. The capacitors chosen are rated at 2.5A each, so three are adequate. Without
the 2-phase operation, six capacitors would be required to
handle two 15A loads.
Output capacitors need to have a ripple current rating
greater than the RMS value of the inductor ripple current.
This is a function of the operating frequency and inductor
value, as well as input and output voltages. Because the
ripple current is relatively small, the controlling parameter
is generally the capacitor’s ESR (equivalent series resistance). The maximum allowable ESR is equal to the
maximum allowable peak-to-peak output ripple voltage
divided by the peak-to-peak inductor ripple current. In
general, if the ESR is low enough for the ripple voltage and
transient requirements, the capacitors will have more than
adequate ripple current capability.
Inductor Selection
Inductor selection is not extremely critical. The inductor
used here was chosen for fairly low cost and ready
availability. The main concerns in choosing an appropriate inductor are the inductance value required, the saturation current rating and the temperature rise. Most
manufacturers specify a DC current rating that produces
a temperature rise of 40°C. If a design will not see high
ambient temperatures, a larger temperature rise can
usually be tolerated. Another maximum current specification is related to core saturation. A manufacturer may
specify that maximum rated current is the point at which
inductance is down by 10% (some specify 25%). Since
most core materials and structures will result in a gentle,
Burst Mode is a trademark of Linear Technology Corporation.
5
DEMO MANUAL DC275
DC/DC CONVERTER
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OPERATIO
controlled roll off of inductance with DC bias, there is no
magical point where the inductor is no longer useful. Look
at what the inductance will be at the maximum load current
expected and determine if the output ripple will remain
within specified limits. If it will, the inductor will most likely
work correctly. Ripple current is generally designed for
between 10% and 40% of output current.
conventional soft-start pin, enforcing a duty cycle limit
proportional to the voltage at RUN/SS. An internal 4µA
current source pull-up is connected to each RUN/SS pin,
allowing a soft-start ramp to be generated with a single
external capacitor (C7 for side 1 and C17 for side 2) to
ground.
Current Limit
MOSFET Selection
The main concern with FET selection in very low voltage
applications is thermal management. At high current levels, power devices will get hot. The trick is to keep the
temperature rise within acceptable limits. Most of the
FETs’ power dissipation will be due to conduction losses.
Therefore, by choosing a FET with a sufficiently low RDS(ON),
the power dissipation, and therefore, the temperature rise,
can be made arbitrarily low. The price paid for very low
temperature rise is more expensive FETs. Switching losses
are a concern only for the high side FET. The low side FET
turns on and off into a forward-biased diode, so its transition losses are very small. The high side FET, in contrast,
must provide all of the reverse recovery charge that the
low side FETs body diode will demand. This can result in
a significant amount of switching loss in this device.
Although it may seem that a lower on-resistance FET is
always desirable from an efficiency perspective, this is not
necessarily true. A smaller device will have a lower gatecharge power requirement and will also exhibit faster
switching transition times. The resulting reduction in AC
losses may more than offset the increase in conduction
losses. A smaller, higher on-resistance FET may prove the
more efficient, as well as the lower cost solution. As the
load current increases, gate-drive losses become less of a
concern. At output currents on the order of 15A, lower
resistance FETs will probably be better in terms of overall
efficiency, but not necessarily the most cost effective
choice. Each application will place a different value on a
few points of efficiency.
Shutdown/Soft-Start
Each half or the LTC1702 has a RUN/SS pin. This pin
performs two functions: when pulled to ground, each
shuts down its half of the LTC1702, and each acts as a
6
The IMAX resistor, R2, sets the current limit by setting the
maximum allowable voltage drop across the bottom
MOSFET before the current limit circuit engages. The
voltage across the bottom MOSFET is determined by its
on-resistance and by the current flowing in the inductor,
which is the same as the output current. To set the current
limit, connect an RIMAX resistor from IMAX to GND. The
value of RIMAX is calculated as follows:
RIMAX = [(ILIM • RDS(ON)) + 100mV]/10µA
ILIM should be chosen to be 150% of the maximum
operating load current to account for MOSFET RDS(ON)
variations with temperature.
How to Measure Voltage Regulation and Efficiency
When trying to measure load regulation or efficiency,
voltage measurements should be made directly across the
VOUT and GND terminals and should not be taken at the end
of test leads at the load. Similarly, input voltage should be
measured directly on the VIN and GND terminals of the
LTC1702 demo board. Input and output current should be
measured by placing an ammeter in series with the input
supply and load. Refer to Figure 2 for the proper test
equipment setup. Refer to page one for typical efficiency
curves for VIN = 5V, VOUT = 3.3V, 2.5V and 1.8V, for
IL = 1A to 15A.
How to Measure Output Voltage Ripple
In order to measure output voltage ripple, care must be
taken to avoid a long ground lead on the oscilloscope
probe. Therefore, a sturdy wire should be soldered on the
output side of the GND terminal. The other end of the wire
is looped around the ground side of the probe and should
be kept as short as possible. The tip of the probe is touched
directly to VOUT (see Figure 3). Bandwidth is generally
DEMO MANUAL DC275
DC/DC CONVERTER
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OPERATIO
limited to 20MHz for ripple measurements. Also, if multiple pieces of line-powered test equipment are used, be
sure to use isolation transformers on their power lines to
prevent ground loops, which can cause erroneous results.
Figures 4 and 5 show the output voltage ripple for the 3.3V
and the 2.5V supplies for a 15A load.
GND
VOUT
Transient Response
The LTC1702 uses true 25MHz gain bandwidth op amps
as the feedback amplifiers. This allows the use of an
OPTI-LOOPTM compensation scheme that can precisely
tailor the loop response. The high gain-bandwidth product allows the loop to be crossed over beyond 50kHz
while maintaining good stability, and significantly
enhances load transient response. Figures 6 and 7 show
the transient response of the 3.3V and the 2.5V output
supplies for a 0A to 10A load step. For more information
about loop compensation and stability analysis, consult
the LTC1702 data sheet.
OPTI-LOOP is a trademark of Linear Technology Corporation.
Figure 3. Measuring Output Voltage Ripple
Figure 4. 3.3V Output Voltage Ripple, IL = 15A
Figure 6. 3.3V Transient Response, IL = 0A to 10A
Figure 5. 2.5V Output Voltage Ripple, IL = 15A
Figure 7. 2.5V Transient Response, IL = 0A to 10A
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DEMO MANUAL DC275
DC/DC CONVERTER
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OPERATIO
Heat Dissipation Issues
Since each side of the LTC1702 demo board can supply
15A of continuous load current, care must be taken not to
exceed the maximum junction temperature for the power
MOSFETs. A few possibilities for dissipating the power are
to use heat sinks and/or forced air cooling. Another
possibility is to use the PC board as a heat sink. On the
LTC1702 demo board, power MOSFETS Q1 to Q8 are
surrounded by ground and power planes on both sides of
the PC board. Also, there is metal on the inner layers
directly underneath the power MOSFETs. This helps in
spreading the heat and improves the power dissipation
capability of the PCB.
Layout Guidelines
Since the LTC1702 is a switching regulator, a good layout
is essential for good load regulation and minimizing radiated/conducted noise. If you want a layout that is guaranteed to work, copy the LTC1702 Gerber files provided with
this demo board; otherwise, be sure to follow the layout
guidelines below:
1. The inductor L1, MOSFETs Q1 to Q4 and the Schottky
diode (D3) should be placed as close as possible to
each other; similarly, L2, Q5 to Q8 and D4 should be
placed as close together as possible. This junction
forms the switch node and should be kept as small as
possible to minimize radiated emissions. It must also
be large enough to carry the full rated output current.
2. The SW1 and the SW2 pins should be connected
directly to the respective switch nodes with a short
trace.
3. C4 (10µF) should be as close as possible to Pin 13 on
the LTC1702.
8
4. C5 (1µF) should be as close as possible to Pin 1 on the
LTC1702.
5. R2 should be connected directly to the sources of Q3
and Q4.
6. R10 should be connected directly to the sources of Q7
and Q8.
7. Keep the trace from the FB1 pin to the junction of R4
and R5 short and use a long trace from the top of
resistor R4 to the output terminal, rather than vice
versa.
8. Keep the trace from the FB2 pin to the junction of R12
and R13 short and use a long trace from the top of
resistor R12 to the output terminal, rather than vice
versa.
9. The sources of the bottom MOSFETs Q3, Q4, Q7 and
Q8 should be tied back to the ground of input capacitors C1 to C3 by means of a wide trace, not by the
ground plane.
10. The grounds of the output capacitors C19–C20,
C25–C27 and C8–C12 should be tied directly to the
input capacitor’s ground by means of a wide trace or
by the ground plane.
11. The grounds of the feedback resistors, soft-start capacitors and C4 should be referenced to the chip SGND
pin, which is then tied to the input bulk capacitors’
grounds.
12. PGND, Pin 19, should connect directly to the ground
plane.
DEMO MANUAL DC275
DC/DC CONVERTER
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PCB LAYOUT A D FIL
Top Silkscreen
Top Solder Mask
Top Pastemask
9
DEMO MANUAL DC275
DC/DC CONVERTER
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PCB LAYOUT A D FIL
10
Layer 1, Top Layer
Layer 2, VIN Plane
Layer 3, GND Plane
Layer 4, Bottom Layer
DEMO MANUAL DC275
DC/DC CONVERTER
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PCB LAYOUT A D FIL
Bottom Silkscreen
Bottom Solder Mask
Bottom Pastemask
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
DEMO MANUAL DC275
DC/DC CONVERTER
U
PC FAB DRAWI G
2.50"
D
A
F
D
A
E
F
E
C
A
A
C
B
2.10"
E
E
A
NOTES: UNLESS OTHERWISE SPECIFIED
1. MATERIAL: FR4 OR EQUIVALENT EPOXY,
2 OZ COPPER CLAD, THICKNESS 0.062 ±0.006
TOTAL OF 4 LAYERS
2. FINISH: ALL PLATED HOLES 0.001 MIN/0.0015 MAX
COPPER PLATE, ELECTRODEPOSITED TIN-LEAD COMPOSITION
BEFORE REFLOW, SOLDER MASK OVER BARE COPPER (SMOBC)
3. SOLDER MASK: BOTH SIDES USING LPI OR EQUIVALENT
4. SILKSCREEN: USING WHITE NONCONDUCTIVE EPOXY INK
5. UNUSED SMD COMPONENTS SHOULD BE FREE OF SOLDER
6. FILL UP ALL VIAS WITH SOLDER
7. SCORING
A
0.017
E
E
A
F
A
F
D
12
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900
FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com
SYMBOL
DIAMETER
NUMBER
OF HOLES
A
0.015
70
B
0.035
5
C
0.064
5
D
0.070
3
E
0.094
6
F
0.125
4
TOTAL HOLES
93
dc275f LT/TP 0100 500 • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2000