DEMO MANUAL DC275 DC/DC CONVERTER DESCRIPTIO LTC1702 Dual 550kHz Synchronous 2-Phase 15A DC/DC Converter U Demonstration circuit DC275 is a dual, high efficiency regulator using the LTC®1702 switching regulator controller. The LTC1702 is optimized for high efficiency with low input voltages. Typical applications are power for a digital signal processor (DSP), microprocessor and/or an application specific integrated circuit (ASIC). The input voltage of the LTC1702 can range from 3V to 7V. One of the output voltages (VOUT2) is fixed at 3.3V and the other (VOUT1) is programmable from 1.6V to 2.5V by means of a jumper. The LTC1702 includes two complete, on-chip, independent switching regulator controllers, each designed to drive a pair of external N-channel MOSFET devices in a voltage mode control, synchronous buck configuration. The LTC1702 also provides opendrain logic outputs (PGOOD1 and PGOOD2) that indicate whether either output has risen to within 5% of the final output voltage. An optional latching fault mode protects the load if the output rises 15% above the intended voltage. The LTC1702 uses a constant 550kHz switching frequency, minimizing external component size and maximizing load transient performance. Operating efficiencies exceeding 90% are obtained for load current currents from 1A to 14A. Additionally, the supply current in shutdown is less than 100µA. Gerber files for this circuit board are available. Call the LTC factory. , LTC and LT are registered trademarks of Linear Technology Corporation. W U WW PERFOR A CE SU ARY PARAMETER CONDITIONS VALUE VIN Input Voltage Range 4.75V to 7V VOUT2 Fixed Output Voltage 3.3V IOUT2 Maximum Output Load Current Typical Output Ripple IOUT = 15A VOUT1 Jumper Selectable Output Voltage IOUT1 Maximum Output Load Current Typical Output Ripple IOUT = 15A 17mV IQ Supply Current in Shutdown 100µA 15A 18mV 1.6V, 1.8V, 2V or 2.5V 15A W U U TYPICAL PERFOR A CE CHARACTERISTICS A D BOARD PHOTO LTC1702 Efficiency EFFICIENCY (%) 100 VIN = 5V VOUT = 3.3V VOUT = 2.5V 90 VOUT = 1.6V 80 70 0 5 10 LOAD CURRENT (A) 15 1702 G01 1 2 E2 PWRGD1 E3 SD1 E8 GND + + + R8 10k VIN C9 180µF C8 C10 1µF 180µF VOUT1 1.6V, 1.8V, 2V OR 2.5V AT 15A E1 C11 180µF C12 180µF + R7 20k 2V 1% JP2 JP1 C13 820pF Q3 R6 40.2k 1.8V 1% D3 Q1 1.6V 1.8V 2.0V 2.5V JP1 OUT IN OUT OUT JP2 OUT OUT IN OUT JP3 R16 8.87k 2.5V 1% Q4 Q2 JP3 OUT OUT OUT IN VOUT1 JUMPER SELECT R5 10k 1.6V 1% R4 10k 1% R3 1.2k L1 1µH C23 1µF C14 680pF R9 47k C15 27pF C7 1µF R2 27k C5 1µF 12 11 10 9 8 7 6 5 4 3 2 1 FB1 SGND COMP1 FAULT PGOOD2 PGND SW2 TG2 BG2 BOOST2 IMAX2 VCC FB2 COMP2 RUN/SS2 LTC1702 RUN/SS1 FCB PGOOD1 IMAX1 SW1 TG1 BG1 BOOST1 PVCC 13 14 15 16 17 18 19 20 21 22 23 24 C17 1µF C16 1µF R10 27k D1 C2 330µF 22 BG2 21 TG2 20 SW2 19 PGND 18 PGOOD2 17 FAULT 16 RUN/SS2 4 5 6 7 8 9 TG1 SW1 IMAX1 PGOOD1 FCB RUN/SS1 13 VCC FB1 12 LTC1702CGN GN PACKAGE 24-LEAD NARROW PLASTIC SSOP 14 FB2 SGND 11 15 COMP2 23 BOOST2 3 BG1 COMP1 10 24 IMAX2 2 BOOST1 TOP VIEW 1 PVCC + C1 330µF C21 27pF + Figure 1. Dual 550kHz Synchronous 2-Phase 15A DC/DC Converter BURST JP4 CONT VIN C6 1µF D2 C4 10µF C22 3300pF C18 680pF Q8 Q6 C3 330µF R15 68k Q7 Q5 + D4 R13 4.99k 1% R12 15.8k 1% R11 1.6k L2 1µH C24 1µF + + R14 10k VIN + C26 C19 180µF 180µF + C25 180µF DC175 F01 E5 PWRGD2 E6 FAULT E7 SD2 E11 GND C20 1µF C27 180µF E4 VOUT2 3.3V/15A E9 GND E10 VIN U W R1 10Ω 5% PACKAGE A D SCHE ATIC DIAGRA SM W NOTES: UNLESS OTHERWISE SPECIFIED D1, D2: ON SEMICONDUCTOR MBR0520LT1 D3, D4: ON SEMICONDUCTOR MBRS340T3 Q1 TO Q8: FAIRCHILD FDS6670A DEMO MANUAL DC275 DC/DC CONVERTER DEMO MANUAL DC275 DC/DC CONVERTER PARTS LIST REFERENCE DESIGNATOR QUANTITY PART NUMBER DESCRIPTION VENDOR TELEPHONE 330µF 10V 10% Tantalum Capacitor Kemet (408) 986-0424 C1 to C3 3 T510X337K010AS C4 1 1206ZG106ZAT1A 10µF 10V Y5V Capacitor AVX (843) 946-0362 C5 to C8, C16, C17, C20, C23 C24 9 06036D105MAT1A 1µF 6V X5R Capacitor AVX (843) 946-0362 C9 to C12, C19, C25 to C27 8 EEFUEOG181R 180µF 4V SP Capacitor Panasonic (714) 373-7334 C13 1 06035C821MAT1A 820pF 50V X7R Capacitor AVX (843) 946-0362 C14, C18 2 06035C681MAT1A 680pF 50V X7R Capacitor AVX (843) 946-0362 C15, C21 2 06035A270MAT1A 27pF 50V NPO Capacitor AVX (843) 946-0362 C22 1 06035C332MAT1A 3300pF 50V X7R Capacitor AVX (843) 946-0362 D1, D2 2 MBR0520LT1 Schottky Diode ON Semiconductor (602) 244-6600 D3, D4 2 MBRS340T3 Schottky Diode ON Semiconductor (602) 244-6600 E1, E4, E8 to E11 6 2501-2 1-Pin Terminal Mill-Max (516) 922-6000 E2, E3, E5 to E7 5 2308-2 1-Pin Terminal Mill-Max (516) 922-6000 JP1 to JP3 3 3801S-02G2 0.100"CC 2-Pin Jumper Comm Con (626) 301-4200 JP4 1 3801S-03G2 0.100"CC 3-Pin Jumper Comm Con (626) 301-4200 JP1, JP4 2 CCIJ230-G 0.100"CC Shunt Comm Con (626) 301-4200 L1, L2 2 CEP125-1R0MC-H or ETQP6F1R0SSP 1µH 20A SMT Inductor Sumida Panasonic (847) 956-0667 (714) 373-7334 Q1 to Q8 8 FDS6670A SO-8 N-Channel MOSFET Fairchild (408) 822-2126 R1 1 CR16-100JM 10Ω 1/16W 5% Chip Resistor Tad (714) 255-9123 R2, R10 2 CR16-273JM 27k 1/16W 5% Chip Resistor Tad (714) 255-9123 R3 1 CR16-122JM 1.2k 1/16W 5% Chip Resistor Tad (714) 255-9123 R4, R5 2 CR16-1002FM 10k 1/16W 1% Chip Resistor Tad (714) 255-9123 R6 1 CR16-4022FM 40.2k 1/16W 1% Chip Resistor Tad (714) 255-9123 R7 1 CR16-2002FM 20k 1/16W 1% Chip Resistor Tad (714) 255-9123 R8, R14 2 CR16-103JM 10k 1/16W 5% Chip Resistor Tad (714) 255-9123 R9 1 CR16-473JM 47k 1/16W 5% Chip Resistor Tad (800) 508-1521 R11 1 CR16-162JM 1.6k 1/16W 5% Chip Resistor Tad (800) 508-1521 R12 1 CR16-1582FM 15.8k 1/16W 1% Chip Resistor Tad (800) 508-1521 R13 1 CR16-4991FM 4.99k 1/16W 1% Chip Resistor Tad (800) 508-1521 R15 1 CR16-683JM 68k 1/16W 5% Chip Resistor Tad (800) 508-1521 R16 1 CR16-8871FM 8.87k 1/16W 1% Chip Resistor Tad (800) 508-1521 U1 1 LTC1702CGN 24-Lead SSOP IC LTC (408) 432-1900 3 DEMO MANUAL DC275 DC/DC CONVERTER QUICK START GUIDE Refer to Figure 2 for proper measurement equipment setup and follow the procedure outlined below: 6. The SD1 and SD2 pins should be left floating for normal operation and tied to GND for shutdown. 1. Connect the input power to the VIN and GND terminals on the board using 12-gauge or heavier wire soldered to the terminals. The input voltage is limited to between 4.75V and 7V. 7. Connect a voltmeter across the VIN and GND terminals to measure input voltage. 8. Connect a voltmeter across the VOUT1 and GND terminals and another across the VOUT2 and GND terminals to measure the output voltages. 2. Connect an ammeter in series with the input supply to measure input current. 9. For applications where the minimum load current is greater than 1A, set jumper JP4 to the “Continuous” position. 3. Since this demo board operates from a low input voltage and supplies high output current, it is essential that the input supply voltage be well regulated. If the input power supply is equipped with remote sense lines, connect SENSE + to the VIN terminal and SENSE – to GND terminal on the board. 10. Set the desired output voltage (VOUT1) with jumpers JP1 to JP3, as shown in Table 1. 11. After all connections are made, turn on the power and verify that VOUT1 and VOUT2 are correct. 4. Connect either power resistors or an electronic load to the VOUT1, VOUT2 and GND terminals using 12-gauge or heavier wire, soldered to the terminals. Table 1 5. Connect an ammeter in series with each of the output loads to measure output currents. POSITION OUTPUT VOLTAGE No Jumper 1.6V JP1 1.8V JP2 2.0V JP3 2.5V A INPUT SUPPLY V PWRGD2 FAULT PWRGD1 GND VIN SD1 SD2 BURST CONT GND JP2 JP1 JP3 LOAD GND LOAD V V A VOUT2 DC275A-LTC1702CGN DUAL OUTPUT BUCK REGULATORS Figure 2. Proper Measurement Setup 4 A VOUT1 DEMO MANUAL DC275 DC/DC CONVERTER U OPERATIO The circuit in Figure 1 highlights the capabilities of the LTC1702. This design provides one fixed 3.3V output (VOUT2) and one output (VOUT1) that is jumper selectable from 1.6V to 2.5V. The LTC1702 is a voltage mode controller, designed to drive a pair of external N-channel MOSFETs using a fixed 550kHz switching frequency. The synchronous buck architecture automatically shifts to discontinuous operation and then to Burst ModeTM operation as the output load decreases, ensuring maximum efficiency over a wide range of load currents. This mode is recommended for load currents less than 1A and can be implemented on the demo board by moving jumper JP4 to the “Burst” position. Theory of Operation The LTC1702 has two independent switching regulators. For the sake of simplicity and to minimize repetition, only side “1” will be discussed. The divided output (VOUT1) is compared to the 0.8V reference. The difference voltage is multiplied by the error amplifier’s (FB) gain. The resulting error signal is then compared to an internally generated, fixed frequency sawtooth waveform by the PWM comparator, which generates a pulse width modulated signal. This PWM signal drives the external MOSFETs through TG1 and BG1. The output of this chopper circuit is then filtered by L1 and C9 to C12 to produce the desired DC output voltage. 2-Phase Operation The LTC1702 dual switching regulator controller also features the considerable benefits of 2-phase operation. The LTC1702 includes a single master clock that drives the two sides such that side 1 is 180° out of phase with side 2. This technique, known as 2-phase switching, has the effect of doubling the frequency of the switching pulses seen by the input capacitor and significantly reduces their RMS value. With 2-phase switching, the input capacitor is sized as required to support the larger of the two sides at maximum load current. As the load current increases on the lower current side, it tends to cancel, rather than add to, the RMS current seen by the input capacitor; thus no additional capacitance is needed. Capacitor Considerations The input capacitors are Kemet T510X337K010AS, 330µF, 10V tantalums. The input capacitors must be rated for the RMS input ripple. A good rule of thumb is that the input ripple current will be 50% of the output current. Since the LTC1702 uses 2-phase switching, the input bulk capacitors should be able to fully handle the RMS ripple current of just one load. As the load current increases on the other side, it tends to cancel, rather than to add to, the ripple current requirements for the input capacitors. For a continuous output current of 15A, the ripple current rating of the input capacitors should be 7.5A. The capacitors chosen are rated at 2.5A each, so three are adequate. Without the 2-phase operation, six capacitors would be required to handle two 15A loads. Output capacitors need to have a ripple current rating greater than the RMS value of the inductor ripple current. This is a function of the operating frequency and inductor value, as well as input and output voltages. Because the ripple current is relatively small, the controlling parameter is generally the capacitor’s ESR (equivalent series resistance). The maximum allowable ESR is equal to the maximum allowable peak-to-peak output ripple voltage divided by the peak-to-peak inductor ripple current. In general, if the ESR is low enough for the ripple voltage and transient requirements, the capacitors will have more than adequate ripple current capability. Inductor Selection Inductor selection is not extremely critical. The inductor used here was chosen for fairly low cost and ready availability. The main concerns in choosing an appropriate inductor are the inductance value required, the saturation current rating and the temperature rise. Most manufacturers specify a DC current rating that produces a temperature rise of 40°C. If a design will not see high ambient temperatures, a larger temperature rise can usually be tolerated. Another maximum current specification is related to core saturation. A manufacturer may specify that maximum rated current is the point at which inductance is down by 10% (some specify 25%). Since most core materials and structures will result in a gentle, Burst Mode is a trademark of Linear Technology Corporation. 5 DEMO MANUAL DC275 DC/DC CONVERTER U OPERATIO controlled roll off of inductance with DC bias, there is no magical point where the inductor is no longer useful. Look at what the inductance will be at the maximum load current expected and determine if the output ripple will remain within specified limits. If it will, the inductor will most likely work correctly. Ripple current is generally designed for between 10% and 40% of output current. conventional soft-start pin, enforcing a duty cycle limit proportional to the voltage at RUN/SS. An internal 4µA current source pull-up is connected to each RUN/SS pin, allowing a soft-start ramp to be generated with a single external capacitor (C7 for side 1 and C17 for side 2) to ground. Current Limit MOSFET Selection The main concern with FET selection in very low voltage applications is thermal management. At high current levels, power devices will get hot. The trick is to keep the temperature rise within acceptable limits. Most of the FETs’ power dissipation will be due to conduction losses. Therefore, by choosing a FET with a sufficiently low RDS(ON), the power dissipation, and therefore, the temperature rise, can be made arbitrarily low. The price paid for very low temperature rise is more expensive FETs. Switching losses are a concern only for the high side FET. The low side FET turns on and off into a forward-biased diode, so its transition losses are very small. The high side FET, in contrast, must provide all of the reverse recovery charge that the low side FETs body diode will demand. This can result in a significant amount of switching loss in this device. Although it may seem that a lower on-resistance FET is always desirable from an efficiency perspective, this is not necessarily true. A smaller device will have a lower gatecharge power requirement and will also exhibit faster switching transition times. The resulting reduction in AC losses may more than offset the increase in conduction losses. A smaller, higher on-resistance FET may prove the more efficient, as well as the lower cost solution. As the load current increases, gate-drive losses become less of a concern. At output currents on the order of 15A, lower resistance FETs will probably be better in terms of overall efficiency, but not necessarily the most cost effective choice. Each application will place a different value on a few points of efficiency. Shutdown/Soft-Start Each half or the LTC1702 has a RUN/SS pin. This pin performs two functions: when pulled to ground, each shuts down its half of the LTC1702, and each acts as a 6 The IMAX resistor, R2, sets the current limit by setting the maximum allowable voltage drop across the bottom MOSFET before the current limit circuit engages. The voltage across the bottom MOSFET is determined by its on-resistance and by the current flowing in the inductor, which is the same as the output current. To set the current limit, connect an RIMAX resistor from IMAX to GND. The value of RIMAX is calculated as follows: RIMAX = [(ILIM • RDS(ON)) + 100mV]/10µA ILIM should be chosen to be 150% of the maximum operating load current to account for MOSFET RDS(ON) variations with temperature. How to Measure Voltage Regulation and Efficiency When trying to measure load regulation or efficiency, voltage measurements should be made directly across the VOUT and GND terminals and should not be taken at the end of test leads at the load. Similarly, input voltage should be measured directly on the VIN and GND terminals of the LTC1702 demo board. Input and output current should be measured by placing an ammeter in series with the input supply and load. Refer to Figure 2 for the proper test equipment setup. Refer to page one for typical efficiency curves for VIN = 5V, VOUT = 3.3V, 2.5V and 1.8V, for IL = 1A to 15A. How to Measure Output Voltage Ripple In order to measure output voltage ripple, care must be taken to avoid a long ground lead on the oscilloscope probe. Therefore, a sturdy wire should be soldered on the output side of the GND terminal. The other end of the wire is looped around the ground side of the probe and should be kept as short as possible. The tip of the probe is touched directly to VOUT (see Figure 3). Bandwidth is generally DEMO MANUAL DC275 DC/DC CONVERTER U OPERATIO limited to 20MHz for ripple measurements. Also, if multiple pieces of line-powered test equipment are used, be sure to use isolation transformers on their power lines to prevent ground loops, which can cause erroneous results. Figures 4 and 5 show the output voltage ripple for the 3.3V and the 2.5V supplies for a 15A load. GND VOUT Transient Response The LTC1702 uses true 25MHz gain bandwidth op amps as the feedback amplifiers. This allows the use of an OPTI-LOOPTM compensation scheme that can precisely tailor the loop response. The high gain-bandwidth product allows the loop to be crossed over beyond 50kHz while maintaining good stability, and significantly enhances load transient response. Figures 6 and 7 show the transient response of the 3.3V and the 2.5V output supplies for a 0A to 10A load step. For more information about loop compensation and stability analysis, consult the LTC1702 data sheet. OPTI-LOOP is a trademark of Linear Technology Corporation. Figure 3. Measuring Output Voltage Ripple Figure 4. 3.3V Output Voltage Ripple, IL = 15A Figure 6. 3.3V Transient Response, IL = 0A to 10A Figure 5. 2.5V Output Voltage Ripple, IL = 15A Figure 7. 2.5V Transient Response, IL = 0A to 10A 7 DEMO MANUAL DC275 DC/DC CONVERTER U OPERATIO Heat Dissipation Issues Since each side of the LTC1702 demo board can supply 15A of continuous load current, care must be taken not to exceed the maximum junction temperature for the power MOSFETs. A few possibilities for dissipating the power are to use heat sinks and/or forced air cooling. Another possibility is to use the PC board as a heat sink. On the LTC1702 demo board, power MOSFETS Q1 to Q8 are surrounded by ground and power planes on both sides of the PC board. Also, there is metal on the inner layers directly underneath the power MOSFETs. This helps in spreading the heat and improves the power dissipation capability of the PCB. Layout Guidelines Since the LTC1702 is a switching regulator, a good layout is essential for good load regulation and minimizing radiated/conducted noise. If you want a layout that is guaranteed to work, copy the LTC1702 Gerber files provided with this demo board; otherwise, be sure to follow the layout guidelines below: 1. The inductor L1, MOSFETs Q1 to Q4 and the Schottky diode (D3) should be placed as close as possible to each other; similarly, L2, Q5 to Q8 and D4 should be placed as close together as possible. This junction forms the switch node and should be kept as small as possible to minimize radiated emissions. It must also be large enough to carry the full rated output current. 2. The SW1 and the SW2 pins should be connected directly to the respective switch nodes with a short trace. 3. C4 (10µF) should be as close as possible to Pin 13 on the LTC1702. 8 4. C5 (1µF) should be as close as possible to Pin 1 on the LTC1702. 5. R2 should be connected directly to the sources of Q3 and Q4. 6. R10 should be connected directly to the sources of Q7 and Q8. 7. Keep the trace from the FB1 pin to the junction of R4 and R5 short and use a long trace from the top of resistor R4 to the output terminal, rather than vice versa. 8. Keep the trace from the FB2 pin to the junction of R12 and R13 short and use a long trace from the top of resistor R12 to the output terminal, rather than vice versa. 9. The sources of the bottom MOSFETs Q3, Q4, Q7 and Q8 should be tied back to the ground of input capacitors C1 to C3 by means of a wide trace, not by the ground plane. 10. The grounds of the output capacitors C19–C20, C25–C27 and C8–C12 should be tied directly to the input capacitor’s ground by means of a wide trace or by the ground plane. 11. The grounds of the feedback resistors, soft-start capacitors and C4 should be referenced to the chip SGND pin, which is then tied to the input bulk capacitors’ grounds. 12. PGND, Pin 19, should connect directly to the ground plane. DEMO MANUAL DC275 DC/DC CONVERTER U W PCB LAYOUT A D FIL Top Silkscreen Top Solder Mask Top Pastemask 9 DEMO MANUAL DC275 DC/DC CONVERTER U W PCB LAYOUT A D FIL 10 Layer 1, Top Layer Layer 2, VIN Plane Layer 3, GND Plane Layer 4, Bottom Layer DEMO MANUAL DC275 DC/DC CONVERTER U W PCB LAYOUT A D FIL Bottom Silkscreen Bottom Solder Mask Bottom Pastemask Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 DEMO MANUAL DC275 DC/DC CONVERTER U PC FAB DRAWI G 2.50" D A F D A E F E C A A C B 2.10" E E A NOTES: UNLESS OTHERWISE SPECIFIED 1. MATERIAL: FR4 OR EQUIVALENT EPOXY, 2 OZ COPPER CLAD, THICKNESS 0.062 ±0.006 TOTAL OF 4 LAYERS 2. FINISH: ALL PLATED HOLES 0.001 MIN/0.0015 MAX COPPER PLATE, ELECTRODEPOSITED TIN-LEAD COMPOSITION BEFORE REFLOW, SOLDER MASK OVER BARE COPPER (SMOBC) 3. SOLDER MASK: BOTH SIDES USING LPI OR EQUIVALENT 4. SILKSCREEN: USING WHITE NONCONDUCTIVE EPOXY INK 5. UNUSED SMD COMPONENTS SHOULD BE FREE OF SOLDER 6. FILL UP ALL VIAS WITH SOLDER 7. SCORING A 0.017 E E A F A F D 12 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com SYMBOL DIAMETER NUMBER OF HOLES A 0.015 70 B 0.035 5 C 0.064 5 D 0.070 3 E 0.094 6 F 0.125 4 TOTAL HOLES 93 dc275f LT/TP 0100 500 • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 2000