TD230 ELECTRONIC CIRCUIT BREAKER . .. . . .. . TWO N-CHANNEL MOSFETs CONTROL AND DUAL INDEPENDANT CURRENT SUPERVISION FOR OVER CURRENT PROTECTION DUAL SUPPLY OPERATION FROM +3/-5 TO +18/-18V OPERATING VOLTAGE STEP-UPCONVERTER : VCC +13.5V OUTPUT VOLTAGE A DJ UST A BL E P ROT ECTI O N MODE (CTRIP1/2) INHIBIT FUNCTION SHUTDOWN OUTPUT STATUS FEW EXTERNAL COMPONENTS N DIP16 (Plastic Package) D SO16 (Plastic Micropackage) ORDER CODES Part Number DESCRIPTION The TD230 is designed to control two N-channel MOSFETs used as power switches in circuit breaking applications. Its current supervision and immediate action on the switches ensure high security for the boards and the supplies thus protected against short-circuit or over current. In case of short-circuit or over current detection,the TD230 immediately switches off the corresponding MOSFET, thus disconnecting the board from the supply. After several automatic restart attempts, a definitive shutdown of the circuit is done if the shortcircuit or over current persists over an externally adjustable time, until the TD230 is reset by temporary INHIBIT signal or temporary switching off of the power supply (hot disconnection/reconnection). If the board is disconnected from the positive supply by the TD230 it will automatically be disjoncted from the negative supply too. TD230 integrates an induction step-up converter that provides 13.5V above the positive rail to drive the high side MOSFET. October 1998 Package Temperature Range o o -40 C, +125 C TD230I N D • • PIN CONNECTIONS 1 16 REF1 LBOOS T 2 15 GC1 CBOOST 3 14 S ENSP OS CGND 4 13 INHIBIT PM1 5 12 SHUTDOWN GND 6 11 SENS N PM2 7 10 GC2 NVcc 8 9 REF2 P Vcc TD230 BLOCK DIAGRAM PVcc LBOOST STEP -UP o sc CBOOS T OS C GND R EF1 VSP 1 IP1 S W1 INHIBIT GC1 PVcc P M1 S E NSP IP 2 VSP 2 S HUTDOWN VSP 3 IP3 VSN3 IN3 G ND P M2 IN2 S ENSN VSN2 P Vcc P Vcc IN1 S W2 GC2 VSN1 R EF2 NVcc ABSOLUTE MAXIMUM RATINGS Symbol + Parameter Value Unit Positive Supply Voltage +22 V VCC - Negative Supply Voltage -22 V Inhibit Input Voltage 7 V Shutdown Input Voltage 7 V PM1/PM2 Input Voltage 7 VCC Tj Operating Junction Temperature V -40 to 150 o C Tamb Operating Ambient Temperature -40 to 125 o Tstg Storage Temperature Range -65 to 150 o C C OPERATING CONDITIONS Symbol VCC Parameter Supply Voltage Value Unit +/-18 V Value Unit 1 µF INSTRUCTIONS FOR USE Symbol Cbypass 2/15 Parameter Bypass Capacitor (each supply) TD230 ELECTRICAL CHARACTERISTICS VCC = +/-5V, Tamb = 25oC, Lboost = 220µH, Cboost = 100nF (unless otherwise specified) Symbol Parameter Min. Typ. Max. Unit SUPPLY VCC+ Positive Supply Voltage 2.7 18 V VCC + ICC Negative Supply Voltage -18 -4.5 V ICC - Positive Supply Current Charge Pump Inactivated 1.8 3 mA Charge Pump Activated Lboost = 220µH, C boost = 100nF 2.3 4 mA Negative Supply Current Charge Pump Activated/Inactivated -0.7 -1.5 mA LOGIC INPUT (INHIBIT) VIH High Input Voltage 2 6 V VIL Low Input Voltage 0 0.8 V 2 µs 0.8 V tp Propagation Delay GC1/2 (without load) 0.5 LOGIC OUTPUT (SHUTDOWN-open drain) VOL Low Output Voltage (2mA) IOH High Output Current (6V) 1 Shutdown Response Time (sens P/N shutdown without load) 8 15 VCC +13.4 VCC +15 V ts nA µs STEP-UP CONVERTER + + + Vboost Step-Up Output Voltage tvboost Rise Time for Vboost (10 to 90%) 250 800 µs Output Ripple Voltge 0.15 0.6 V 63 71 mV 2 3 µs 300 ns Vrip VCC +10 INPUT COMPARATORS Vi Threshold (PVCC - Ref1, NVCC - Ref2) tre Response Time (GC1/2 - without load) ti Inhibition Time (without load) 56 VOLTAGE SOURCES V VSP2 VSN2 Threshold Sense Pos/Neg 0.7 VCC+/- 0.75 VCC+/- 0.8 VCC+/- VSP3 VSN3 Threshold Protection Mode 1.10 1.20 1.30 10 15 20 µA V CURRENT SOURCES IP1, IN1 Soft Start Current Sources IP2, IN2 Protection Mode Current Sources (loading Ctrip 1/2) IP3, IN3 Protection Mode Current Sources (discharging Ctrip 1/2) 3 4 5 µA 0.6 1 1.4 µA 90 200 SWITCHES R on On-Resistance of the switches SW1/SW2 Ω 3/15 TD230 Figure 1 : DUAL ELECTRONIC CIRCUIT BREAKER APPLICATION RS1 Vcc+ REF1 P Vcc 1 LBOOST 16 LBO OS T 2 15 CBO OS T S ENS P OS CGND INHIBIT 3 CBOOS T NMos GC1 4 5 P M1 S HUTDO WN 6 GND SE NS N CTRIP1 CTRIP2 CS S1 to BOARD 14 13 CONTR OL 12 from BOARD 11 NMOS G ND 7 P M2 GC2 10 CS S2 REF2 NVcc 8 9 VccRS2 Figure 2 : SINGLE ELECTRONIC CIRCUIT BREAKER APPLICATION RS 1 Vcc+ P Vcc LBOO ST 16 LBOO S T 2 CBOOS T REF1 1 3 15 CBOO S T S ENS P OS CGND INHIBIT 4 13 P M1 S HUTDOWN 12 6 GND S ENS N 11 7 P M2 GND NVcc 8 GC2 REF2 CS S 1 14 5 CTRIP1 4/15 NMos GC1 10 9 CONTROL to BOARD TD230 TIMING DIAGRAMS P owe r E vents Sta tus Norma l Curre nt S hort Circuit Norma l Curre nt Short Circuit Norma l Function Current Limitation Norma l Function Curre nt Limitation Inhibit OFF ON Circuit OFF Norma l Curre nt HI LO Norma l Curre nt Norma l Function OFF Norma l Func tion PVc c- Vre f (=Vrs) Vi S e ns P P Vc c -ε GC1-S e ns P = Vgs 13.4V ~5V P M1=Vctrip1 ts s # PVc c. Css Ip1 PVc c Vsp3 S hutdown tpm1 # (Ctrip1.Vsp3) IP2-IP3 ts HiZ Inhibit tp TTL PVcc -Vref (=Vrs) Vi G C1 -Se ns P = Vgs 13.4V ~5V PVcc -Vref (=Vrs) t < ti ti tre toff # Ro n.Cs s Vi G C1 -Se ns P = Vgs 1 3 .4V ~5V 5/15 TD230 APPLICATION NOTE ELECTRONIC CIRCUIT BREAKER by R. LIOU INTRODUCTION Over current and short circuit protection is a constant concern for today’s engineers. More and more applications in different segments (Telecom, Automotive, Industrial, Computer...) require always improved reliability after delivery : maintenance costs are an ever more worrying source of expenses and customers’ dissatisfaction. Alternatives for short circuit or over current protections are the fuses and the PTC (Positive Temperature Coefficient) resistors. The first are a cheap but destructive solution ; the second are tied to a time constant due to self heating which is often incompatible with the host equipment’s requirements. In both cases, a coil can be added for an efficient limitation of current surges, to the detriment of weight and volume. None of these solutions is fully satisfactory for a reliable, immediate and non destructible short circuit and over current protection. 1. ELECTRONIC CIRCUIT BREAKER The electronic circuit breaker TD230 is the convenient solution for any industrial who wants at the same time : • immediate, efficient and resettable protection for his equipment • versatility regarding different applications • easy and quick design-in • low component count • low cost The electronic circuit breaker TD230 is to be used with a minimal amount of external and low cost components to drive one or two N-channel MOSFETs (in respectively single or dual supply applications) used as power switches between the DC power supplies and the equipments to be protected. The TD230 immediately reacts (3µs max. without load) whenever an over current is detected by switching off the corresponding MOSFET. Several automatic restart attempts are made unless the fault persists over an externally adjustable amount 6/15 of time after which the power MOSFET is definitively switched off, waiting for a reset. If the fault is detected on the positive supply, the definitive shutdown will also disconnect the negative power supply and set a warning low level on the Shutdown pin. If the fault is detected on the negative supply, the definitive shutdown will disconnect only the negative power supply,and let the positive part of the circuit undisturbed. The whole system can be reset in three ways : • by switching off the power supplies • by unplugging and re-plugging the card (live insertion) • by setting the INHIBIT pin active during a short time (allowing remote reset) 2. HOW TO USE THE TD230 ? The typical configuration of the TD230 - Electronic Circuit Breaker - in a dual supply topology is shown in figure 1. In this configuration, both NMOS 1/2 are used as power switches which connect the equipments to the power supplies, thus ensuring low voltage drop through the ON-resistances (Rdson) of NMOS 1/2. 2.1. Current Limitation When an over current condition (IOC ) is detected through the low ohmic shunt resistors RS 1/2 as given under equation (i) : • VRS 1/2 = IOC x RS > 63mV typ. (i) the gate of the corresponding MOSFET 1/2 is discharged immediately, thus disconnecting the board/equipment from the power supply. Note that the over current condition is given by the constant product IOC x RS = 63mV, which means that the IOC limit is directly given by the choice of the shunt resistors RS1/2 values. The TD230 automatically makes restart attempts by slowly recharging the gate of the MOSFET 1/2 with a 15µA typ. current source ensuring thus slow ramp with the typical time constant before reconduction shown in equation (ii) : • tON = CISS x VTH / 15µA (ii) TD230 where CISS is the input capacitance of the power MOSFET1/2 and VTH, the threshold voltage of the MOSFET (typically 5V). This reconduction time can be extended with an external soft start capacitor CSS1/2 as shown in figure 1 CISS will therefore simply be replaced by CISS + CSS 1/2. Figure 1 : Dual Electronic Circuit Breaker Application RS 1 Vcc+ R EF1 P Vcc LBOO S T 1 16 2 CBO OS T 15 S E NSP CB OOS T 3 O S C GND INHIBIT 4 5 P M1 S HUTDOWN 6 GND S ENSN CTRIP 1 CTRIP 2 GND NMos GC1 LBOOS T CS S 1 to BOARD 14 13 C ONTRO L 12 from BO ARD 11 NMOS 7 P M2 GC 2 10 CS S 2 NVcc R EF2 8 9 VccRS 2 If the fault (over current condition) still remains after the reconductionstate of the MOSFET1/2 has been reached, the current through NMOS1/2 will overpass the limitation given by equation (i), and the NMOS 1/2 will immediately be switched off again. Figure 2 shows the current limitation which is operated on every restart attempt. Figure 2 : TD230 as Current Limitor Trace A represents the Gate-Source Voltage of the Power Mosfet (0 to 13,4V). Trace B represents the voltage across the Sense Resistor (68mΩ) in direct relation with the current through it (0 to ~1A). Note that the first current peak which is due to an over current is limited only by the reaction time of the TD230. This off time is tied to the value of the external soft start capacitor CSS 1/2 by equation (iii) : • tOFF = RDSON x CSS (iii) While in current limitation mode, the NMOS1/2 dissipates low power due to the fact that the ON/OFF cycle time rate is very low. Note that the higher the value of CSS1/2 are, the more the NMOS1/2 will stay in linear mode during current limitation. Note that at Power ON, or in the case of live insertion, the inrush current is automatically limited thanks to the slow gate charge of the MOSFET which switches ON softly due to the time constant given in equation (ii). 2.2. Fault Time Limitation The repetitive switching off of the MOSFET will come to an end under two conditions : • either the fault has disappeared, and the current through the shunt resistors RS 1/2 has come back to its nominal value : the system keeps running normally. External line defaults (lightning, line breakage, etc...) are usual causes for such temporary over currents. • either the repetitive switching off has lasted over an externally adjustable time and the TD230 has definitively switched off the corresponding NMOS : the system waits to be reset. Equipment faults (component short circuit, over heat, etc ...) are usual causes for lasting over currents. This fault time supervision is done by the comparison of the output voltage to 75% of the nominal supply voltage. As soon as the output voltage is detected under 0.75xVcc(+/-), the corresponding external capacitors CTRIP1/2 is charged by a fixed current source IP/N2 - IP/N3 (3µA). When the voltage across CTRIP1/2 reaches 1.20V, the corresponding NMOS is definitively switched off and the SHUTDOWN pin is active low. 7/15 TD230 To avoid cumulative charging of the protection capacitors CTRIP 1/2 in case of successive overcurrent conditions, the capacitors CTRIP 1/2 are conFigure 3 : Figure 4 : Step Up Converter External Components Fault Time Limitation Rs e ns e Lboos t Sense S tep Up Cboos t MOS Driver TD230 The principle of this inductive step-up converter is to pump charges in the tank capacitor CBOOST following the equation (v) : stantly discharged by another fixed current source IP/N3 which value is a fourth of IP/N2 (1µA). Trace 1 represents the CBOOST Voltage (0 to 5+13,4 = 18,4V) Trace 2 represents the CTRIP1 Voltage. The value of the capacitors CTRIP 1/2 should be chosen in relation with the required protection time as indicated in equation (iv) : • CTRIP1/2 = (IP/N2 - IP/N3) x tPROTECT1/2 / VSPN/3(iv) where tPROTECT 1/2 is the time defined by the user before a definitive resettable shutdown of MOSFET 1/2. Equation (iv) can be translated to : • CTRIP 1/2 = tPROTE CT 1/2 x 3µA / 1.20V (iv) Note that the positive power supply disjonction leads to the negative power supply disjonction, whereas the opposite is not true. 2.3. Step-Up Converter To ensure proper voltage on the gate of the positive supply NMOS1 (VGS = 13.4V typ), the TD230 integrates a step-up converter which is to be boosted with two small low cost external components : an inductor LBOOST and a capacitor CBOOST, as shown in figure 4. 8/15 Figure 5 : Internal Step Up Schematic Lboos t Cboos t Os c Re gulation TD230 • V(CBOOST) = VCC+ + 13.4V typ (v) Charges are pumped by means of an oscillator commanded switch, and stored in the CBOOST tank capacitor through a diode as shown on figure 5. Wh en th e vo lt a ge across CBOOST reaches VCC++13.4V typ, the oscillator is stopped. This creates a ripple voltage with an amplitude of 0.2V. Note that the min and max values of V(CBOOST) comprised between VCC+ +10V and VCC+ +15V already take the ripple voltage into account. TD230 Proper operation of this step-up converter is guaranteed at as low as 2.7V with a rise time (0 to 90% of V(CBOOST)) in the range of 700µs at 2.7V which Figure 6 : Step Up Converter Rise Time 2.4. Single Supply Breaker Application The TD230 is perfectly suited to fit in single supplied applications (ex 0-5V), and can drive only one power MOSFET used as high side power switch. Figure 7 shows how TD230 can be used as a single circuit breaker with the same performances. Figure 7 : Single Electronic Circuit Breaker Application RS1 Vcc+ P Vcc LBOOS T 16 LBOOS T 2 CBOOS T R EF 1 1 3 4 GC 1 CBO OS T S ENS P OS C GND INHIBIT 13 P M1 S HUTDOWN 12 6 GND S ENS N 11 7 P M2 GC 2 10 Table (a) : Recommended values for Cboost and Lboost VCC+ V 2.7 5 10 12 14 18 Cboost nF 47 100 100 100 220 220 220 220 Lboost µH 68 Ipk mA 60 220 470 35 33 470 680 1000 39 34 31 Vrip mV 190 100 120 220 100 150 150 200 Icc mA 5 Cby µF >1 2.5 2.2 1 1 2.2 2.4 2.7 1 1 1 GND NVcc 8 R EF 2 CSS 1 to BOARD 14 5 CTRIP1 is the worst case. At 5V, the rise time of V(Cboost) is 250µs typ. The CBOOST voltage wave form at power ON under 5V supply voltage is shown on figure 6. Trace 1 represents the power supply voltage (0 to 5V). Trace 2 represents the CBOOST Voltage at power ON (0 to 5+13,4 = 18,4V). Table (a) summerizes the recommended values of the CBOOST and LBOOST to ensure optimized gate charge and low ripple voltage with their corresponding maximum current surge (IPK) and nominal consumption (I CC) of the TD230 for the most common power supply values. For each power supply value is also given the recommended value of a bypass capacitor (CBY) on the power supplies. Note that both CBOOST and LBOOST are available in surface mount packages. NMos 15 C ONTR OL 9 In this case, the external components consist in : one boost inductor, one sense resistor, three capacitors, and one power MOSFET. 2.5. Typical Telecom Line Cards Protection Application One of the typical applications where the TD230 can display all its technical advantages is in an exchange Telecom Cards protection. Sometimes fifty cards or more are to be supplied with the same power supply (+/-5V, 1kW), and a decentralized protection is needed because one card may be faulty, but should not penalize the others with unadapted protection system. The risk of complete breakdown of the system must be eradicated. In this application the two above described over current causes (external line perturbation or internal component fault) are likely to happen. In the first case, the current limitation on each card will ensure undammagingon-board conditions, and in the second case, the faulty card will be disjoncted from the power supply until reset. Figure 8 shows a typical telecom application with decentralized protection. In this application, the positive power supply serves the logic control and analog signals whereas the negative power supply is dedicated to the analog. 9/15 TD230 Figure 8 : Decentralized Protection Vcc+ P owe r S upply TD230 TD230 TD230 BOARD1 BOARD2 BOARD3 TD230 GND BOARDN Vcc- Therefore, when a fault appears onthe positive rail, the definitive shutdown of the positive NMOS will lead to the shutdown of the negative NMOS, but when a fault appears on the negative rail, the definitive shutdown of the negativeNMOS will have no effect on the positive NMOS. Several possibilities are offered to reset the whole system when it has been led to definitive shutdown : • the card can be unplugged and plugged back (live insertion) • the INHIBIT pin can be set to active state during a short time (100µs typ or more) in the case of remote control facilities • CTRIP1 = 10µF • CTRIP2 = 10µF • RS1 = 68mΩ • RS2 = 68mΩ • CSS1 = 1nF • CSS2 = 1nF • Positive Bypass = 4.7µF (plastic) • Negative Bypass = 4.7µF (plastic) The evaluation board is available and allows to test the performances of the TD230. The layout and schematic of this evaluation board are given on figures 9A-9B-9C. 3. PERFORMANCES AND EVALUATION All the curves shown in this application note have been realized with the TD230 Evaluation Board. The external conditions and components were as listed hereafter : • Vcc+ = 5V • Vcc- = -5V • Suppliable output short circuit current = 5A • IC = TD230 • MOSFET 1 = BUZ71 • MOSFET 2 = BUZ71 • LBOOST = 220µH • CBOOST = 100nF For proper use of the TD230 as a reliable protection device, a few precautions must be taken : 1. Proper bypass capacitors must be connected as close as possible to the power pins of the TD230 (PVcc, NVcc, GND). Some recommended values are given in table (a). 2. The OSCGND pin must be tied to the GND pin externally (printed board) to ensure proper step-up converter reference. If not, the step-up converter will not start. 3. TheINHIBIT pin is a CMOS/TTL compatible input which should therefore not be left unconnected. The absolute maximum rating of this input is 7V. It should be tied to the TTL compatible output of an 10/15 4. CAUTIONS TD230 Figure 9A : PCB (not to scale) Figure 9B : Silkscreen Figure 9C : Schematic eventual control block, or, if it should not be used, tied to the GND pin. 4. Th e SHUTDOWN pin is an open drain CMOS/TTL compatible output which should be tied to the TTL compatible input of an eventual control block. The absolute maximum rating of this output is 7V. In the case of a visual alarm, a LED is likely to be tied to the positive power supply which can be destructive for the Shutdown output if the power supply is over 7V. An easy way to eliminate this is to add a 6V zener diode between the Shutdown output and the Ground as shown on figure 10. 5. The time constant of the protection mode (given by the charge of CTRIP1/2 capacitors) must be greater than the time constant of the restart attempts (given by the charge of the CSS 1/2 soft start capacitors). This condition can be described as follows : • VSP1/2 x CTRIP1/2 / IP/N2 > VTH1/2 x (CSS1/2+CISS1/2) / IP/N1 11/15 TD230 Figure 10 : Visual Alarm-Shutdown Vcc + TD230 which, in most cases are, are not worrying. But in some very demanding applications, it is necessary to remove this noise. A good way to eliminate such peaks is to add a resistor connected in series with the inductance and an electrolytic capacitor between the common point of resistor and inductance, and ground of the Step-Up Converter as shown on figure 11. S hutdown Figure 11 : Step Up Noise Reduction Rs e ns e Table (b) : Protection Mode Time Constants C TRIP1/2 Time Constant Range for Protection Mode - Shutdown - 22nF #10ms 220nF #100ms 2.2µF #1s 22µF #10s 5. ENHANCEMENTS The performances of TD230 are well adapted to most of the circuit breaking applications in many differents industry segments (Telecom, Automotive, Industrial, Computer etc...), but in the case of very demanding environment, or outstanding features, the few following advices may be helpful. 5.1. Step-Up Noise Reduction The inductive step-up converter inevitably generates current peaks in the output of the power switch 12/15 C TD230 Cboost where CISS1/2 , CSS1/2, VTH1/2 , IP/N1 are respectively the input capacitance, the soft start capacitor, the threshold voltage and the internal gate current sources of NMOS1/2 ; and where VSP1/2 , CTRIP1/2 , IP/N2 are respectively the voltage source, current source and external capacitor of the protection mode pins PM1/2. Considering the typical values of VSP1/2, IP/N2, IP/N1, and the fact that classical power MOSFETs have a threshold voltage around 5V, this condition can be translated to inequation (vi) : • CTRIP1/2 > 0.8 x (CSS1/2 + CISS ) (vi) If CISS = 1nF and CSS1/2 = 4.7nF, CTRIP1/2 should be superior to 4.56nF. Table (b) summerizes Protection Mode Time Constants corresponding to different CTRIP1/2 values. Lboost R The resistor’s voltage drop will be due to the product of the average consumption current with the resistor’s value and the inductive current peaks will be totally absorbed by the capacitor. With a 100Ω resistor, the voltage drop is negligible and the attenuationgood with a 4.7µF as shown on figure 12. Figure 12 : Step Up Noise Reduction TD230 Trace A represents the ripple voltage on CBOOST (200mV width). Trace B represents the voltage perturbation due to the Step-Up converter on the output (source of the power Mosfet = Board power supply). Traces 1 and 2 represent the same, but improved thanks to the Step-Up Noise reduction RC. 5.2. Precision Enhancement If the system needs accurate current limitation in an environment subject to very wide temperature Figure 13 : Wide Temperature Variations Rs e ns e variations, a good way to compensate fluctuations due to temperature variations is to use a CTN as described in figure 13. 5.3. Temporisation In some cases, it can be useful to let short current peaks pass without reaction of the breaker, though these are of higher value than the fixedcurrent limit. This enables the Electronic Circuit Breaker to behave as a thermal disjonctor. This behaviour can easily be given by adding an RC constant as shown on figure 14. Figure 14 : Temporisation Rse nse R R TD230 CTN R C TD230 13/15 TD230 PACKAGE MECHANICAL DATA 16 PINS - PLASTIC DIP Dimensions a1 B b b1 D E e e3 F i L Z 14/15 Min. 0.51 0.77 Millimeters Typ. Max. 1.65 0.5 0.25 Min. 0.020 0.030 Inches Typ. Max. 0.065 0.020 0.010 20 8.5 2.54 17.78 0.787 0.335 0.100 0.700 7.1 5.1 3.3 0.280 0.201 0.130 1.27 0.050 TD230 PACKAGE MECHANICAL DATA 16 PINS - PLASTIC MICROPACKAGE (SO) Dimensions A a1 a2 b b1 C c1 D E e e3 F G L M S Min. Millimeters Typ. 0.1 0.35 0.19 Max. 1.75 0.2 1.6 0.46 0.25 Min. Inches Typ. 0.004 0.014 0.007 0.5 Max. 0.069 0.008 0.063 0.018 0.010 0.020 o 45 (typ.) 9.8 5.8 10 6.2 0.386 0.228 1.27 8.89 3.8 4.6 0.5 0.394 0.244 0.050 0.350 4.0 5.3 1.27 0.62 0.150 0.181 0.020 0.157 0.209 0.050 0.024 o 8 (max.) Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such info rmation nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publi cation are subject to change without notice. This publ ication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for useas critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a trademark of STMicroelectronics 1998 STMicroelectronics – Printed in Italy – All Rights Reserved STMicroelectronics GROUP OF COMPANIES Australia - Brazil - Canada - China - France - Germany - Italy - Japan - Korea - Malaysia - Malta - Mexico - Morocco The Netherlands - Singapore - Spain - Sweden - Switzerland - Taiwan - Thailand - United Kingdom - U.S.A. http://www.st.com 15/15