Electrical Specifications Subject to Change LT3799 Offline Isolated Flyback LED Controller with Active PFC FEATURES DESCRIPTION Isolated PFC LED Driver with Minimum Number of External Components n TRIAC Dimmable n V and V IN OUT Limited Only by External Components n Active Power Factor Correction (Typical PFC > 0.97) n Low Harmonic Content n No Opto-Coupler Required n Accurate Regulated LED Current (±5% Typical) n Open LED and Shorted LED Protection n Thermally Enhanced 16-lead MSOP Package The LT®3799 is an isolated flyback controller with power factor correction specifically designed for driving LEDs. The controller operates using critical conduction mode allowing the use of a small transformer. Using a novel current sensing scheme, the controller is able to deliver a well regulated current to the secondary side without using an opto-coupler. A strong gate driver is included to drive an external high voltage MOSFET. Utilizing an onboard multiplier, the LT3799 typically achieves power factors of 0.97. The FAULT pin provides notification of open and short LED conditions. n APPLICATIONS Offline 4W to 100W+ LED Applications High DC VIN LED Applications n n L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patents pending. The LT3799 uses a micropower hysteretic start-up to efficiently operate at offline input voltages, with a third winding to provide power to the part. An internal LDO provides a well regulated supply for the part’s internal circuitry and gate driver. TYPICAL APPLICATION TRIAC Dimmable 20W LED Driver LED Current vs Input Voltage 1.20 100k 0.22µF 0.1µF 499k 200Ω 1.15 20Ω 100k 4.7pF 10µF 499k 4:1:1 1.05 2k VIN DCM VIN_SENSE FB VREF 100k 32.4k 40.2k 100k NTC 16.2k FAULT CTRL3 GATE CTRL2 SENSE CTRL1 VINTVCC GND 0.1µF COMP– 1.00 0.95 0.90 560µF ×2 0.85 20W LED POWER 20Ω 0.80 90 100 110 120 130 VIN (VAC) 140 150 3799 TA01b 0.05Ω 4.7µF FAULT CT COMP+ 1A 100k 4.99k LT3799 6.34k 1.10 ILED (A) 90V TO 150V AC 2.2nF 3799 TA01a 0.1µF 3799p 1 LT3799 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) VIN, FAULT..................................................................32V GATE, INTVCC............................................................18V CTRL1, CTRL2, CTRL3, VIN_SENSE, COMP–.................4V FB, CT, VREF, COMP+,....................................................3V SENSE.......................................................................0.4V DCM........................................................................±3mA Maximum Junction Temperature........................... 125°C Operating Temperature Range (Note 2) LT3799E............................................. –40°C to 125°C LT3799I.............................................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C TOP VIEW CTRL1 CTRL2 CTRL3 VREF FAULT CT COMP+ COMP– 1 2 3 4 5 6 7 8 17 GND 16 15 14 13 12 11 10 9 VIN_SENSE SENSE GATE INTVCC NC VIN DCM FB MSE PACKAGE 16-LEAD PLASTIC MSOP θJA = 50°C/W, θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3799EMSE#PBF LT3799EMSE#TRPBF 3799 16-Lead Plastic MSOPE –40°C to 125°C LT3799IMSE#PBF LT3799IMSE#TRPBF 3799 16-Lead Plastic MSOPE –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS VIN Turn-On Voltage 22.2 23 24.2 V VIN Turn-Off Voltage 11.8 12.3 13.0 V VIN Hysteresis VTURNON – VTURNOFF 10.7 V VIN Shunt Regulator Voltage I = 1mA 25.0 V VIN Shunt Regulator Current Limit 15 mA VIN Quiescent Current Before Turn-On After Turn-On 55 65 70 75 µA µA INTVCC Quiescent Current Before Turn-On After Turn-On 12 1.5 16 1.2 20.0 2.6 µA mA VIN_SENSE Threshold Turn-Off 30 65 VIN_SENSE Linear Range 0 2 1.98 mV V 2.02 2.02 V V VREF Voltage 0µA Load 200µA Load Error Amplifier Voltage Gain ∆VCOMP+/∆VCOMP–, CTRL1 = 1V, CTRL2 = 2V, CTRL3 = 2V 100 V/V Error Amplifier Transconductance ∆I = 5µA 50 µmhos l l 1.975 1.9555 90 1.3 3799p 2 LT3799 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, unless otherwise noted. PARAMETER CONDITIONS MIN FB Pin Bias Current (Note 3), FB = 1V CTRL1/CTRL2/CTRL3 Pin Bias Current CTRL/CTRL2/CTRL3 = 1V SENSE Current Limit Threshold 96 TYP MAX UNITS 100 600 nA ±30 nA 106 mV 100 SENSE Input Bias Current Current Out of Pin, SENSE = 0V 15 µA Current Loop Voltage Gain ∆VCTRL /∆VSENSE, 1000pF Cap from COMP+ to COMP– 21 V/V CT Pin Charge Current 10 µA CT Pin Discharge Current 200 nA mV CT Pin Low Threshold Falling Threshold 240 CT Pin High Threshold Rising Threshold 1.25 V 100 mV CT Pin Low Hysteresis FB Pin High Threshold 1.22 1.25 1.29 V DCM Current Turn-On Threshold Current Out of Pin 45 µA Maximum Oscillator Frequency COMP+ = 1.2V, V 300 kHz Minimum Oscillator Frequency COMP+ = 0V, V 25 kHz 20 kHz IN_SENSE = 1V IN_SENSE Back-Up Oscillator Frequency Linear Regulator INTVCC Regulation Voltage 9.8 10 10.4 V 500 900 mV Dropout (VIN – INTVCC) INTVCC = –10mA Current Limit Below Undervoltage Threshold 17 25 mA Current Limit Above Undervoltage Threshold 80 120 mA Gate Driver tr GATE Driver Output Rise Time CL = 3300pF, 10% to 90% 20 ns tf GATE Driver Output Fall Time CL = 3300pF, 90% to 10% 20 ns GATE Output Low (VOL) GATE Output High (VOH) Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3799E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C 0.05 V INTVCC – 0.05 V to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3799I is guaranteed to meet performance specifications from –40°C to 125°C operating junction temperature. Note 3: Current flows out of the FB pin. 3799p 3 LT3799 TYPICAL PERFORMANCE CHARACTERISTICS VIN Start-Up Voltage vs Temperature Input Voltage Hysteresis vs Temperature VIN IQ vs Temperature 24.0 12.0 140 100 IQ (µA) INPUT VOLTAGE (V) HYSTERESIS VOLTAGE (V) 120 23.5 23.0 VIN = 24V 80 VIN = 12V 60 40 22.5 11.6 11.2 10.8 10.4 20 –25 50 25 0 75 TEMPERATURE (°C) 100 0 –50 125 3799 G01 2.100 2.100 2.075 2.075 2.050 2.050 2.025 2.025 VREF (V) VREF (V) VREF vs Temperature 2.000 NO LOAD 1.975 200µA LOAD 1.925 1.925 –25 50 25 0 75 TEMPERATURE (°C) 100 125 1.900 –25 50 25 0 75 TEMPERATURE (°C) 100 125 3799 G03 SENSE Pin Threshold Current vs Temperature MAX ILIM 100 NO LOAD 200µA LOAD 80 60 40 MIN ILIM 20 14 16 18 20 3799 G05 22 24 VIN (V) 26 28 32 30 0 –50 –25 3799 G05 50 25 0 75 TEMPERATURE (°C) 100 125 3799 G06 Minimum Oscillator Frequency vs Temperature 375 70 350 60 325 50 FREQUENCY (kHz) FREQUENCY (kHz) 10.0 –50 3799 G02 Maximum Oscillator Frequency vs Temperature 300 275 250 225 –50 125 120 1.975 1.950 100 VREF vs VIN 2.000 1.950 1.900 –50 50 25 0 75 TEMPERATURE (°C) –25 THRESHOLD (mV) 22.0 –50 40 30 20 –25 50 25 0 75 TEMPERATURE (°C) 100 125 3799 G07 10 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 3799 G08 3799p 4 LT3799 TYPICAL PERFORMANCE CHARACTERISTICS CT Pin Discharge Current vs Temperature CT Pin Charge Current vs Temperature 200 CT DISCHARGE CURRENT (nA) 8 6 4 2 –25 50 25 0 75 TEMPERATURE (°C) 100 180 170 150 –50 125 1.2 100 10.05 9.6 10.00 3799 G12 25.00 24.75 50 25 0 75 TEMPERATURE (°C) 100 125 3799 G15 100 125 9.95 3799 G13 1.2 25 1.0 20 0.8 15 220V APPLICATION 0.4 5 0.2 50 25 0 75 TEMPERATURE (°C) 100 125 3799 G16 PAGE 17 SCHEMATIC 0.6 10 –25 10 12 14 16 18 20 22 24 26 28 30 34 VIN (V) LED Current vs TRAIC Angle 30 0 –50 PART OFF 3799 G14 Maximum Shunt Current vs Temperature SHUNT CURRENT (mA) ISHUNT = 10mA 125 3799 G11 10.10 9.8 50 0 25 75 TEMPERATURE (°C) 100 PART ON 10.15 10mA LOAD –25 50 25 0 75 TEMPERATURE (°C) INTVCC vs VIN NO LOAD 9.4 –50 –25 3799 G10 10.0 125 25.50 –25 0 –50 125 10.20 25.75 24.50 –50 100 10.4 VIN Shunt Voltage vs Temperature 25.25 50 25 0 75 TEMPERATURE (°C) 10.25 1.1 50 25 0 75 TEMPERATURE (°C) 0.1 10.6 10.2 1.3 INTVCC (V) CT PIN VOLTAGE (V) 1.4 –25 0.2 INTVCC vs Temperature 1.5 VIN SHUNT VOLTAGE (V) –25 3799 G09 CT Pin High Threshold vs Temperature 1.0 –50 0.3 160 INTVCC (V) 0 –50 190 ILED (A) CT CHARGE CURRENT (µA) 10 0.4 CT PIN VOLTAGE (V) 12 26.00 CT Pin Low Threshold vs Temperature 0 120V APPLICATION 0 30 120 90 60 TRIAC ANGLE (°C) 150 180 3799 G17 3799p 5 LT3799 TYPICAL PERFORMANCE CHARACTERISTICS LED Current vs Input Voltage 1.20 PAGE 17 SCHEMATIC: OPTIMIZED FOR 120V 1.15 PAGE 17 SCHEMATIC: OPTIMIZED FOR 220V 1.05 1.05 1.05 ILED (A) 1.10 1.00 PAGE 17 SCHEMATIC: UNIVERSAL 1.15 1.10 1.00 1.00 0.95 0.95 0.95 0.90 0.90 0.90 0.85 0.85 0.85 0.80 170 180 190 200 210 220 230 240 250 260 270 VIN (VAC) 0.80 100 110 120 130 VIN (VAC) 140 150 3799 G18 Power Factor vs Input Voltage 0.99 0.99 0.98 0.98 0.97 0.97 POWER FACTOR ( ) 1.00 0.96 0.95 0.94 0.93 0.92 0.90 PAGE 17 SCHEMATIC: OPTIMIZED FOR 120V 90 100 110 120 130 VIN (VAC) 140 150 3799 G21 95 1.00 PAGE 17 SCHEMATIC: OPTIMIZED FOR 220V 0.98 0.95 0.94 0.93 0.97 0.96 0.95 0.94 0.93 0.92 0.92 0.91 0.91 0.90 170 180 190 200 210 220 230 240 250 260 270 VIN (VAC) 0.90 PAGE 17 SCHEMATIC: OPTIMIZED FOR 120V 95 Efficiency vs Input Voltage 100 PAGE 17 SCHEMATIC: 95 UNIVERSAL PAGE 17 SCHEMATIC: OPTIMIZED FOR 220V EFFICIENCY (%) 90 85 80 75 90 85 80 75 85 80 75 70 70 70 65 65 65 60 170 180 190 200 210 220 230 240 250 260 270 VIN (VAC) 60 60 90 100 110 120 130 VIN (VAC) 140 150 3799 G24 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3799 G23 Efficiency vs Input Voltage 100 PAGE 17 SCHEMATIC: UNIVERSAL 0.99 3799 G22 90 EFFICIENCY (%) Power Factor vs Input Voltage 0.96 Efficiency vs Input Voltage 100 3799 G20 Power Factor vs Input Voltage 1.00 0.91 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3799 G19 POWER FACTOR ( ) 90 EFFICIENCY (%) 0.80 POWER FACTOR ( ) LED Current vs Input Voltage 1.20 1.10 ILED (A) ILED (A) 1.15 LED Current vs Input Voltage 1.20 3799 G25 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3799 G26 3799p 6 LT3799 PIN FUNCTIONS VIN (Pin 11): Input Voltage. This pin supplies current to the internal start-up circuitry and to the INTVCC LDO. This pin must be locally bypassed with a capacitor. A 25V shunt regulator is internally connected to this pin. INTVCC (Pin 13): Regulated Supply for Internal Loads and GATE Driver. Supplied from VIN and regulates to 10V (typical). INTVCC must be bypassed with a 4.7µF capacitor placed close to the pin. COMP+, COMP– (Pin 7, Pin 8): Compensation Pins for Internal Error Amplifier. Connect a capacitor between these two pins to compensate the internal feedback loop. DCM (Pin 10): Discontinuous Conduction Mode Detection Pin. Connect a capacitor and resistor in series with this pin to the third winding. VIN_SENSE (Pin 16): Line Voltage Sense Pin. The pin is used for sensing the AC line voltage to perform power factor correction. Connect the output of a resistor divider from the line voltage to this pin. The voltage on this pin should be between 1.25V to 1.5V at the maximum input voltage. CTRL1, CTRL2, CTRL3 (Pin 1, Pin 2, Pin 3): Current Output Adjustment Pins. These pins control the output current. The lowest value of the three CTRL inputs is compared to the negative input of the operational amplifier. Due to the unique nature of the LT3799 control loop, the maximum current does not directly correspond to the VCTRL voltages. SENSE (Pin 15): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor, RSENSE, and the source of the N-channel MOSFET. The negative terminal of the current sense resistor should be connected to the GND plane close to the IC. GATE (Pin 14): N-Channel MOSFET Gate Driver Output. Switches between INTVCC and GND. This pin is pulled to GND during shutdown state. FB (Pin 9): Voltage Loop Feedback Pin. FB is used to detect open LED conditions by sampling the third winding voltage. An open LED condition is reported if the CT pin is high and the FB pin is higher than 1.25V. CT (Pin 6): Timer Fault Pin. A capacitor is connected between this pin and ground to provide an internal timer for fault operations. During start-up, this pin is pulled to ground and then charged with a 10µA current. Faults related to the FB pin will be ignored until the CT pin reaches 1.25V. If a fault is detected, the controller will stop switching and begin to discharge the CT capacitor with a 200nA pull-down current. When the pin reaches 240mV, the controller will start to switch again. FAULT (Pin 5): Fault Pin. An open-collector pull-down on FAULT asserts if FB is greater than 1.25V with the CT pin higher than 1.25V. VREF (Pin 4): Voltage Reference Output Pin, Typically 2V. This pin drives a resistor divider for the CTRL pin, either for analog dimming or for temperature limit/compensation of LED load. Can supply up to 200µA. GND (Exposed Pad Pin 17): Ground. The exposed pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The exposed pad must be soldered to the circuit board for proper operation. 3799p 7 LT3799 BLOCK DIAGRAM D2 R4 C1 R5 FB 6 C4 5 FAULT DETECTION 3 16 VIN_SENSE VOUT + L1B N:1 C7 VOUT – 11 VIN + A8 – 1.22V INTVCC ONE SHOT CURRENT COMPARATOR A7 CTRL2 CTRL3 S R DRIVER Q MASTER LATCH COMP +– A5 + + – A1 + C5 R8 S SW1 – CTRL1 13 R7 600mV COMP+ 1M 2 10 DCM FAULT C6 1 L1A R2 A3 – 8 C2 D1 S&H CT + A2 + – 7 T1 R1 C3 L1C R10 9 VIN R3 GATE SENSE A4 MULTIPLIER LOW OUTPUT CURRENT OSCILLATOR M1 15 R6 GND A6 14 17 4 VREF 3799 BD 3799p 8 LT3799 OPERATION The LT3799 is a current mode switching controller IC designed specifically for generating an average current output in an isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage and current on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with an opto-isolator. The LT3799 uses a novel method of using the external MOSFETs peak current information from the sense resistor to calculate the output current of a flyback converter without the need of an opto-coupler. In addition, it also detects open LED conditions by examining the third winding voltage when the main power switch is off. Power factor has become an important specification for lighting. A power factor of one is achieved if the current drawn is proportional to the input voltage. The LT3799 modulates the peak current limit with a scaled version of the input voltage. This technique provides power factors of 0.97 or greater. The Block Diagram shows an overall view of the system. The external components are in a flyback topology configuration. The third winding senses the output voltage and also supplies power to the part in steady-state operation. The VIN pin supplies power to an internal LDO that generates 10V at the INTVCC pin. The novel control circuitry consists of an error amplifier, a multiplier, a transmission gate, a current comparator, a low output current oscillator and a master latch, which will be explained in the following sections. The part also features a sample-and-hold to detect open LED conditions, along with a FAULT pin. A comparator is used to detect discontinuous conduction mode (DCM) with a cap connected to the third winding. The part features a 1.9A gate driver. The LT3799 employs a micropower hysteretic start-up feature to allow the part to work at any combination of input and output voltages. In the Block Diagram, R3 is used to stand off the high voltage supply voltage. The internal LDO starts to supply current to the INTVCC when VIN is above 23V. The VIN and INTVCC capacitors are charged by the current from R3. When VIN exceeds 23V and INTVCC is in regulation at 10V, the part will began to charge the CT pin with 10µA. Once the CT pin reaches 340mV, switching begins. The VIN pin has 10.7V of hysteresis to allow for plenty of flexibility with the input and output capacitor values. The third winding provides power to VIN when its voltage is higher than the VIN voltage. A voltage shunt is provided for fault protection and can sink up to 15mA of current when VIN is over 25V. During a typical cycle, the gate driver turns the external MOSFET on and a current flows through the primary winding. This current increases at a rate proportional to the input voltage and inversely proportional to the magnetizing inductance of the transformer. The control loop determines the maximum current and the current comparator turns the switch off when the current level is reached. When the switch turns off, the energy in the core of the transformer flows out the secondary winding through the output diode, D1. This current decreases at a rate proportional to the output voltage. When the current decreases to zero, the output diode turns off and voltage across the secondary winding starts to oscillate from the parasitic capacitance and the magnetizing inductance of the transformer. Since all windings have the same voltage across them, the third winding rings too. The capacitor connected to the DCM pin, C1, trips the comparator, A2, which serves as a dv/dt detector, when the ringing occurs. This timing information is used to calculate the output current (description to follow). The dv/dt detector waits for the ringing waveform to reach its minimum value and then the switch turns back on. This switching behavior is similar to zero volt switching and minimizes the amount of energy lost when the switch is turned back on, improving efficiency as much as 5%. Since this part operates on the edge of continuous conduction mode and discontinuous conduction mode, this operating mode is called critical conduction mode (or boundary conduction mode). Primary-Side Current Control Loop The CTRL1/CTRL2/CTRL3 pins control the output current of the flyback controller. To simplify the loop, assume the VIN_SENSE pin is held at a constant voltage above 1V, eliminating the multiplier from the control loop. The error amplifier, A5, is configured as an integrator with the external capacitor, C6. The COMP+ node voltage is 3799p 9 LT3799 OPERATION converted to a current into the multiplier with the V/I converter, A6. Since A7’s output is constant, the output of the multiplier is proportional to A6 and can be ignored. The output of the multiplier controls the peak current with its connection to the current comparator, A1. The output of the multiplier is also connected to the transmission gate, SW1. The transmission gate, SW1, turns on when the secondary current flows to the output capacitor. This is called the flyback period (when the output diode D1 is on). The current through the 1M resistor gets integrated by A5. The lowest CTRL input is equal to the negative input of A5 in steady state. A current output regulator normally uses a sense resistor in series with the output current and uses a feedback loop to control the peak current of the switching converter. In this isolated case the output current information is not available, so instead the LT3799 calculates it using the information available on the primary side of the transformer. The output current may be calculated by taking the average of the output diode current. As shown in Figure 1, the diode current is a triangle waveform with a base of the flyback time and a height of the peak secondary winding current. In a flyback topology, the secondary winding current is N times the primary winding current, where N is the primary to secondary winding ratio. Instead of taking the area of the triangle, think of it as a pulse width modulation (PWM) waveform. During the flyback time, the average current is half the peak secondary winding current and zero dur- SECONDARY DIODE CURRENT IPK(sec) SWITCH WAVEFORM ing the rest of the cycle. The equation for expressing the output current is: IOUT = 0.5 • IPK • N • D where D is equal to the percentage of the cycle represented by the flyback time. The LT3799 has access to both the primary winding current, the input to the current comparator, and when the flyback time starts and ends. Now the output current can be calculated by averaging a PWM waveform with the height of the current limit and the duty cycle of the flyback time over the entire cycle. In the feedback loop previously described, the input to the integrator is such a waveform. The integrator adjusts the peak current until the calculated output current equals the control voltage. If the calculated output current is low compared to the control pin, the error amplifier increases the voltage on the COMP+ node, thus increasing the current comparator input. When the VIN_SENSE voltage is connected to a resistor divider of the supply voltage, the current limit is proportional to the supply voltage if COMP+ is held constant. The output of the error amplifier is multiplied with the VIN_SENSE pin voltage. If the LT3799 is configured with a fast control loop, slower changes from the VIN_SENSE pin will not interfere with the current limit or the output current. The COMP+ pin will adjust to the changes of the VIN_SENSE. The only way for the multiplier to function properly is to set the control loop to be an order of magnitude slower than the fundamental frequency of the VIN_SENSE signal. In the offline case, the fundamental frequency of the supply voltage is 120Hz, so the control loop unity gain frequency needs to be set less than approximately 120Hz. Without a large amount of energy storage on the secondary side, the output current is affected by the supply voltage changes, but the DC component of the output current is accurate. TRIAC Dimming Features TFLYBACK TPERIOD 3799 F01 Figure 1. Secondary Diode Current and Switch Waveforms The LT3799 incorporates some special features that aid in the design of an offline LED current source when used with a TRIAC dimmer. TRIAC dimmers are not ideal switches when turned off and allow milliamps of current to flow through them. This is an issue if used with a low quiescent part such as the LT3799. Instead of turning the main power 3799p 10 LT3799 OPERATION MOSFET off when the TRIAC is off, this power device is kept on and sinks the current to properly load the TRIAC. When the TRIAC turns on, the VIN_SENSE pin detects this and enables the loop, but the current comparator is always enabled and turns the switch off if it is tripped. Start-Up The LT3799 uses a hysteretic start-up to operate from high offline voltages. A resistor connected to the supply voltage protects the part from high voltages. This resistor is connected to the VIN pin on the part and also to a capacitor. When the resistor charges the part up to 23V and INTVCC is in regulation at 10V, the part begins to charge the CT pin to 340mV and then starts to switch. The resistor does not provide power for the part in steady state, but relies on the capacitor to start-up the part, then the third winding begins to provide power to the VIN pin along with the resistor. An internal voltage clamp is attached to the VIN pin to prevent the resistor current from allowing VIN to go above the absolute maximum voltage of the pin. The internal clamp is set at 25V and is capable of 28mA (typical) of current at room temperature. But, ideally, the resistor connected between the input supply and the VIN pin should be chosen so that less than 10mA is being shunted by this internal clamp. CT Pin and Faults The CT pin is a timing pin for the fault circuitry. When the input voltages are at the correct levels, the CT pin sources 10µA of current. When the CT pin reaches 340mV, the part begins to switch. The output voltage information from the FB pin is sampled but ignored until the CT pin reaches 1.25V. When this occurs, if the FB pin is above 1.25V, the fault flag pulls low. The FAULT pin is meant to be used with a large pull-up resistor to the INTVCC pin or another supply. The CT pin begins to sink 200nA of current. When the CT pin goes below 240mV, the part will re-enable itself, begin to switch, and start to source 10µA of current to the CT pin but not remove the fault condition. When the CT pin reaches 1.25V and FB is below 1.25V, the FAULT pin will no longer pull low and switching will continue. If not below 1.25V, the process repeats itself. Programming Output Current The maximum output current depends on the supply voltage and the output voltage in a flyback topology. With the VIN_SENSE pin connected to 1V and a DC supply voltage, the maximum output current is determined at the minimum supply voltage, and the maximum output voltage using the following equation: N IOUT(MAX) = 2 • (1− D) • 42 • R SENSE where D= VOUT • N VOUT • N + VIN The maximum control voltage to achieve this maximum output current is 2V • (1-D). It is suggested to operate at 95% of these values to give margin for the part’s tolerances. When designing for power factor correction, the output current waveform is going to have a half sine wave squared shape and will no longer be able to provide the above currents. By taking the integral of a sine wave squared over half a cycle, the average output current is found to be half the value of the peak output current. In this case, the recommended maximum average output current is as follows: IOUT(MAX) = (1− D) • N • 47.5% 42 • R SENSE where D= VOUT • N VOUT • N + VIN The maximum control voltage to achieve this maximum output current is (1-D) • 47.5%. For control voltages below the maximum, the output current is equal to the following equation: IOUT = CTRL • N 42 • R SENSE 3799p 11 LT3799 OPERATION The VREF pin supplies a 2V reference voltage to be used with the control pins. To set an output current, a resistor divider is used from the 2V reference to one of the control pins. The following equation sets the output current with a resistor divider: 2N R1= R2 − 1 42 • IOUT • RSENSE with AC, the following equation should be used with the correction factor: N IOUT = CTRL • 42 • R SENSE − CF where R1 is the resistor connected to the VREF pin and the CTRL pin and R2 is the resistor connected to the CTRL pin and ground. where CR is the output current correction factor on the Y-axis in Figure 3. When used with an AC input voltage, the LT3799 senses when the VIN_SENSE goes below 65mV and above 65mV for detecting when the TRIAC is off. During this low input voltage time, the output current regulation loop is off but the part still switches. This helps with output current regulation with a TRIAC but introduces a line regulation error. When VIN_SENSE is low, very little power is being delivered to the output and since the output current regulation loop is off, this time period needs to be accounted for in setting the output current. This time period slightly varies with line voltage. Figure 2 shows the correction factor in selecting the resistor divider resistors. When used 2N R1= R2 − 1 (42 • IOUT • RSENSE • CF) Setting Control Voltages for LED Over Temperature and Brownout Conditions Critical Conduction Mode Operation Critical conduction mode is a variable frequency switching scheme that always returns the secondary current to zero with every cycle. The LT3799 relies on boundary mode and discontinuous mode to calculate the critical current because the sensing scheme assumes the secondary current returns to zero with every cycle. The DCM pin uses a fast current input comparator in combination with a small capacitor to detect dv/dt on the third winding. To eliminate false tripping due to leakage inductance ringing, OUTPUT CURRENT CORRECTION FACTOR 1.16 1.14 1.12 1.10 1.08 1.06 1.04 1.02 0 0.5 1 PEAK VIN_SENSE Figure 2. Correction Factor in Selecting the Resistor Divider Resistors 1.5 3799 F03 Figure 3. Output Current Correction Factor 3799p 12 LT3799 OPERATION a blanking time of between 600ns and 2.25µs is applied after the switch turns off, depending on the current limit shown in the Leakage Inductance Blanking Time vs Current Limit curve in the Typical Performance Characteristics section. The detector looks for 40µA of current through the DCM pin due to falling voltage on the third winding when the secondary diode turns off. This detection is important since the output current is calculated using this comparator’s output. This is not the optimal time to turn the switch on because the switch voltage is still close to VIN + VOUT • N and would waste all the energy stored in the parasitic capacitance on the switch node. Discontinuous ringing begins when the secondary current reaches zero and the energy in the parasitic capacitance on the switch node transfers to the input capacitor. This is a secondorder network composed of the parasitic capacitance on the switch node and the magnetizing inductance of the primary winding of the transformer. The minimum voltage of the switch node during this discontinuous ring is VIN – VOUT • N. The LT3799 turns the switch back on at this time, during the discontinuous switch waveform, by sensing when the slope of the switch waveform goes from negative to positive using the dv/dt detector. This switching technique may increase efficiency by 5%. Sense Resistor Selection The resistor, RSENSE, between the source of the external N-channel MOSFET and GND should be selected to provide an adequate switch current to drive the application without exceeding the current limit threshold . For applications without power factor correction, select a resistor according to: RSENSE = 2(1− D)N • 95% IOUT • 42 where D= VOUT • N VOUT • N + VIN For applications with power factor correction, select a resistor according to: RSENSE = (1− D)N • 47.5% IOUT • 42 where D= VOUT • N VOUT • N + VIN Minimum Current Limit The LT3799 features a minimum current limit of approximately 7% of the peak current limit. This is necessary when operating in critical conduction mode since low current limits would increase the operating frequency to a very high frequency. The output voltage sensing circuitry needs a minimum amount of flyback waveform time to sense the output voltage on the third winding. The time needed is 350ns. The minimum current limit allows the use of smaller transformers since the magnetizing primary inductance does not need to be as high to allow proper time to sample the output voltage information. Errors Affecting Current Output Regulation There are a few factors affecting the regulation of current in a manufacturing environment along with some systematic issues. The main manufacturing issues are the winding turns ratio and the LT3799 control loop accuracy. The winding turns ratio is well controlled by the transformer manufacturer’s winding equipment, but most transformers do not require a tight tolerance on the winding ratio. We have worked with transformer manufacturers to specify ±1% error for the turns ratio. Just like any other LED driver, the part is tested and trimmed to eliminate offsets in the control loop and an error of ±3% is specified at 80% of the maximum output current. The error grows larger as the LED current is decreased from the maximum output current. At half the maximum output current, the error doubles to ±6%. There are a number of systematic offsets that may be eliminated by adjusting the control voltage from the ideal voltage. It is difficult to measure the flyback time with complete accuracy. If this time is not accurate, the control voltage needs to be adjusted from the ideal value to eliminate the offset but this error still causes line regulation errors. If the supply voltage is lowered, the time error becomes a smaller portion of the switching cycle period so the offset becomes smaller and vice versa. This error may be compensated for at the primary supply voltage, but this does 3799p 13 LT3799 OPERATION not solve the problem completely for other supply voltages. Another systematic error is that the current comparator cannot instantaneously turn off the main power device. This delay time leads to primary current overshoot. This overshoot is less of a problem when the output current is close to its maximum, since the overshoot is only related to the slope of the primary current and not the current level. The overshoot is proportional to the supply voltage, so again this affects the line regulation. Universal Input The LT3799 operates over the universal input range of 90VAC to 265VAC . Output current regulation error may be minimized by using two application circuits for the wide input range: one optimized for 120VAC and another optimized for 220VAC . The first application pictured in the Typical Applications section shows three options: universal input, 120VAC , and 220VAC . The circuit varies by three resistors. In the Typical Performance Characteristics section, the LED Current vs VIN graphs show the output current line regulation for all three circuits. Selecting Winding Turns Ratio Boundary mode operation gives a lot of freedom in selecting the turns ratio of the transformer. We suggest to keep the duty cycle low, lower NPS, at the maximum input voltage since the duty cycle will increase when the AC waveform is decreases to zero volts. A higher NPS increases the output current while keeping the primary current limit constant. Although this seems to be a good idea, it comes at the expense of a higher RMS current for the secondary-side diode which might not be desirable because of the primary side MOSFET’s superior performance as a switch. A higher NPS does reduce the voltage stress on the secondary-side diode while increasing the voltage stress on the primaryside MOSFET. If switching frequency at full output load is kept constant, the amount of energy delivered per cycle by the transformer also stays constant regardless of the NPS. Therefore, the size of the transformer remains the same at practical NPS’s. Adjusting the turns ratio is a good way to find an optimal MOSFET and diode for a given application. Switch Voltage Clamp Requirement Leakage inductance of an offline transformer is high due to the extra isolation requirement. The leakage inductance energy is not coupled to the secondary and goes into the drain node of the MOSFET. This is problematic since 400V and higher rated MOSFETs cannot always handle this energy by avalanching. Therefore the MOSFET needs protection. A transient voltage suppressor (TVS) and diode are recommended for all offline application and connected, as shown in Figure 4. The TVS device needs a reverse breakdown voltage greater than (VOUT + Vf)*N where VOUT is the output voltage of the flyback converter, Vf is the secondary diode forward voltage, and N is the turns ratio. VSUPPLY GATE 3799 F04 Figure 4. Clamp 3799p 14 LT3799 OPERATION Transformer Design Considerations Transformer specification and design is a critical part of successfully applying the LT3799. In addition to the usual list of caveats dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. Since the current on the secondary side of the transformer is inferred by the current sampled on the primary, the transformer turns ratio must be tightly controlled to ensure a consistent output current. A tolerance of ±5% in turns ratio from transformer to transformer could result in a variation of more than ±5% in output regulation. Fortunately, most magnetic component manufacturers are capable of guaranteeing a turns ratio tolerance of 1% or better. Linear Technology has worked with several leading magnetic component manufacturers to produce predesigned flyback transformers for use with the LT3799. Table 1 shows the details of several of these transformers. Loop Compensation The current output feedback loop is an integrator configuration with the compensation capacitor between the negative input and output of the operational amplifier. This is a one-pole system therefore a zero is not needed in the compensation. For offline applications with PFC, the crossover should be set an order of magnitude lower than the line frequency of 120Hz or 100Hz. In a typical application, the compensation capacitor is 0.1µF. In non-PFC applications, the crossover frequency may be increased to improve transient performance. The desired crossover frequency needs to be set an order of magnitude below the switching frequency for optimal performance. MOSFET and Diode Selection With a strong 1.9A gate driver, the LT3799 can effectively drive most high voltage MOSFETs. A low Qg MOSFET is recommended to maximize efficiency. In most applications, the RDS(ON) should be chosen to limit the temperature rise of the MOSFET. The drain of the MOSFET is stressed to VOUT • NPS + VIN during the time the MOSFET is off and the secondary diode is conducting current. But in most applications, the leakage inductance voltage spike exceeds this voltage. The voltage of this stress is determined by the switch voltage clamp. Always check the switch waveform with an oscilloscope to make sure the leakage inductance voltage spike is below the breakdown voltage of the MOSFET. A transient voltage suppressor and diode are slower than the leakage inductance voltage spike, therefore causing a higher voltage than calculated. Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted TRANSFORMER PART NUMBER SIZE (L × W × H) LPRI (µH) NPSA (NP:NS:NA) JA4429 7508110210 21.1mm × 21.1mm × 17.3mm 400 15.75mm × 15mm × 18.5mm 2000 750813002 15.75mm × 15mm × 18.5mm 750811330 TARGET APPLICATION (VOUT /IOUT) RPRI (mΩ) RSEC (mΩ) 1:0.24:0.24 252 126 Coilcraft 22V/1A 6.67:1:1.67 5100 165 Würth Elektronik 10V/0.4A 2000 20:1.0:5.0 6100 25 Würth Elektronik 3.8V/1.1A 43.2mm × 39.6mm × 30.5mm 300 6:1.0:1.0 150 25 Würth Elektronik 18V/5A 750813144 16.5mm × 18mm × 18mm 600 4:1:0.71 2400 420 Würth Elektronik 28V/0.5A 750813134 16.5mm × 18mm × 18mm 600 8:1:1.28 1850 105 Würth Elektronik 14V/1A 750811291 31mm × 31mm × 25mm 400 1:1:0.24 550 1230 Würth Elektronik 85V/0.4A 750813390 43.18mm × 39.6mm × 30.48mm 100 1:1:0.22 150 688 Würth Elektronik 90V/1A 750811290 31mm × 31mm × 25mm 460 1:1:0.17 600 560 Würth Elektronik 125V/0.32A X-11181-002 23.5mm × 21.4mm × 9.5mm 500 72:16:10 1000 80 Premo 30V/0.5A MANUFACTURER 3799p 15 LT3799 OPERATION The secondary diode stress may be as much as VOUT + 2 • VIN /NPS due to the anode of the diode ringing with the secondary leakage inductance. An RC snubber in parallel with the diode eliminates this ringing, so that the reverse voltage stress is limited to VOUT + VIN /NPS. With a high NPS and output current greater than 3A, the IRMS through the diode can become very high and a low forward drop Schottky is recommended. Discontinuous Mode Detection The discontinuous mode detector uses AC-coupling to detect the ringing on the third winding. A 10pF capacitor with a 500Ω resistor in series is recommended in most designs. Depending on the amount of leakage inductance ringing, an additional current may be needed to prevent false tripping from the leakage inductance ringing. A resistor from INTVCC to the DCM pin adds this current. Up to an additional 100µA of current may be needed in some cases. The DCM pin is roughly 0.7V, therefore the resistor value is selected using the following equation: R= 10V − 0.7V I where I is equal to the additional current into the DCM pin. Power Factor Correction/Harmonic Content Protection from Open LED and Shorted LED Faults The LT3799 detects output overvoltage conditions by looking at the voltage on the third winding. The third winding voltage is proportional to the output voltage when the main power switch is off and the secondary diode is conducting current. Sensing the output voltage requires delivering power to the output. Using the CT pin, the part turns off switching when a overvoltage condition occurs and rechecks to see if the overvoltage condition has cleared, as described in “CT Pin and Faults” in the Operation section. This greatly reduces the output current delivered to the output but a Zener is required to dissipate 2% of the set output current during an open LED condition. The Zener diode’s voltage needs to be 10% higher than the output voltage set by the resistor divider connected to the FB pin. Multiple Zener diodes in series may be needed for higher output power applications to keep the Zener’s temperature within the specification. During a shorted LED condition, the LT3799 operates at the minimum operating frequency. In normal operation, the third winding provides power to the IC, but the third winding voltage is zero during a shorted LED condition. This causes the part’s VIN UVLO to shutdown switching. The part starts switching again when VIN has reached its turn-on voltage. The LT3799 attains high power factor and low harmonic content by making the peak current of the main power switch proportional to the line voltage by using an internal multiplier. A power factor of >0.97 is easily attainable for most applications by following the design equations in this datasheet. With proper design, LT3799 applications meet IEC 6100-3-2 Class C harmonic standards. 3799p 16 LT3799 TYPICAL APPLICATIONS Universal TRIAC Dimmable 20W LED Driver L2 800µH L1 33mH 90V TO 265V AC BR1 C3 0.22µF C1 0.1µF C2 0.1µF R1 200Ω R3 499k R4 499k R7 100k R6 D2 20Ω R8 100k C4 C5 4.7pF 10µF D3 R13 2k VIN DCM VIN_SENSE FB R5 3.48k R18 100k 4:1:1 R4 100k Z1 R15 4.99k LT3799 R16 20Ω VREF R16 32.4k 100k NTC R9 40.2k CTRL3 GATE CTRL2 SENSE CTRL1 VINTVCC R10 24.9k C9 4.7µF GND FAULT CT COMP+ FAULT 1A D4 C10 560µF ×2 D1 20W LED POWER Z2 M1 RS 0.05Ω C8 2.2nF COMP– 3799 TA02 C7, 0.1µF BR1: DIODES, INC. HD06 D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: DIODES INC. BAV20W DR: CENTRAL SEMICONDUCTOR CMR1U-02M Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ5938B T1: COILCRAFT JA4429-AL M1: FAIRCHILD FDPF15N65 Component Values for Input Voltage Ranges R5 (Ω) R10 (Ω) RS (Ω) R1 (Ω) C2 (µF) C3 (µF) Optimized for 110V 6.34k 16.2k 0.05 200 0.1 0.22 Optimized for 220V 3.48k 24.9k 0.075 1.00k 0.033 0.1 Universal 3.48k 15.4k 0.05 200 0.1 0.22 3799p 17 LT3799 TYPICAL APPLICATIONS Universal Input TRIAC Dimmable 4W LED Driver L1 3.3mH R20, 10k 90V TO 265V AC C1 33nF BR1 L1 3.3mH R21, 10k L2, 3.3mH C3 68nF C2 22nF R1 750Ω R3 499k R4 499k R7 100k R6 D2 20Ω R8 100k C4 C5 4.7pF 10µF D3 R13 10k VIN DCM VIN_SENSE FB R5 3.48k R18 100k 20:5:1 LT3799 CTRL3 GATE CTRL2 SENSE CTRL1 VINTVCC R10 32.4k FAULT BR1: DIODES, INC. HD06 D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: CENTRAL SEMICONDUCTOR CMMSHI-100 D4: CENTRAL SEMICONDUCTOR CMSH2-40L Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ59198 T1: WÜRTH ELEKTRONIK WE-750813002 M1: FAIRCHILD FQU5N60 GND FAULT CT COMP+ C6 0.1µF COMP– 1A D4 Z1 R15 4.99k VREF R9 40.2k R4 100k R16 20Ω C9 4.7µF C10 1500µF D1 M1 RS 0.3Ω 4W LED POWER Z2 C8 2.2nF 3799 TA03 C7, 0.1µF 3799p 18 LT3799 PACKAGE DESCRIPTION MSE Package 16-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1667 Rev A) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ± 0.102 (.112 ± .004) 5.23 (.206) MIN 2.845 ± 0.102 (.112 ± .004) 0.889 ± 0.127 (.035 ± .005) 8 1 1.651 ± 0.102 (.065 ± .004) 1.651 ± 0.102 3.20 – 3.45 (.065 ± .004) (.126 – .136) 0.305 ± 0.038 (.0120 ± .0015) TYP 16 0.50 (.0197) BSC 4.039 ± 0.102 (.159 ± .004) (NOTE 3) RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOSE 0.280 ± 0.076 (.011 ± .003) REF 16151413121110 9 DETAIL “A” 0° – 6° TYP 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 4.90 ± 0.152 (.193 ± .006) GAUGE PLANE 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.17 – 0.27 (.007 – .011) TYP 1234567 8 0.50 (.0197) BSC NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.86 (.034) REF 0.1016 ± 0.0508 (.004 ± .002) MSOP (MSE16) 0608 REV A 3799p Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3799 TYPICAL APPLICATION Universal Input TRIAC Dimmable 14W LED Driver L2 750µH L1 39mH 90V TO 265V AC BR1 C1 47nF C2 0.1µF C3 0.22µF R1 250Ω R3 499k R2 250Ω R4 499k R7 100k R6 D2 20Ω R8 100k C4 C5 4.7pF 10µF D3 4:1:0.71 R13 2k VIN DCM VIN_SENSE FB R5 3.48k LT3799 R16 10k R9 40.2k PHOTOCELL R17 10k CTRL3 GATE CTRL2 SENSE CTRL1 VINTVCC R10 23.2k GND FAULT CT COMP FAULT C6 0.1µF BR1: DIODES, INC. HD06 D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: DIODES INC. BAV20W D4: DIODES INC. DFLS1150 Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ5938B T1: WÜRTH ELEKTRONIK WE750813144 M1: ST MICRO STD12N65M5 + COMP– 0.5A D4 Z1 R15 5.90k VREF R18 100k R4 100k C10 390µF ×2 D1 R16 20Ω C9 4.7µF 14W LED POWER Z2 RS 0.10Ω C8 2.2nF 3799 TA04 C7, 0.1µF RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT3755/ LT3755-1/ LT3755-2 High Side 60V, 1MHz LED Controller with 3000:1 True Color PWM™ Dimming VIN : 4.5V to 40V, VOUT(MAX) = 60V, Dimming: 3000:1 True Color PWM, ISD < 1µA, 3mm × 3mm QFN-16 and MSOP-16E Packages LT3756/ LT3756-1/ LT3756-2 High Side 100V, 1MHz LED Controller with 3000:1 True Color PWM Dimming VIN : 6V to 100V, VOUT(MAX) = 100V, Dimming: 3000:1 True Color PWM, ISD < 1µA, 3mm × 3mm QFN-16 and MSOP-16E Packages LT3743 Synchronous Step-Down 20A LED Driver with Three-State LED Current Control VIN : 5.5V to 36V, Dimming: 10000:1 True Color PWM, ISD < 1µA, 5mm × 8mm QFN-52 Package LT3518 2.3A, 2.5MHz High Current LED Driver with 3000:1 Dimming VIN : 3V to 30V, Dimming: 3000:1 True Color PWM, ISD < 1µA, 4mm × 4mm QFN-16 Package LT3517 1.3A, 2.5MHz High Current LED Driver with 3000:1 Dimming VIN : 3V to 30V, Dimming: 3000:1 True Color PWM, ISD < 1µA, 4mm × 4mm QFN-16 Package LT3741 High Power, Constant-Current, Constant-Voltage Synchronous Step-Down Controller VIN : 6V to 36V, Average Current Mode Control, ISD < 1µA, 4mm × 4mm QFN-20 and TSSOP-20E Packages 3799p 20 Linear Technology Corporation LT 0211 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2011