TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 CONSTANT CURRENT LED DRIVER FEATURES • • • • • • • • • • • • • DESCRIPTION Current Source with 18 V Overvoltage Protection Powers Up to 4 LEDs in Series Input Voltage Range . . 1.8 V to 6.0 V Internal 30 V Switch Up to 85% Efficiency Precise Brightness Control Using PWM Signal or Analog Signal Switching Frequency . . Up to 1 MHz Internal Power MOSFET Switch . . 400 mA Operates With Small Output Capacitors Down to 100 nF Disconnects LEDs During Shutdown No Load Quiescent Current . . 38 µA Typ Shutdown Current . . 0.1 µA Typ Available in a Small 3 mm × 3 mm QFN Package APPLICATIONS • White LED Supply for Display Backlight/Sidelight in – PDA, Pocket PC, Smart Phones – Handheld Devices – Cellular Phones The TPS61043 is a high frequency boost converter with constant current output that drives white LEDs or similar. The LED current is set with the external sense resistor (RS) and is directly regulated by the feedback pin (FB) that regulates the voltage across the sense resistor RS to 252 mV (typ). To control LED brightness, the LED current can be pulsed by applying a PWM (pulse width modulated) signal with a frequency range of 100 Hz to 50 kHz to the control pin (CTRL). To allow higher flexibility, the device can be configured where the brightness can be controlled by an analog signal as well, as described in the application information section. To avoid possible leakage currents through the LEDs during shutdown, the control pin (CTRL) disables the device and disconnects the LEDs from ground. For maximum safety during operation, the output has integrated overvoltage protection that prevents damage to the device by limiting the output voltage to typically 18 V in case of a high impedance output (e.g. faulty LED). The TPS61042 device provides a solution for applications where higher LED currents or more than four LEDs in series need to be powered. L1 4.7 H VIN 1.8 V to 6 V CIN 4.7 F 3 VIN D1 SW CO 100 nF 25 V 8 5 7 CTRL OVP 6 GND LED 1 4 FB R 2 S Enable/PWM Brightness Control 100 Hz to 50 kHz A. RS 13 Output capacitor values like 1 µF and larger, reduce the LED ripple current and improve line regulation. Figure 1. Typical Application Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2003, Texas Instruments Incorporated TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 ORDERING INFORMATION (1) (1) TA PACKAGE PACKAGE MARKING -40°C to 85°C TPS61043DRB AQN The DRB package is available taped and reeled. Add R suffix (TPS61043DRBR) to order quantities of 3000 devices per reel. Add T suffix (TPS61043DRBT) to order quantities of 250 devices per reel. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) TPS61043 Supply Voltages, v(VIN) (2) -0.3 V to 7 V Voltages, V(Rs), V(CTRL), V(FB) -0.3 V to Vin + 0.3 V Voltages, V(SW), V(LED) (2) 30 V Voltage, V(OVP) 30 V Continuous power dissipation See Dissipation Rating Table Operating junction temperature range -40°C to 150°C Storage temperature range, TSTG -65°C to 150°C Lead temperature (soldering, 10 sec) (1) (2) 260°C Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. DISSIPATION RATING (1) PACKAGE THERMAL RESISTANCE TA ≤ 25°C POWER RATING DERATING FACTOR ABOVE TA= 25°C TA = 70°C POWER RATING TA = 85°C POWER RATING 8-pin QFN RΘJA = 270°C/W (1) 370 mW 3.7 mW/°C 204 mW 148 mW The thermal resistance junction to ambient of the 8-pin QFN package is measured with a standard 2-layer PCB without vias for the thermal pad. RECOMMENDED OPERATING CONDITIONS MIN VI Input voltage range 1.8 TA Operating ambient temperature TJ Operating junction temperature NOM MAX UNIT 6.0 V -40 85 °C -40 125 °C MAX UNIT ELECTRICAL CHARACTERISTICS VI = 3.6 V, CTRL= VI, TA = -40°C to + 85°C, typical values are at TA= 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP SUPPLY CURRENT VI Input voltage range 6.0 V I(Q) Operating quiescent current into VIN IO = 0 mA, not switching 1.8 38 65 µA IO(sd) Shutdown current CTRL = GND 0.1 1 µA VUVLO Under-voltage lockout threshold VI falling 1.5 1.7 V CTRL VIH CTRL high level input voltage VIL CTRL low level input voltage IIkg CTRL input leakage current CTRL = GND or VIN ton Minimim CTRL pulse witdh to enable CTRL = low to high 500 toff Minimum CTRL pulse width to disable CTRL = high to low 10 2 1.3 V 0.3 V 0.1 µA 32 ms us TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 ELECTRICAL CHARACTERISTICS (continued) VI = 3.6 V, CTRL= VI, TA = -40°C to + 85°C, typical values are at TA= 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX f(CTRL) PWM switching frequency applied to CTRL 0.1 50 D(CTRL) PWM duty cycle applied to CTRL 1% 100% UNIT kHz POWER SWITCH AND CURRENT LIMIT (SW) VS Maximum switch voltage rds(ON) MOSFET on-resistance VI = 3.6 V; I(SW) = 200 mA Ilkg MOSFET leakage current V(SW) = 28 V ILIM MOFSET current limit TON Power switch maximum on-time TOFF Power switch minimum off-time 320 VO = 15 V 30 V 300 600 mΩ 0.1 10 µA 400 480 mA 4.5 µs 400 ns LED SWITCH AND CURRENT LIMIT (LED) VS Maximum switch voltage I Maximum LED switch current (LED) rds(ON) MOSFET on-resistance VI = 3.6 V; ISW = 20 mA Ilkg MOSFET leakage current V(LED)= 28 V VO Output voltage range OVP connected I(FB) Feedback input bias current (1) V(FB) = 0.252 V VFB Feedback trip point voltage 1.8 V ≤ VI≤ 6.0 V V(OVP) Output overvoltage protection VO rising Vhys(OVP) Output overvoltage protection hysteresis I(OVP) OVP input current 30 V 60 mA 1 2 Ω 0.1 10 µA OUTPUT (1) VI 16.9 V 100 nA 244 252 260 mV 17 18 19 V 23 µA 3.65 VO = 15 V 17 V The feedback input is high impedance MOSFET Gate input. PIN ASSIGNMENT DRB PACKAGE (TOP VIEW) LED 1 RS 2 VIN 3 FB 4 † 8 SW Exposed Thermal † Die Pad 7 OVP 6 GND 5 CTRL The exposed thermal die pad is connected to GND. 3 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 Terminal Functions TERMINAL I/O DESCRIPTION NAME NO. CTRL 5 I Combined enable and PWM control pin. If CTRL is constantly pulled high, the device is enabled and the internal LED switch (Q2) is constantly turned on. When CTRL is pulled to GND, the device is disabled. Apply a PWM signal (100 Hz to 50 kHz) to this pin to control the brightness of the LEDs FB 4 I Feedback. FB regulates the LED current through the sense resistor by regulating the voltage across RS to 252 mV. GND 6 LED 1 I Input of the LED switch (Q2). Connect the LEDs to this pin. OVP 7 I Overvoltage protection. OVP is connected to the output capacitor of the converter. RS 2 O Output of the internal LED switch. The sense resistor that programs the LED current is connected to RS. SW 8 I Drain of the integrated switch (Q1) VIN 3 I Input supply pin. GND FUNCTIONAL BLOCK DIAGRAM EN SW EN VIN UVLO Bias VREF 0.252 V Control Logic Q1 Gate Driver Thermal Shutdown OVP CTRL Enable Control Logic EN R1 1080 k Current Limit Softstart R2 25 k 4.5 s Max On Time GND PWM Gate Drive Overvoltage Protection + - VREF 4 + LED Q2 Error Comparator FB 0.4 V 400 ns Min Off Time RS TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 TYPICAL CHARACTERISTICS L1 4.7 H VIN 1.8 V to 6 V CIN 4.7 F 3 VIN D1 SW CO 1 F 25 V 8 5 7 CTRL OVP 6 GND LED 1 4 FB R 2 S Enable/PWM Brightness Control 100 Hz to 50 kHz RS 13 L1 = Murata LQH32CN4R7 (4.7 H) D1 = Zetex Schottky ZHCS400 CIN = 4.7 F X5R 20% JMK212BJ475MG-T COUT = 1 F X7R 10% TMK316BJ105KL-T 5 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 TYPICAL CHARACTERISTICS Table of Graphs FIGURE η Efficiency vs LED current Figure 2 vs Input voltage Figure 3 IQ Operating Quiescent Current into VIN vs Input voltage and Temperature Figure 4 V(FB) Feedback voltage vs Temperature Figure 5 I(FB) Feedback current vs Temperature Figure 6 rds(on) Main switch Q1 vs Temperature Figure 7 vs Input voltage Figure 8 LED switch Q2 ILED Average LED current vs Temperature Figure 9 vs Input voltage Figure 10 vs PWM duty cycle on CTRL pin Figure 11 Soft start Figure 12 PFM operation (fixed peak current control) Figure 13 Burst mode operation (fixed peak current control) Figure 14 PWM dimming Figure 15 Efficiency vs LED Current Efficiency vs Input Voltage 90 90 4 LEDs ≈ 13 V, CO = 1 µF Efficiency - % 86 82 VI = 3.6 V 80 78 VI = 2.4 V 76 84 82 IO = 15 mA 80 78 76 74 74 72 72 70 0.1 70 1 10 IO - Output Current - mA Figure 2. 6 86 VI = 4.2 V 84 4 LEDs ≈ 13 V, CO = 1 µF 88 Efficiency - % 88 100 1.8 2.8 3.8 VI - Input Voltage - V Figure 3. 4.8 5.8 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 Operating Quiescent Current into VIN vs Input Voltage and Temperature Feedback Voltage vs Temperature 260 45 V(fb) - Feedback Voltage - mV Quiescent Current into VI/µA 50 TA = 85°C 40 35 TA = 25°C 30 25 TA = - 40°C 20 15 10 5 0 258 VCC = 3.7 V 256 254 252 250 248 246 244 242 240 1.8 2.4 3.0 3.6 4.2 4.8 5.4 6.0 - 40 35 Figure 5. Feedback Current vs Temperature rds(on) Main Switch (Q1) vs Temperature 60 85 60 85 60 85 500 rds(on) - On-State Resistance - MΩ I(fb) - Feedback Current - nA 10 Figure 4. 60 40 VCC = 3.6 V 20 0 - 20 VCC = 5 V VCC = 2.4 V - 40 - 60 VCC = 3.6 V 450 400 350 300 250 200 - 40 - 15 10 35 60 85 - 40 TA - Free-Air Temperature - °C - 15 10 35 TA - Free-Air Temperature - °C Figure 6. Figure 7. rds(ON) Main Switch (Q1) vs Input Voltage rds(on) LED Switch (Q2) vs Temperature 600 1.6 rds(on) - On-State Resistance - Ω rds(on) - On-State Resistance - MΩ - 15 TA - Free-Air Temperature - °C VI - Input Voltage - V TA = 27°C 500 400 300 200 100 0 1.5 VCC = 3.6 V 1.4 1.3 1.2 1.1 1.0 0.9 0.8 0.7 0.6 1.8 2.4 3.0 3.6 4.2 VI - Input Voltage - V Figure 8. 4.8 5.4 6.0 - 40 - 15 10 35 TA - Free-Air Temperature - °C Figure 9. 7 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 rds(on) LED Switch (Q2) vs Input Voltage Average LED Current vs PWM Duty Cycle on CTRL Pin 20 TA = 25°C 2.5 IO - Output Current - mA rds(on) - On-State Resistance - Ω 3.0 2.0 1.5 1.0 0.5 0.0 15 10 fPWM = 50 kHz fPWM = 100 Hz 5 fPWM = 25 kHz 0 1.8 2.4 3.0 3.6 4.2 4.8 5.4 6.0 0 VI - Input Voltage - V 20 40 60 Duty Cycle - % Figure 10. Figure 11. Softstart PFM Operation Vsw 5V/Div Vout 10V/Div Vout 500mV/Div CTRL 1V/Div LED Current 20mA/Div Input Current 100mA/Div 50s/Div Figure 12. Figure 13. Bust Mode Operation PWM Dimming Vsw 5V/Div Vsw 5V/Div Vout 50mV/Div Vout 500mV/Div LED Current 20mA/Div LED Current 20mA/Div 2.5s/Div Figure 14. 8 2.5s/Div 25s/Div Figure 15. 80 100 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION Operation The TPS61043 operates like a standard boost converter but regulates the voltage across the sense resistor (RS) instead of the output voltage. This gives an accurate regulated LED current independent of the input voltage and number of LEDs connected. With integrated overvoltage protection (OVP) the TPS61043 is configured as a current source with overvoltage protection ideally suited to drive LEDs. The device can generate output voltages of up to 16.9 V if the OVP-function is used and has an internal 400 mA MOSFET switch (Q1). This allows up to four LEDs to be connected in series to the output. The internal LED switch (Q2) in series with the LEDs has a maximum current rating of 60 mA and disconnects the LEDs from ground during shutdown. The LED switch is driven by a PWM signal applied to the control pin (CTRL), which directly controls the LED brightness. With this control method the LED brightness depends on the PWM duty cycle only and is independent of the PWM frequency and amplitude. If the OVP-function is not needed, the device can be used to generate output voltages up to 28V. Boost Converter The boost converter operates in a pulse frequency modulation (PFM) scheme with constant peak current control. This control scheme maintains high efficiency over the entire load current range and with a switching frequency of up to 1 MHz, enables the use of small external components. The converter monitors the sense voltage across RS with the feedback pin (FB) and, when the feedback voltage falls below the reference voltage (252 mV typ), the main switch turns on and the current ramps up. The main switch turns off when the inductor current reaches the internally set peak current of 400 mA (typ). Refer to the peak current control section for more information. The second criteria that turns off the main switch is the maximum on-time of 4.5 µs (typ). This limits the maximum on-time of the converter in extreme conditions. As the switch is turned off the external Schottky diode is forward biased, delivering the stored inductor energy to the output. The main switch remains off until the minimum off time of 400 ns (typ) has passed and the feedback voltage is below the reference voltage again. Using this PFM peak current control scheme, the converter operates in discontinuous conduction mode (DCM) where the switching frequency depends on the inductor, input and output voltage, and LED current. Lower LED currents reduce the switching frequency, which results in high efficiency over the entire LED current range. This regulation scheme is inherently stable, allowing a wide range for the selection of the inductor and output capacitor. Peak Current Control (Boost Converter) The internal switch is turned on until the inductor current reaches the DC current limit (ILIM) of 400 mA (typ) . Due to the internal current limit delay of 100 ns (typ) the actual current exceeds the DC current limit threshold by a small amount. The typical peak current limit can be calculated: V I I I 100 ns P(typ) (LIM) L V I 400 mA I 100 ns P L The higher the input voltage and the lower the inductor value, the greater the current limit overshoot. Softstart All inductive step-up converters exhibit high in-rush current during start-up if no special precautions are taken. This can cause voltage drops at the input rail during start-up, which may result in an unwanted or premature system shutdown. The TPS61043 limits this in-rush current during start-up by increasing the current limit in two steps starting from ILIM/4 for 256 switch cycles to ILIM/2 for the next 256 switch cycles and then full current limit. See Figure 12 for typical start-up behavior. 9 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION (continued) Control (CTRL) The CTRL pin serves two functions. One is the enable and disable of the device. The other is the PWM control of the internal LED switch (Q2). The CTRL pin can be used as a standard enable pin for the device if no PWM signal is applied to the CTRL pin. To enable the device, the CTRL pin must be pulled high for a time period of at least 500 µs. The device starts with the Softstart cycle. Pulling the CTRL pin to GND for a time period ≥32 ms disables the device, disconnecting the LEDs from GND by opening the LED switch (Q2) to avoid any LED leakage current. See Figure 16 for the CTRL pin timing. tp toff ton ton High Low Minimum On-Time to Enable the Device (500 s) t D = tp/t Minimum Off-Time to Disable the Device (32 ms) Figure 16. CTRL Timing Diagram To enable the device, the CTRL signal must be high for 500 µs (see Figure 16). The PWM signal can then be applied with a pulse width (tp) greater or smaller than tON. To force the device into shutdown mode, the CTRL signal must be low for at least 32 ms. Requiring the CTRL pin to be low for 32 mS before the device enters shutdown allows for PWM dimming frequencies as low as 100 Hz. The device is enabled again when a CTRL signal is high for a period of 500 µs minimum. See Figure 11 for the PWM duty cycle versus LED current characteristic. The internal LED switch (Q2) is driven by the PWM signal when applied to the CTRL pin. Applying a PWM signal in the range of 100 Hz to 50 kHz allows the LED current to be pulsed with the duty cycle of the PWM signal. The CTRL pin accepts a PWM duty cycle from D = 1% to 100%. Duty cycles below 1% are also possible with the restriction that the device is forced into shutdown as the off time of the applied PWM signal exceeds 10 ms. When a PWM signal is applied to the CTRL pin the LED switch (Q2) turns on immediately. The internal error comparator is disabled for 400 ns. This 400 ns delay time is required to establish the correct voltage level across the sense resistor RS after the LED switch (Q2) is closed. To achieve good LED current accuracy and linearity, the switching frequency of the converter must be higher than the PWM frequency applied to the CTRL pin. This CTRL pin must be terminated. Overvoltage Protection (OVP) As with any current source, the output voltage rises as the output impedance increases as for example with a disconnected load. To prevent the output voltage from exceeding the maximum main switch (Q1) voltage rating, an overvoltage protection (OVP) circuit is integrated. With an OVP threshold voltage of 19 V maximum, up to 4 LEDs can be connected in series. This allows the use of a cheaper output capacitor with a 25 V voltage rating. When the output voltage exceeds the OVP threshold voltage, (Q1) turns off. The converter switch remains off until the output voltage falls below the OVP threshold voltage. As long as the output voltage is below the OVP threshold the converter continues its normal operation, until the output voltage exceeds the OVP threshold again. If overvoltage protection is not needed, then the OVP pin should be connected to GND. In this case the TPS61043 can be used to generate output voltages up to 28V. 10 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION (continued) Undervoltage Lockout An undervoltage lockout feature prevents mis-operation of the device at input voltages below 1.5 V (typ). As long as the input voltage is below the undervoltage threshold the device remains off, with the main MOSFET switch (Q1) and the LED switch (Q2) open. Thermal Shutdown An internal thermal shutdown is implemented in the TPS61043 that shuts down the device if the typical junction temperature of 160°C is exceeded. If the device is in thermal shutdown mode, the main MOSFET switch (Q1) and the LED switch (Q2) are open. Inductor Selection, Maximum Load Current, and Switching Frequency The PFM peak current control scheme of the TPS61043 is inherently stable. The inductor value does not affect the stability of the regulator. The selection of the inductor together with the nominal LED current, input, and output voltage of the application determines the switching frequency of the converter. The first step is to calculate the maximum load current the converter can support using the selected inductor. The inductor value has less effect on the maximum available load current and is only of secondary order. A good inductor value to start with is 4.7 µH. Depending on the application, inductor values down to 1.0 µH can be used. The maximum inductor value is determined by the maximum on time of the switch of 4.5 µs (typ). The peak current limit of 400 mA (typ) must be reached within this 4.5 µs for proper operation. The maximum load current of the converter is determined at the operation point where the converter starts to enter the continuous conduction mode. The converter must always operate in discontinuous conduction mode to maintain regulation. Depending on the time period of the inductor current fall time being larger or smaller compared to the minimum off time of the converter (400 ns typ), the maximum load current can be calculated. Inductor fall time: Ip L t f V V O I for tf≥ 400 ns I I V I P LOAD (max) 2V O for tf≤ 400 ns I LOAD (max) V O – V I I2 L V I P 2 I L 2 400 ns V P I with: L = selected inductor value η = expected converter efficiency. Typically between 70% to 85% V I 400 mA I 100 ns P L (Peak inductor current as described in the peak current control section) The above formula contains the expected converter efficiency that allows calculating the expected maximum load current the converter can support. The efficiency can be taken out of the efficiency graphs shown in Figure 2 and Figure 3 or 80% can be used as an accurate estimation. If the converter can support the desired LED current, the next step is to calculate the converter switching frequency at the operation point, which must be ≤1 MHz. Also the converter switching frequency should be much higher than the applied PWM frequency at the CTRL pin to avoid non-linear brightness control. Assuming the converter shows no double pulses or pulse bursts (Figure 13 and Figure 14) on the switch node (SW) the switching frequency at the operation point can be calculated as: 11 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION (continued) ƒs 2I I O LIM V O V V I F V I 100 ns 2 2 1MHz L with: ILIM(min) = minimum switch current limit (320 mA typ) L = selected inductor value IO = nominal load or LED current VF = Rectifier diode forward voltage (typically 0.3 V) The smaller the inductor value, the higher the switching frequency of the converter but the lower the efficiency. The selected inductor must have a saturation current that meets the maximum peak current of the converter as calculated in the peak current control section. Use the maximum value for ILIM (480 mA) for this calculation. Another important inductor parameter is the DC resistance. The lower the DC resistance the higher the efficiency of the converter. See Table 1 and Figure 22 to Figure 26 for a selection of inductors. Table 1. Possible Inductors (or equivalent) INDUCTOR VALUE COMPONENT SUPPLIER SIZE 10 µH muRata LQH43CN100K01 4,5 mm×3,2 mm×2.6 mm 4.7 µH muRata LQH32CN4R7M11 3,2 mm×2,5 mm×2,0 mm 10 µH Coilcraft DO1605T-103MX 5,5 mm ×4,1 mm ×1,8 mm 4.7 µH Sumida CDRH3D16-4R7 3,8 mm×3,8 mm×1,8 mm 3.3 µH Sumida CMD4D11-3R3 3,5 mm×5,3 mm×1,2 mm 4.7 µH Sumida CMD4D11-4R7 3,5 mm×5,3 mm×1,2 mm 3.3 µH Sumida CMD4D11-3R3 3,5 mm×5,3 mm×1,2 mm 4.7 µH Coiltronics SD12-4R7 5,2 mm×5,2 mm×1,2 mm 3.3 µH Coilcraft LPO1704-332M 6,6 mm×5,5 mm×1,0 mm 4.7 µH Coilcraft LPO1704-472M 6,6 mm×5,5 mm×1,0 mm Output Capacitor Selection and Line Regulation For better output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low ESR value, but depending on the application, tantalum capacitors can be used. The selection of the output capacitor value directly influences the output voltage ripple of the converter which also influences line regulation. The larger the output voltage ripple, the larger the line regulation, which means that the LED current changes if the input voltage changes. If a certain change in LED current gives a noticeable change in LED brightness, depends on the LED manufacturer and on the application. Applications requiring good line regulation ≤1%/V (typ) must use output capacitor values ≥1 µF. See Table 2 and Figure 22 to Figure 26 for the selection of the output capacitor. Assuming the converter does not show double pulses or pulse bursts (see Figure 13 and Figure 14) on the switch node (SW), the output voltage ripple is calculated as: 12 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION (continued) V I I 100 ns L LIM(min) 2 I I ESR V O 1 – P O ƒs C V V V O O F I with: ILIM(min) = minimum switch current limit (320 mA typ) L = selected inductor value IO = nominal load current fS = switching frequency at the nominal load current as calaculated with equation (5). VF = rectifier diode forward voltage (0.3 V typ) CO = selected output capacitor ESR = output capacitor ESR value Input Capacitor Selection For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 µF ceramic input capacitor is sufficient for most applications. For better input voltage filtering the capacitor value can be increased. Refer to Table 2 and Figure 22 to Figure 26 for input capacitor selection. Table 2. Possible Input and Output Capacitors (or equivalent) CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER 4.7 µF/X5R/0805 6.3 V Tayo Yuden JMK212BY475MG 10 µF/X5R/0805 6.3 V COMMENTS CI Tayo Yuden JMK212BJ106MG CI 100 nF Any CO 220 nF Any CO 470 nF Any CO 1.0 µF/X7R/1206 25 V Tayo Yuden TMK316BJ105KL CO 1.0 µF/X7R/1206 35 V Tayo Yuden GMK316BJ105KL CO 4.7 µF/X5R/1210 25 V Tayo Yuden TMK325BJ475MG CO Diode Selection To achieve high efficiency a Schottky diode must be used. The current rating of the diode must meet the peak current rating of the converter as it is calculated in the peak current control section. Use the maximum value for ILIM for this calculation. See Table 3 and Figure 22 to Figure 26 for the Schottky diode selection. Table 3. Possible Diodes (or equivalent) COMPONENT SUPPLIER REVERSE VOLTAGE ON Semiconductor MBR0530 30 V ON Semiconductor MBR0520 20 V Toshiba CRS02 30 V Zetex ZHCS400 40 V 13 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 Efficiency The overall efficiency of the application depends on the specific application conditions and mainly on the selection of the inductor. A lower inductor value increases the switching frequency and switching losses yielding in a lower efficiency. A lower inductor dc resistance has lower copper losses, giving a higher efficiency. Therefore, the efficiency can typically vary ±5% depending on the selected inductor. Figure 2 and Figure 3 can be used as a guideline for the application efficiency. These curves show the typical efficiency powering four LEDs using a 4.7 µH inductor with just 1,2 mm height. The efficiency curve in Figure 2 and Figure 3 show the efficiency delivering the power to the LEDs rather than the overall converter efficiency and is calculated as: V I LED LED V I I I Setting the LED Current The converter regulates the LED current by regulating the voltage across the current sense resistor (RS). The voltage across the sense resistor is regulated to the internal reference voltage of V(FB) = 252 mV. PWM 100 Hz to 50 kHz 3 VIN SW 8 5 7 CTRL OVP 6 1 GND LED 4 FB RS 2 Rs Figure 17. Setting the LED Current The LED current can be calculated: V I FB 0.252 V LED R R S S The current programming method is used when the brightness of the LEDs is fixed or controlled by a PWM signal applied to the CTRL pin. When using a PWM signal on the CTRL pin, the LED brightness is only dependent on the PWM duty cycle, independent of the PWM frequency, or amplitude, which simplifies the system. Analog Control Signal for Brightness Control Alternatively, an analog voltage can be used as well to control the LED brightness. 14 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION (continued) 3 Enable: CTRL = High Disable: CTRL = Low 5 VIN SW CTRL OVP 6 GND 4 LED FB RS 8 7 1 2 R1 PWM Signal VADJ (Brightness Control) R I1 Rs Vs R2 C Optional Filter for the use of a PWM Signal Figure 18. Setting the LED Current using an analog control signal In Figure 18 the LED current is determined by the voltage applied to R2 (VADJ) and the selection of R1, R2 and the sense resistor (RS). In this configuration, the LED current is linear controlled instead of pulsed as in the configuration before. To select the resistor values following steps are required. 1. Select the voltage VADJ(max) to turn the LEDs off, e.g. 3.3 V 2. Select the voltage VADJ(min) to turn the LEDs fully on, e.g. 0.0 V 3. Select the maximum and minimum LED current IO(max) and IO(min), e.g. IO(max) = 20 mA, IO(min) = 0 mA 4. Calculate R2 to achieve a feedback current in the range of I1 = 3 µA to 10 µA as the LEDs are fully turned on: V R2 ref V ADJ(min) I1 5. Calculate R1 I R1 V ref R2 V I R2 V O(max) ADJ(min) O(min) ADJ(max) V I V I V I V I ADJ(max) O(max) ref O(min) ADJ(min) O(min) ref O(max) V V 1 R1 R1 V S (V ref ADJ(min) R2 R2 6. Calculate the sense voltage S) at maximum LED current 7. Calculate the required sense resistor (RS) V R S I S O(max) PWM Control With Separate Enable The control pin (CTRL) combines the enable function as well as the PWM brightness control function in one pin. For some systems an independent enable function is required. One way to implement this is to use the brightness control configuration as shown in the previous section Figure 18. Other possible solutions are shown in Figure 19, Figure 20, Figure 21. 15 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 DETAILED DESCRIPTION (continued) PWM Brightness Control 100 Hz to 50 kHz Enable (EN) 3 VIN SW 8 5 7 CTRL OVP 6 1 GND LED 4 FB RS 2 Figure 19. Separate Enable and PWM Control Using a Schottky Diode PWM Brightness Control 100 Hz to 50 kHz 3 VIN SW 8 5 CTRL OVP 7 6 1 GND LED 4 FB RS 2 Enable (EN) Figure 20. Separate Enable and PWM Control Using a Transistor PWM Brightness Control 100 Hz to 50 kHz Enable (EN) 3 VIN SW 8 5 CTRL OVP 7 6 GND LED 1 4 2 FB RS Figure 21. Separate Enable and PWM Control Using an AND Gate Layout Considerations In all switching power supplies the layout is an important step in the design, especially at high peak currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems and duty cycle jitter. The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The inductor and diode must be placed as close as possible to the switch pin to minimize noise coupling into other circuits. It is important to connect the output capacitor directly across the diode cathode pin and ground rather than connecting the output capacitor across the LEDs. This minimizes EMI. Since the feedback pin and network is a high impedance circuit, the feedback network should be routed away from the inductor. Thermal Considerations The TPS61043 comes in a thermally enhanced QFN package. The package includes a thermal pad improving the thermal capabilities of the package. See the QFN/SON PCB Attachment application note (SLUA271). The thermal resistance junction to ambient RΘJA of the QFN package greatly depends on the PCB layout. Using thermal vias and wide PCB traces improves the thermal resistance RΘJA. Under normal operation conditions no PCB vias are required for the thermal pad. However, the thermal pad must be soldered to the PCB. 16 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 APPLICATION INFORMATION L1 4.7 H Coilcraft LPO1704-472 D1 ZHCS400 CO 1 F VIN 3.0 V to 6.0 V 3 VIN SW 8 5 CTRL OVP 7 C(IN) 4.7 F 6 GND 4 FB LED 1 RS 2 RS 13 Enable/PWM Brightness Control 100 HZ to 50 kHz Figure 22. TPS61043 With 1,0 mm Total System Height L1 4.7 H SUMIDA CMD4D11 D1 ZHCS400 CO 4.7 F VIN = 3.0 V to 6.0 V 3 5 C(IN) 4.7 F 6 4 Enable/PWM Brightness Control 100 HZ to 50 kHz VIN SW CTRL OVP GND LED FB RS 8 7 1 2 RS 13 Figure 23. TPS61043 With Low LED Ripple Current and Higher Accuracy Using a 4.7 µF Output Capacitor 17 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 Application Information (continued) L1 4.7 H Coilcraft LPO1704-472 D1 ZHCS400 CO 1 F VIN = 2.7 V to 6 V 3 VIN 5 CIN 4.7 F 6 CTRL SW 8 OVP 7 GND LED 4 FB 1 RS 2 RS 13 Enable/PWM Brightness Control 100 Hz to 50 kHz Figure 24. TPS61043 Powering 3 LEDs D1 ZHCS400 L1 4.7 H CO 100 nF VIN = 3.0 V to 6.0 V 3 VIN 5 CTRL C(IN) 4.7 F 6 4 Analog Brightness Control 3.3 V LED Off 0 V ILED = 20 mA GND SW 8 OVP 7 1 LED FB R1 10 k RS 2 RS 13 R2 120 k Figure 25. Adjustable Brightness Control Using an Analog Voltage 18 TPS61043 www.ti.com SLVS465A – DECEMBER 2003 – REVISED DECEMBER 2003 Application Information (continued) L1 4.7 H D1 ZHCS400 CO 100 nF VIN = 3.0 V to 6.0 V 3 VIN 5 CIN 4.7 F 6 CTRL SW 8 OVP 7 GND LED 4 FB 3.3 V PWM Signal 0 % LEDs on 100 % LEDs Off R1 10 k R C 1 RS 2 RS 13 R2 120 k Figure 26. 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