TI TPS61042DRB

TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
CONSTANT CURRENT LED DRIVER
FEATURES
DESCRIPTION
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The TPS61042 is a high frequency boost converter with
constant current output that drives white LEDs or similar.
The LED current is set with the external sense resistor
(RS) and is directly regulated by the feedback pin (FB)
that regulates the voltage across the sense resistor RS to
252 mV (typ). To control LED brightness, the LED
current can be pulsed by applying a PWM (pulse width
modulated) signal with a frequency range of 100 Hz to 50
kHz to the control pin (CTRL). To allow higher flexibility,
the device can be configured where the brightness can
be controlled by an analog signal as well, as described in
the application information section. To avoid possible
leakage currents through the LEDs during shutdown, the
control pin (CTRL) disables the device and disconnects
the LEDs from ground. For maximum safety during
operation, the output has integrated overvoltage
protection that prevents damage to the device in case of
a high impedance output (e.g. faulty LED).
Current Source With Overvoltage Protection
Input Voltage Range . . 1.8 V to 6.0 V
Internal 30 V Switch
Up to 85% Efficiency
Precise Brightness Control Using PWM Signal
or Analog Signal
Switching Frequency . . Up to 1 MHz
Internal Power MOSFET Switch . . 500 mA
Operates With Small Output Capacitors Down
to 100 nF
Disconnects LEDs During Shutdown
No Load Quiescent Current . . 38 µA Typ
Shutdown Current . . 0.1 µA Typ
Available in a Small 3 mm × 3 mm QFN
Package
APPLICATIONS
•
White LED Supply for Backlight/Sidelight
Displays
– PDA, Pocket PC, Smart Phones
– Handheld Devices
– Cellular Phones
L1
4.7 H
Vin
1.8 V to 6 V
Cin
4.7 F
Enable/PWM Brightness Control
100 Hz to 50 kHz
3
VIN
SW
D1
CO†
100 nF
8
5 CTRL OVP 7
6 GND LED 1
4 FB
RS 2
Rs
13 † Larger output capacitor values like 1 µF and larger, reduce the LED ripple current and improve line regulation.
Figure 1. Typical Application
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date. Products
conform to specifications per the terms of Texas Instruments standard warranty.
Production processing does not necessarily include testing of all parameters.
Copyright © 2003, Texas Instruments Incorporated
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
ORDERING INFORMATION (1)
(1)
TA
Package
Package Marking
-40 to 85°C
TPS61042DRB
BHS
The DRB package is available taped and reeled. Add R suffix (TPS61042DRBR) to order quantities of 3000 devices per reel. Add T
suffix (TPS61042DRBT) to order quantities of 250 devices per reel.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
TPS61042
Supply Voltages, v(VIN)
(2)
-0.3 V to 7 V
Voltages, V(Rs) , V(CTRL), V(FB)
-0.3 V to Vin + 0.3 V
Voltages, V(SW), V(LED) (2)
30 V
Voltage, V(OVP)
30 V
Continuous power dissipation
See Dissipation Rating Table
Operating junction temperature range
-40°C to 150°C
Storage temperature range, TSTG
-65°C to 150°C
Lead temperature (soldering, 10 sec)
(1)
(2)
260°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltage values are with respect to network ground terminal.
DISSIPATION RATING
(1)
PACKAGE
TA ≤ 25°C POWER
RATING
DERATING FACTOR
ABOVE TA=25°C
TA=70°C POWER
RATING
TA=85°C POWER
RATING
8 pin QFN (1)
370 mW
3.7 mW/°C
204 mW
148 mW
The thermal resistance junction to ambient of the 8-pin QFN package is 270 °C/W. Standard 2-layer PCB without vias for the thermal
pad. See the application section on how to improve the thermal resistance RθJA.
RECOMMENDED OPERATING CONDITIONS
MIN
TYP
MAX
UNIT
VI
Input voltage range
1.8
6.0
V
Vs
Output voltage range
VIN
27.5
V
VSW
Switch voltage
30
V
I(LED)
Maximum LED switch current
60
mA
L
Inductor (1)
f
Switching frequency (1)
CI
Input capacitor (1)
4.7
µF
CO
Output capacitor (1)
100
nF
TA
Operating ambient temperature
-40
85
°C
TJ
Operating junction temperature
-40
125
°C
(1)
2
See application section for further information
4.7
µH
1
MHz
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
ELECTRICAL CHARACTERISTICS
VI = 3.6 V, CTRL= VI, TA = -40°C to + 85°C, typical values are at TA= 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Supply current
VI
Input voltage range
1.8
6.0
V
I(Q)
Operating quiescent current into VIN
IO =0 mA, not switching
IO(sd)
Shutdown current
CTRL=GND
38
65
µA
0.1
1
µA
VUVLO
Under-voltage lockout threshold
VI falling
1.5
1.7
V
CTRL
VIH
CTRL high level input voltage
1.3
VIL
CTRL low level input voltage
IIkg
CTRL input leakage current
CTRL=GND or VIN
ton
Minimim CTRL pulse witdh to enable
CTRL=low to high
500
toff
Minimum CTRL pulse width to disable
CTRL=high to low
f(CTRL)
PWM switching frequency applied to
CTRL
D(CTRL)
PWM duty cycle applied to CTRL
V
0.3
V
0.1
µA
10
32
ms
0.1
50
kHz
1
100
%
us
Power switch and current limit (SW)
VS
Maximum switch voltage
rds(ON)
MOSFET on-resistance
VI=3.6 V; I(SW)=200 mA
Ilkg
MOSFET leakage current
V(SW)=28 V
ILIM
MOFSET current limit
300
400
30
V
600
mΩ
0.1
10
µA
500
600
mA
LED switch and current limit (LED)
VS
Maximum switch voltage
rds(ON)
MOSFET on-resistance
VI=3.6 V; IS =20 mA
Ilkg
MOSFET leakage current
V(LED)=28 V
30
V
1
2
Ω
0.1
10
µA
27.5
V
Output
VO
Output voltage range
VI
I(FB)
Feedback input bias current (1)
V(FB) =0.252 V
100
nA
VFB
Feedback trip point voltage
1.8 V ≤ VI≤ 6.0 V
244
252
260
mV
V(OVP)
Output overvoltage protection
VO rising
27.5
29
30
V
Vhys(OVP)
Output overvoltage protection hysteresis
I(OVP)
OVP input current
(1)
5
VO=15 V
9
7
V
12
µA
The feedback input is high impedance MOSFET Gate input.
3
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
DRB PACKAGE
(TOP VIEW)
LED 1
RS 2
VIN 3
8 SW
Exposed
Thermal
Die Pad
FB 4
7 OVP
6 GND
5 CTRL
NOTES:The exposed thermal die pad is connected to GND.
Terminal Functions
TERMINAL NAME NO.
4
I/O
DESCRIPTION
CTRL
5
I
Combined enable and PWM control pin. If CTRL is constantly pulled high, the device is enabled and the
internal LED switch (Q2) is constantly turned on. When CTRL is pulled to GND, the device is disabled.
Apply a PWM signal (100 Hz to 50 kHz) to this pin to control the brightness of the LEDs
FB
4
I
Feedback. FB regulates the LED current through the sense resistor by regulating the voltage across RS to
252 mV.
GND
6
LED
1
GND
I
Input of the LED switch (Q2). Connect the LEDs to this pin.
OVP
7
I
Overvoltage protection. OVP is connected to the output capacitor of the converter.
RS
2
O
Output of the internal LED switch. The sense resistor that programs the LED current is connected to RS.
SW
8
I
Drain of the integrated switch (Q1)
VIN
3
I
Input supply pin.
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
FUNCTIONAL BLOCK DIAGRAM
EN
SW
EN
VIN
UVLO
Bias
VREF
0.252 V
Control
Logic
Q1
Gate
Driver
Thermal
Shutdown
OVP
CTRL
Enable
Control
Logic
EN
R1
2 M
Current
Limit
Softstart
R2
30 k
6 s Max
On Time
GND
PWM
Gate
Drive
Overvoltage Protection
+
–
–
VREF
+
LED
Q2
Error
Comparator
FB
0.4 V
400 ns Min
Off Time
RS
5
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
FIGURE
η
Efficiency
vs Input voltage
Figure 3
IQ
Operating Quiescent Current into VIN
vs Input voltage and Temperature
Figure 4
V(FB)
Feedback voltage
vs Temperature
Figure 5
vs LED current
Figure 2
I(FB)
Feedback current
vs Temperature
Figure 6
rds(on)
Main switch Q1
vs Temperature
Figure 7
vs Input voltage
Figure 8
vs Temperature
Figure 9
vs Input voltage
Figure 10
vs PWM duty cycle on CTRL pin
Figure 11
LED switch Q2
ILED
Average LED current
Soft start
Figure 12
PFM operation (fixed peak current control)
Figure 13
Burst mode operation (fixed peak current
control)
Figure 14
PWM dimming
Figure 15
Efficiency
vs
LED Current
Efficiency
vs
Input Voltage
90
88
88
86
VI = 4.2V
84
IO = 15 mA
84
82
Efficiency – %
Efficiency – %
86
VI = 3.6V
80
78
VI = 2.4V
76
74
82
80
78
76
74
4 LEDs ≈ 13 V, CO = 1 µF
72
4 LEDs ≈ 13 V, CO = 1 µF
72
70
70
0.1
1
10
100
1
2
3
IO – Output Current – mA
Figure 2
Figure 3
Operating Quiescent Current into VIN
vs
Input Voltage and Temperature
Feedback Voltage
vs
Temperature
5
6
60
85
260
45
V(fb) - Feedback Voltage - mV
Quiescent Current into VI/µA
50
TA = 85°C
40
35
TA = 25°C
30
25
TA = - 40°C
20
15
10
5
0
258
VCC = 3.7 V
256
254
252
250
248
246
244
242
240
1.8
2.4
3.0
3.6
4.2
VI - Input Voltage - V
Figure 4
6
4
VI – Input Voltage – V
4.8
5.4
6.0
- 40
- 15
10
35
TA - Free-Air Temperature - ° C
Figure 5
TPS61042
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SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
Feedback Current
vs
Temperature
rds(on) Main Switch (Q1)
vs
Temperature
500
rds(on) - On-State Resistance - MΩ
I(fb) - Feedback Current - nA
60
40
VCC = 3.6 V
20
0
- 20
VCC = 5 V
VCC = 2.4 V
- 40
- 60
VCC = 3.6 V
450
400
350
300
250
200
- 40
- 15
10
35
60
85
- 40
- 15
TA - Free-Air Temperature - ° C
Figure 7
rds(ON) Main Switch (Q1)
vs
Input Voltage
rds(on) LED Switch (Q2)
vs
Temperature
60
85
60
85
1.6
rds(on) - On-State Resistance - Ω
rds(on) - On-State Resistance - MΩ
35
Figure 6
600
TA = 27° C
500
400
300
200
100
0
1.5
VCC = 3.6 V
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
1.8
2.4
3.0
3.6
4.2
4.8
5.4
6.0
- 40
- 15
10
35
TA - Free-Air Temperature - ° C
VI - Input Voltage - V
Figure 8
Figure 9
rds(on) LED Switch (Q2)
vs
Input Voltage
Average LED Current
vs
PWM Duty Cycle on CTRL Pin
3.0
20
TA = 25° C
2.5
IO - Output Current - mA
rds(on) - On-State Resistance - Ω
10
TA - Free-Air Temperature - ° C
2.0
1.5
1.0
0.5
0.0
15
10
fPWM = 50 kHz
fPWM = 100 Hz
5
fPWM = 25 kHz
0
1.8
2.4
3.0
3.6
4.2
VI - Input Voltage - V
Figure 10
4.8
5.4
6.0
0
20
40
60
80
100
Duty Cycle - %
Figure 11
7
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
PFM Operation
SOFTSTART
Vsw
5V/Div
Vout
10V/Div
Vout
500mV/Div
CTRL
1V/Div
LED Current
20mA/Div
Input Current
100mA/Div
50s/Div
Figure 13
Bust Mode Operation
PWM Dimming
Vsw
5V/Div
Vsw
5V/Div
Vout
50mV/Div
Vout
500mV/Div
LED Current
20mA/Div
LED Current
20mA/Div
2.5s/Div
Figure 14
8
2.5s/Div
Figure 12
25s/Div
Figure 15
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
DETAILED DESCRIPTION
OPERATION
The TPS61042 operates like a standard boost converter but regulates the voltage across the sense resistor (RS)
instead of the output voltage. This gives an accurate regulated LED current independent of the input voltage and
number of LEDs connected. With integrated overvoltage protection (OVP) the TPS61042 is configured as a
current source with overvoltage protection ideally suited to drive LEDs. With the 30V internal switch, the device
can generate output voltages of up to 27.5 V and has an internal 500mA MOSFET switch (Q1). This allows
several LEDs to be connected in series to the output. The internal LED switch (Q2) in series with the LEDs has a
maximum current rating of 60 mA and disconnects the LEDs from ground during shutdown. The LED switch is
driven by a PWM signal applied to the control pin (CTRL), which directly controls the LED brightness. With this
control method the LED brightness depends on the PWM duty cycle only and is independent of the PWM
frequency and amplitude.
BOOST CONVERTER
The boost converter operates in a pulse frequency modulation (PFM) scheme with constant peak current control.
This control scheme maintains high efficiency over the entire load current range and with a switching frequency
of up to 1 MHz, enables the use of small external components. The converter monitors the sense voltage across
RS with the feedback pin (FB) and, when the feedback voltage falls below the reference voltage (252 mV typ),
the main switch turns on and the current ramps up. The main switch turns off when the inductor current reaches
the internally set peak current of 500 mA (typ). Refer to the peak current control section for more information.
The second criteria that turns off the main switch is the maximum on-time of 6 µs (typ). This limits the maximum
on-time of the converter in extreme conditions. As the switch is turned off the external Schottky diode is forward
biased, delivering the stored inductor energy to the output. The main switch remains off until the minimum off
time of 400 ns (typ) has passed and the feedback voltage is below the reference voltage again. Using this PFM
peak current control scheme, the converter operates in discontinuous conduction mode (DCM) where the
switching frequency depends on the inductor, input and output voltage, and LED current. Lower LED currents
reduce the switching frequency, which results in high efficiency over the entire LED current range. This regulation
scheme is inherently stable, allowing a wide range for the selection of the inductor and output capacitor.
PEAK CURRENT CONTROL (BOOST CONVERTER)
The internal switch is turned on until the inductor current reaches the DC current limit (ILIM) of 500 mA (typ) . Due
to the internal current limit delay of 100 ns (typ) the actual current exceeds the DC current limit threshold by a
small amount. The typical peak current limit can be calculated:
V
I
I
I 100 ns
P(typ)
(LIM)
L
I
V
500 mA I 100 ns
P(typ)
L
The higher the input voltage and the lower the inductor value, the greater the current limit overshoot.
SOFTSTART
All inductive step-up converters exhibit high in-rush current during start-up if no special precautions are taken.
This can cause voltage drops at the input rail during start-up, which may result in an unwanted or premature
system shutdown.
The TPS61042 limits this in-rush current during start-up by increasing the current limit in two steps starting from
ILIM/4 for 256 switch cycles to ILIM/2 for the next 256 switch cycles and then full current limit. See Figure 12 for
typical start-up behavior.
9
TPS61042
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SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
CONTROL (CTRL)
The CTRL pin serves two functions. One is the enable and disable of the device. The other is the PWM control of
the internal LED switch (Q2). If no PWM signal is applied to the CTRL pin, then the CTRL pin can be used as a
standard enable pin for the device. To enable the device, the CTRL pin must be pulled high for time period of at
least 500 µs. The device starts with the softstart cycle. Pulling the CTRL pin to GND for a time period ≥32 ms
disables the device, disconnecting the LEDs from GND by opening the LED switch (Q2) to avoid any LED
leakage current. See Figure 16 for the CTRL pin timing.
The internal LED switch (Q2) is driven by the PWM signal when applied to the CTRL pin. Applying a PWM signal
in the range of 100 Hz to 50 kHz allows the LED current to be pulsed with the duty cycle of the PWM signal. The
CTRL pin accepts a PWM duty cycle from D = 1% to 100%. Duty cycles below 1% are also possible with the
restriction that the device is forced into shutdown as the off time of the applied PWM signal exceeds 10 ms.
When a PWM signal is applied to the CTRL pin the LED switch (Q2) turns on immediately. The internal error
comparator is disabled for 400 ns. This 400 ns delay time is required to establish the correct voltage level across
the sense resistor RS after the LED switch (Q2) is closed.
To achieve good LED current accuracy and linearity, the switching frequency of the converter must be higher
than the PWM frequency applied to the CTRL pin.
tp
toff
ton
ton
High
Low
Minimum
On-Time
to Enable
the Device
(500 s)
t
D = tp/t
Minimum
Off-Time
to Disable
the Device
(32 ms)
Figure 16. CTRL Timing Diagram
The CTRL timing diagram is shown in Figure 16. To enable the device, the CTRL signal must be high for 500 µs.
The PWM signal can then be applied with a pulse width (tp) greater or smaller than tON. To force the device into
shutdown mode, the CTRL signal must be low for at least 32 ms. Requiring the CTRL pin to be low for 32mS
before the device enters shutdown allows for PWM dimming frequencies as low as 100 Hz. The device is
enabled again when a CTRL signal is high for a period of 500 µs minimum. See Figure 11 for the PWM duty
cycle versus LED current characteristic.
This CTRL pin must be terminated.
OVERVOLTAGE PROTECTION (OVP)
As with any current source, the output voltage rises as the output impedance increases or is disconnected. To
prevent the output voltage from exceeding the maximum main switch (Q1) voltage rating of 30 V, an overvoltage
protection circuit is integrated. When the output voltage exceeds the OVP threshold voltage, (Q1) turns off. The
converter switch remains off until the output voltage falls below the OVP threshold voltage. As long as the output
voltage is below the OVP threshold the converter continues its normal operation, until the output voltage exceeds
the OVP threshold again.
10
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TPS61042
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
UNDERVOLTAGE LOCKOUT
An undervoltage lockout feature prevents mis-operation of the device at input voltages below 1.5 V (typ). As long
as the input voltage is below the undervoltage threshold the device remains off, with the main MOSFET switch
(Q1) and the LED switch (Q2) open.
THERMAL SHUTDOWN
An internal thermal shutdown is implemented in the TPS61042 that shuts down the device if the typical junction
temperature of 160°C is exceeded. If the device is in thermal shutdown mode, the main MOSFET switch (Q1)
and the LED switch (Q2) are open.
11
TPS61042
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SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
APPLICATION INFORMATION
INDUCTOR SELECTION, MAXIMUM LOAD CURRENT, AND SWITCHING FREQUENCY
The PFM peak current control scheme of the TPS61042 is inherently stable. The inductor value does not affect
the stability of the regulator. The selection of the inductor together with the nominal LED current, input, and
output voltage of the application determines the switching frequency of the converter.
The first step is to calculate the maximum load current the converter can support using the selected inductor.
The inductor value has less effect on the maximum available load current and is only of secondary order. A good
inductor value to start with is 4.7 µH. Depending on the application, inductor values down to 1.0 µH can be used.
The maximum inductor value is determined by the maximum on time of the switch of 6 µs (typ). The peak current
limit of 500 mA (typ) must be reached within this 6 µs for proper operation. The maximum load current of the
converter is determined at the operation point where the converter starts to enter the continuous conduction
mode. The converter must always operate in discontinuous conduction mode to maintain regulation.
Depending on the time period of the inductor current fall time being larger or smaller compared to the minimum
off time of the converter (400ns typ), the maximum load current can be calculated.
Inductor fall time:
ip L
t
fall
Vout–Vin
For tfall≥400ns
I
load max
ip Vin
2 Vout
for tfall≤ 400ns
I
load max
ip2 L Vin
(Vout Vin) (2 ip L 2 400 ns Vin)
with:
L = selected inductor value
η = expected converter efficiency. Typically between 70% to 85%
i p 400 mA Vin 100 ns
2
(Peak inductor current as described in the peak current control section)
The above formula contains the expected converter efficiency that allows calculating the expected maximum load
current the converter can support. The efficiency can be taken out of the efficiency graphs shown in Figure 2 and
Figure 3 or 80% can be used as an accurate estimation.
If the converter can support the desired LED current, the next step is to calculate the converter switching
frequency at the operation point, which must be ≤1 MHz. Also the converter switching frequency should be much
higher than the applied PWM frequency at the CTRL pin to avoid non-linear brightness control. Assuming the
converter shows no double pulses or pulse bursts (Figure 13, Figure 14) on the switch node (SW) the switching
frequency at the operation point can be calculated as:
2I
fs
(ILOAD)
LOAD
V
O
V V
I
F
V
I
I 100 ns
(LIM)
2
2
L
with:
I(LIM) = minimum switch current limit (500 mA typ)
L = selected inductor value
I(LOAD) = nominal load or LED current
VF = Rectifier diode forward voltage (typically 0.3 V)
12
TPS61042
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SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
The smaller the inductor value, the higher the switching frequency of the converter but the lower the efficiency.
The selected inductor must have a saturation current that meets the maximum peak current of the converter as
calculated in the peak current control section. Use the maximum value for I(LIM) (600mA) for this calculation.
Another important inductor parameter is the DC resistance. The lower the DC resistance the higher the efficiency
of the converter. See Table 1and Figure 22to Figure 27for inductor selection.
Table 1. Possible Inductor
Inductor Value
Component Supplier
Size
10 µH
muRata LQH43CN100K01
4,5 mm×3,2 mm×2.6 mm
4.7 µH
muRata LQH32CN4R7M11
3,2 mm×2,5 mm×2,0 mm
10 µH
Coilcraft DO1605T-103MX
5,5 mm ×4,1 mm ×1,8 mm
4.7 µH
Sumida CDRH3D16-4R7
3,8 mm×3,8 mm×1,8 mm
3.3 µH
Sumida CMD4D11- 3R3
3,5 mm×5,3 mm×1,2 mm
4.7 µH
Sumida CMD4D11- 4R7
3,5 mm×5,3 mm×1,2 mm
3.3 µH
Sumida CMD4D11- 3R3
3,5 mm×5,3 mm×1,2 mm
4.7 µH
Coiltronics SD12-4R7
5,2 mm×5,2 mm×1,2 mm
3.3 µH
Coilcraft LPO1704-332M
6,6 mm×5,5 mm×1,0 mm
4.7 µH
Coilcraft LPO1704-472M
6,6 mm×5,5 mm×1,0 mm
output capacitor selection and line regulation
For better output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value, but depending on the application, tantalum capacitors can be used.
The selection of the output capacitor value directly influences the output voltage ripple of the converter which
also influences line regulation. The larger the output voltage ripple, the larger the line regulation, which means
that the LED current changes if the input voltage changes. If a certain change in LED current gives a noticeable
change in LED brightness, depends on the LED manufacturer and on the application. Applications requiring good
line regulation ≤1%/V (typ) must use output capacitor values ≥1 µF.
See Table 2 and Figure 22 to Figure 27 for the selection of the output capacitor.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output
voltage ripple is calculated as (see Figure 13, Figure 14):
V
I
I 100 ns L
(LIM)
2
I
I ESR
1 V O P
O
C
f
V V V
O s(ILOAD)
O
F
I
with:
I(LIM) = minimum switch current limit (400 mA typ)
L = selected inductor value
I(LOAD) = nominal load current
fS = switching frequency at the nominal load current as calaculated before.
VF = rectifier diode forward voltage (0.3 V typ)
CO = selected output capacitor
ESR = output capacitor ESR value
13
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
INPUT CAPACITOR SELECTION
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 µF ceramic input capacitor
is sufficient for most applications. For better input voltage filtering the capacitor value can be increased. Refer to
Table 2 and Figure 22 to Figure 27 for input capacitor selection.
Table 2. Possible Input and Output Capacitors
Capacitor
Voltage rating
Component Supplier
Comments
4.7 µF/X5R/0805
6.3 V
Tayo Yuden JMK212BY475MG
CI
10 µF/X5R/0805
6.3 V
Tayo Yuden JMK212BJ106MG
CI
100 nF
Any
CO
220 nF
Any
CO
470 nF
Any
CO
1.0 µF/X7R/1206
25 V
Tayo Yuden TMK316BJ105KL
CO
1.0 µF/X7R/1206
35 V
Tayo Yuden GMK316BJ105KL
CO
4.7 µF/X5R/1210
25 V
Tayo Yuden TMK325BJ475MG
CO
DIODE SELECTION
To achieve high efficiency a Schottky diode must be used. The current rating of the diode must meet the peak
current rating of the converter as it is calculated in the peak current control section. Use the maximum value for
I(Lim) for this calculation. See Table 3 and Figure 22 to Figure 27 for the Schottky diode selection.
Table 3. Possible Diodes
Component Supplier
Reverse voltage
ON Semiconductor MBR0530
30 V
ON Semiconductor MBR0520
20 V
Toshiba CRS02
30 V
Zetex ZHCS400
40 V
EFFICIENCY
The overall efficiency of the application depends on the specific application conditions and mainly on the
selection of the inductor. A lower inductor value increases the switching frequency and switching losses yielding
in a lower efficiency. A lower inductor dc resistance has lower copper losses, giving a higher efficiency.
Therefore, the efficiency can typically vary ±5% depending on the selected inductor. Figure 2 and Figure 3 can
be used as a guideline for the application efficiency. These curves show the typical efficiency powering four
LEDs using a 4.7 µH inductor with just 1,2 mm height. The efficiency curve in Figure 2 and Figure 3 show the
efficiency delivering the power to the LEDs rather than the overall converter efficiency and is calculated as:
V
I
LED
LED
V I
I
I
SETTING THE LED CURRENT
The converter regulates the LED current by regulating the voltage across the current sense resistor (RS). The
voltage across the sense resistor is regulated to the internal reference voltage of V(FB) = 252 mV.
14
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
VIN
PWM
100 Hz to 50 kHz
SW
CTRL OVP
GND
FB
LED
RS
Rs
Figure 17. Setting the LED Current
The LED current can be calculated:
V
I
FB 0.252 V
LED
R
R
S
S
The current programming method is used when the brightness of the LEDs is fixed or controlled by a PWM
signal applied to the CTRL pin. When using a PWM signal on the CTRL pin, the LED brightness is only
dependent on the PWM duty cycle, independent of the PWM frequency, or amplitude, which simplifies the
system.
Alternatively, an analog voltage can be used as well to control the LED brightness.
VIN
Enable: CTRL = High
Disable: CTRL = Low
SW
CTRL OVP
LED
GND
FB
RS
R1
PWM
Signal
VADJ
(Brightness Control)
R
I1
Rs
Vs
R2
C
Optional Filter for the
use of a PWM Signal
Figure 18. Setting the LED Current
15
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
In Figure 18 the LED current is determined by the voltage applied to R2 (V(adj)) and the selection of R1, R2 and
the sense resistor (RS). In this configuration, the LED current is linear controlled instead of pulsed as in the
configuration before. To select the resistor values following steps are required.
Select the voltage V(adjmax) to turn the LEDs off. → V(adjmax) (e.g. 3.3 V)
Select the voltage V(adjmin) to turn the LEDs fully on. → V(adjmin) (e.g. 0.0 V)
Select the maximum and minimum LED current IO(max) and IO(min). → (e.g. IO(max) = 20 mA, IO(min) = 0 mA)
Calculate R2 to achieve a feedback current in the range of I1 = 3 µA to 10 µA as the LEDs are fully turned
on:
•
•
•
•
V
R2 •
ref
V
adj(min)
I1
Calculation of R1
I
R1 V
ref
R2 V
I
R2 V
O(max)
adj(min)
O(min)
adj(max)
V
I
V I
V
I
V I
adj(max)
O(max)
ref
O(min)
adj(min)
O(min)
ref
O(max)
•
Calculation of the sense voltage (VS) at maximum LED current
V V 1 R1 – R1 V
S
ref
adj(min)
R2 R2
•
Calculation of the required sense resistor (RS)
V
R S
I
16
S
O(max)
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
PWM CONTROL WITH SEPARATE ENABLE
The control pin (CTRL) combines the enable function as well as the PWM brightness control function in one pin.
For some systems an independent enable function is required. One way to implement this is to use the
brightness control configuration as shown in the previous section Figure 18.
Other possible solutions are shown in Figure 19, Figure 20, Figure 21.
VIN
PWM Brightness Control
100 Hz to 50 kHz
SW
CTRL OVP
LED
GND
Enable (EN)
FB
RS
Figure 19. Separate Enable and PWM Control Using a Schottky Diode
VIN
PWM Brightness Control
100 Hz to 50 kHz
SW
CTRL OVP
GND
Enable (EN)
LED
FB
RS
Figure 20. Separate Enable and PWM Control Using a Transistor
PWM Brightness Control
100 Hz to 50 kHz
VIN
SW
CTRL OVP
Enable (EN)
GND
FB
LED
RS
Figure 21. Separate Enable and PWM Control Using an AND Gate
Layout Considerations
In all switching power supplies the layout is an important step in the design, especially at high peak currents and
switching frequencies. If the layout is not carefully done, the regulator might show noise problems and duty cycle
jitter.
The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The
inductor and diode must be placed as close as possible to the switch pin to minimize noise coupling into other
circuits. Since the feedback pin and network is a high impedance circuit, the feedback network should be routed
away from the inductor.
THERMAL CONSIDERATIONS
The TPS61042 comes in a thermally enhanced QFN package. The package includes a thermal pad improving
the thermal capabilities of the package. See the QFN/SON PCB Attachment application note (SLUA271).
The thermal resistance junction to ambient RΘJA of the QFN package greatly depends on the PCB layout. Using
thermal vias and wide PCB traces improves the thermal resistance RΘJA. Under normal operation conditions no
PCB vias are required for the thermal pad. However, the thermal pad must be soldered to the PCB.
17
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
L1
4.7 H
Coilcraft LPO1704–472
D1
ZHCS400
CO
1 F
VIN
1.8 V to 6 V
VIN
C(IN)
4.7 F
SW
CTRL OVP
GND
FB
LED
RS
RS
13 Enable/PWM Brightness
Control 100 HZ to 50 kHz
Figure 22. TPS61042 With 1,0 mm Total System Height. Efficiency = 82.7%@VI = 3.0 V/19 mA
L1
4.7 H
SUMIDA CMD4D11
VIN
1.8 V to 6 V
C(IN)
4.7 F
CO
4.7 F
VIN
SW
CTRL
OVP
GND
LED
FB
Enable/PWM Brightness
Control 100 HZ to 50 kHz
D1
ZHCS400
RS
RS
13 Figure 23. TPS61042 With Low LED Ripple Current and Higher Accuracy Using a 4.7 µF Output Capacitor
18
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
L1
D1
Sumida CDRH3D16–6R8
Zetex ZHCS400
6.8 H
CO
1 F
VIN
2.7 V to 6 V
C(IN)
4.7 F
VIN
SW
CTRL
OVP
GND
LED
FB
RS
RS
13 Enable/PWM Brightness
Control 100 HZ to 50 kHz
Figure 24. TPS61042 Powering 6 LEDs, Efficiency = 84%@VI = 3.6 V/19 mA
L1
10.0H
Coilcraft DO1605T–103MX
VIN
2.7 V to 6 V
CIN
4.7 F
CO
100 nF
VIN
SW
CTRL OVP
GND
FB
Enable/PWM Brightness
Control 100 HZ to 50 kHz
D1
Zetex ZHCS400
LED
Rs
RS
6.5 R1
30
R2
30
Figure 25. TPS61042 Powering 8 LEDs, Efficiency = 81%@VI = 3.6 V/18.6 mA
19
TPS61042
www.ti.com
SLVS441B — DECEMBER 2002 — REVISED JANUARY 2003
D1
ZHCS400
L1
4.7 H
CO
100 nF
VCC = 1.8 V to 6 V
C(IN)
4.7 F
VIN
SW
CTRL
OVP
GND
LED
RS
FB
R1
10 k
Analog Brightness Control
3.3 V LED Off
0 V ILED = 20 mA
RS
13 R2
120 k
Figure 26. Adjustable Brightness Control Using an Analog Voltage
L1
4.7 H
D1
ZHCS400
CO
100 nF
VCC = 1.8 V to 6 V
VIN
SW
CTRL OVP
CIN
4.7 F
GND
LED
FB
3.3 V PWM Signal
0 % LEDs on
100 % LEDs Off
R1
10 k
R
C
RS
RS
13 R2
120 k
Figure 27. Alternative Adjustable Brightness Control Using PWM Signal
20
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