TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 DC-DC INVERTER FEATURES • • • • • • • • • • DESCRIPTION Adjustable Output Voltage Down to –15 V 2.7-V to 5.5-V Input Voltage Range Up to 360-mA Output Current 1000-mA Typical Switch Current Limit Up to 84% Efficiency Typical 1.4-MHz Fixed-Frequency PWM Operation Thermal Shutdown Typical –19 V Output Overvoltage Protection 1.5-µA Shutdown Current Small 3-mm x 3-mm SON-10 Package (DRC) The inverter operates with a fixed-frequency PWM control topology. The device has an internal current limit, overvoltage protection, and a thermal shutdown for highest reliability under fault conditions. APPLICATIONS • • • • The TPS63700 is an inverting dc-dc converter generating a negative output voltage down to –15 V with output currents up to 360-mA, depending on input-voltage to output-voltage ratio. With a total efficiency up to 84%, the device is ideal for portable battery-powered equipment. The input voltage range of 2.7-V to 5.5-V allows the TPS63700 to be directly powered from a Li-ion battery, from 3-cell NiMH/NiCd, from a 3.3-V or 5-V supply rail. The TPS63700 comes in a small 3-mm x 3-mm SON-10 package. Furthermore, the high switching frequency of typically 1.4 MHz allows the use of small external components. This, and the small package make a small power supply solution possible. Generic Negative Voltage Supply Small-to-Medium Size OLED Displays PDAs, Pocket PCs, Smartphones Bias Supply TPS63700 C2 VIN R1 C1 0.1 F R2 VREF 0.22 F EN FB R3 OUT PS_GND D1 IN VIN 2.7 V To 5.5 V C4 10 F GND VOUT −5 V SW PowerPAD COMP L1 4.7 H C5 22 F C6 4.7 nF Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2005, Texas Instruments Incorporated TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION (1) (1) (2) TA SWITCH CURRENT LIMIT PACKAGE TYPE SYMBOL PART NUMBER (2) –40°C to 85°C 1000 mA SON-10 NUB TPS63700DRC For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI Web site at www.ti.com. The DRC package is available taped and reeled. Add an R suffix to the device type (i.e., TPS63700DRCR) to order quantities of 3000 devices per reel. Add a T suffix to the device type (i.e., TPS63700DRCT) to order quantities of 250 devices peer reel. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range unless otherwise noted (1) TPS63700 Input voltage range at VIN (2) –0.3 V to +6.0 V Input voltage range at IN (2) Minimum voltage at VOUT VIN (2) Voltage at EN, FB, COMP, PS –18 V (2) –0.3 V to VIN + 0.3 V Differential voltage between OUT to VIN (2) 24 V Operating virtual junction temperature, TJ –40°C to 150°C Storage temperature range, TSTG –65°C to 150°C (1) (2) Stresses beyond those listed under "absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal, unless otherwise noted. DISSIPATION RATINGS TABLE (1) (1) PACKAGE TA≤ 25°C POWER RATING DRC 2053 mW DERATING FACTOR TA = 70°C ABOVE TA = 25°C POWER RATING 21 mW/°C 1130 mW TA = 85°C POWER RATING 821 mW The thermal resistance junction to ambient of the 10-pin DRC is ΘJA = 48.7 °C/W. Exceeding the maximum junction temperature forces the device into thermal shutdown. RECOMMENDED OPERATING CONDITIONS MIN NOM MAX UNIT Input voltage range, VI 2.7 5.5 V Operating free-air temperature range, TA –40 85 °C Operating virtual junction temperature range, TJ –40 125 °C 2 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 ELECTRICAL CHARACTERISTICS –40°C to 85°C, over recommended input voltage range, typical at an ambient temperature of 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT DC-DC STAGE VOUT Adjustable output voltage range VIN Input voltage range PIN VIN, IN 2.7 VREF Reference voltage IREF = 10 µA 1.2 IFB Negative feedback input bias current VFBN = 0.1 VREF VFB Negative feedback regulation voltage VIN = 2.7 V to 5.5 V VOUT DC output accuracy PWM mode, device switching, VOVP Output overvoltage protection RDS(ON) Inverter switch on-resistance VIN = 3.6 V 440 600 VIN = 5 V 370 500 ILIM Inverter switch current limit 2.7 V < VIN < 5.5 V 1000 1140 mA DMAX Maximum duty cycle inverting converter 87.5% DMIN Minimum duty cycle inverting converter 12.5% 1500 kHz –15 1.213 –2 V 5.5 V 1.225 V 2 –0.024 860 0 nA 0.024 V ±3 % –19 V mΩ CONTROL STAGE fS Oscillator frequency VEN High level input voltage VEN Low level input voltage IEN Input current VIN I(Q) Quiescent current ISD Shutdown supply current VUVLO Undervoltage lockout threshold IN 1250 V 0.4 V EN = VIN or GND 0.01 0.1 µA VIN = 3.6 V, IOUT = 0, EN = VIN, no switching VOUT = –5 V 330 400 µA 640 750 µA EN = GND 0.2 1.5 µA 2.35 2.7 V 2.1 Thermal shutdown Thermal shutdown hysteresis 1380 1.4 Junction temperature decreasing 150 °C 5 °C 3 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 PIN ASSIGNMENTS DRC PACKAGE PowerPAD™ (TOP VIEW) COMP 1 10 GND 2 9 FB VIN 3 8 OUT EN 4 7 PS_GND IN 5 6 SW PowerPAD VREF Terminal Functions TERMINAL I/O DESCRIPTION NAME NO. COMP 1 I/O EN 4 I Enable pin (EN=GND: disabled; EN=VIN: enabled) FB 9 I Feedback pin for the voltage divider GND 2 IN 5 I supply voltage for the power switch OUT 8 I Output voltage sense input PS_GND 7 I Connect to GND for control logic SW 6 O Inverter switch output VIN 3 I supply voltage for control logic, connect a RC filter of 10R and 100nF VREF 10 O Reference voltage output. Connect a 220-nF capacitor to ground. Connect the lower resistor of the negative output voltage divider to this pin. 4 Compensation pin for control, connect a 4.7nF capacitor between this pin and GND Ground pin TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 FUNCTIONAL BLOCK DIAGRAM VIN VIN VIN Temperature GND Oscillator Control VIN PS_GND OUT Control Logic EN − COMP FB + VREF Gate IN IN Control + − SW 5 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 TYPICAL CHARACTERISTICS PARAMETER MEASUREMENT INFORMATION TPS63700 C2 VIN VREF EN FB R2 C3 0.22 F 10 C1 0.1 F PS_GND R3 OUT D1 IN VIN C4 10 F SW R4 100 k VOUT, −5 V SL02/SL03 GND PowerPAD COMP L1 C6 4.7 nF List of Components REFERENCE C1, C2, C3, C4, DESCRIPTION X7R/X5R ceramic C5 4 x 4.7 µF X7R/X5R ceramic D1 SL03/SL02 Vishay L1 –5V: TDK VLF4012 4R7, TDK SLF6025-4R7, Coilcraft LPS4018-472, –12V: Sumida CDRH5D18 10 µH 6 10 pF C5 4x4.7 F TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 Table of Graphs GRAPH DESCRIPTION Figure 1 Maximum output current versus input voltage, VOUT = –5 V, –12 V, –15 V Figure 2 Efficiency versus output current, VOUT = –5 V Figure 3 Efficiency versus output current, VOUT = –12 V Figure 4 Efficiency versus output current, VOUT = –15V Figure 5 Efficiency versus input voltage, VOUT = –5 V Figure 6 Efficiency versus input voltage, VOUT = –12 V Figure 7 Output voltage versus output current, VOUT = –5 V Figure 8 Output voltage versus output current, VOUT = –12 V Figure 9 Output voltage in discontinuous conduction mode, VIN= 3.6 V, VOUT = –5 V Figure 10 Output voltage in continuous conduction mode, VIN= 3.6 V, VOUT = –5 V Figure 11 Load transient response, VIN= 3.6 V, VOUT = –5 V, 45 to 150 mA Figure 12 Line transient response, VIN= 3.6 V to 4.2 V, VOUT = –5 V Figure 13 Start-up after enable,VI = 3.6 V, VOUT = –5 V PERFORMANCE GRAPHS MAXIMUM OUTPUT CURRENT vs INPUT VOLTAGE EFFICIENCY vs OUTPUT CURRENT, VOUT –5V 90 400 VIN = 5 V 80 350 VO = −5 V VIN = 3.3 V 70 300 VIN = 4.2 V 60 250 Efficiency % Maximum Output Current − mA VIN = 3.6 V VO = −12 V 200 VO = −15 V 150 50 40 30 100 20 50 10 VOUT = −5 V 0 2.5 0 3 3.5 4 4.5 5 5.5 0 100 200 300 VI − Input Voltage − V IO − Output Current − mA Figure 1. Figure 2. 400 7 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 PERFORMANCE GRAPHS (continued) EFFICIENCY vs OUTPUT CURRENT, VOUT –12 V EFFICIENCY vs OUTPUT CURRENT, VOUT –15 V 90 90 VIN = 5 V 80 80 VIN = 4.2 V 70 VIN = 3.6 V VIN = 3.3 V 70 Efficiency % 60 Efficiency % VIN = 5 V 50 40 VIN = 3.3 V VIN = 4.2 V 60 50 40 30 30 20 20 10 10 VOUT = −15 V VOUT = −12 V 0 0 0 100 50 150 200 250 0 20 40 IO − Output Current − mA Figure 3. Figure 4. EFFICIENCY vs INPUT VOLTAGE, VOUT –5 V EFFICIENCY vs INPUT VOLTAGE, VOUT –12 V 90 90 IOUT = 200 mA IOUT = 50 mA 80 IOUT = 150 mA IOUT = 50 mA 80 IOUT = 20 mA IOUT = 20 mA 70 70 60 Efficiency % 60 Efficiency % 60 80 100 120 140 160 180 200 IO − Output Current − mA 50 40 50 40 30 30 20 20 10 10 VOUT = −12 V VOUT = −5 V 0 2.5 3 3.5 4 4.5 VIN − Input Voltage − V Figure 5. 8 5 5.5 0 2.5 3 3.5 4.5 4 VIN − Input Voltage − V Figure 6. 5 5.5 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 PERFORMANCE GRAPHS (continued) OUTPUT VOLTAGE vs OUTPUT CURRENT OUTPUT VOLTAGE vs OUTPUT CURRENT −5.1 −12.4 VOUT = −5 V VOUT = −12 V VOUT− Output Voltage − V VOUT− Output Voltage − V −12.3 VIN = 5 V −5.05 −5 VIN = 3.6 V VIN = 3.3 V −4.95 −12.2 VIN = 5 V −12.1 VIN = 3.6 V −12 VIN = 3.3 V −11.9 −11.8 −4.9 0 −11.7 50 100 150 200 250 300 350 400 0 50 100 150 200 IOUT − Output Current − mA IOUT − Output Current − mA Figure 7. Figure 8. OUTPUT VOLTAGE IN DISCONTINUOUS CONDUCTION MODE OUTPUT VOLTAGE IN CONTINUOUS CONDUCTION MODE VIN = 3.6 V, ILOAD = 20 mA VOUT = –5 V 250 VIN = 3.6 V, ILOAD = 95 mA VOUT 20 mV/div, AC ICOIL 200 mA/div, DC t - Time - 500 ns/div Figure 9. VOUT 20 mV/div, AC ICOIL 200 mA/div, DC VOUT = –5 V t - Time - 500 ns/div Figure 10. 9 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 PERFORMANCE GRAPHS (continued) LOAD TRANSIENT RESPONSE, –5 V, 45 TO 150 mA VIN = 3.6 V, ILOAD = 45 mA to 150 mA LINE TRANSIENT RESPONSE, –5 V VIN = 3.6 V to 4.2 V, ILOAD = 100 mA, VOUT = –5 V 4.2 V VOUT 100 mV/div, AC VIN 500 mV/div, DC 3.6 V VOUT 100 mV/div, DC VOUT = –5 V ILOAD 50 mA/div, DC t - Time - 2 ms/div t - Time - 2 ms/div Figure 11. Figure 12. START-UP AFTER ENABLE, –5 V EN 2 V/div, DC VIN = 3.6 V, Load = 22 W, VOUT = –5 V ICOIL 200 mA/div, DC VOUT 2 V/div, DC t - Time - 500 ms/div Figure 13. 10 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 DETAILED DESCRIPTION The TPS63700 is a dc-dc converter for negative output voltages using buck-boost topology. It operates with an input voltage range of 2.7 V to 5.5 V and generates a negative output voltage down to –15 V. The output is controlled by a fixed-frequency, pulse-width-modulated (PWM) regulator. In normal operation mode, the converter operates at continuous conduction mode (CCM). At light loads it can enter discontinuous conduction mode (DCM). Power Conversion The converter operates in a fixed-frequency, pulse-width-modulated control scheme. So, the on-time of the switches varies depending on input-to-output voltage ratio and the load. During this on-time, the inductor connected to the converter is charged with current. In the remaining time, the time period set by the fixed operating frequency, the inductor discharges into the output capacitor via the rectifier diode. Usually, at higher loads the inductor current is continuous. During light load, the inductor current of this converter can become discontinuous. In this case, the control circuit of the controller output automatically takes care of these changing conditions to always operate with an optimum control setup. Control The controller circuit of the converter is based on a fixed-frequency, multiple-feedforward controller topology. Input voltage, output voltage, and voltage drop across the switch are monitored and forwarded to the regulator. Changes in the operating conditions of the converter directly affect the duty cycle. The error amplifier compares the voltage on FB pin with GND to generate an accurate and stable output voltage. The error amplifier is internally compensated. At light loads, the converter operates in discontinuous conduction mode (DCM). If the load will be further decreased, the energy transmitted to the output capacitor can't be absorbed by the load and would lead to an increase of the output voltage. In this case, the converter limits the output voltage increase by skipping switch pulses. Enable Applying GND signal at the EN pin disables the converter, where all internal circuitry is turned off. The device now just consumes low shutdown current flowing into the VIN pin. The output load of the converter is also disconnected from the battery as described in the following paragraph. Pulling the EN pin to VIN enables the converter. Internal circuitry, necessary to operate the converter, is then turned on. Load Disconnect The device supports complete load disconnection when the converter is disabled. The converter turns off the internal PMOS switch, thus no DC current path remains between load and input voltage source. Soft Start The converter has a soft-start function. When the converter is enabled, the implemented switch current limit ramps up slowly to its nominal value. Soft start is implemented to limit the input current during start-up to avoid high peak currents at the battery which could interfere with other systems connected to the same battery. Without soft start, uncontrolled input peak currents flow to charge up the output capacitors and to supply the load during start-up. This would cause significant voltage drops across the series resistance of the battery and its connections. Output Overvoltage Protection The converter has an implemented output overvoltage protection. The output voltage is limited to –19 V in case the feedback connection from the output to the FB pin is open. 11 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 DETAILED DESCRIPTION (continued) Undervoltage Lockout An undervoltage lockout prevents the device from starting up and operating if the supply voltage at VIN is lower than the programmed threshold shown in the electrical characteristics table. The device automatically shuts down the converter when the supply voltage at VIN falls below this threshold. Nevertheless, parts of the control circuits remain active, which is different than device shutdown using EN inputs. The undervoltage lockout function is implemented to prevent device malfunction. Overtemperature Shutdown The device automatically shuts down if the implemented internal temperature detector detects a chip temperature above the programmed threshold shown in the electrical characteristics table. It starts operating again when the chip temperature decreases. A built-in temperature hysteresis avoids undefined operation caused by ringing from overtemperature shutdown. 12 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 APPLICATION INFORMATION Design Procedure The TPS63700 dc-dc converter is intended for systems typically powered by a single-cell Li-ion or Li-polymer battery with a terminal voltage between 2.7 V up to 4.2 V. Due to the recommended input voltage going up to 5.5 V, the device is also suitable for 3-cell alkaline, NiCd, or NiMH batteries, as well as regulated supply voltages of 3.3 V or 5 V. TPS63700 R2 150 kW C2 VIN VREF EN FB 0.22 mF 10 W C1 0.1 mF OUT PS_GND VIN 2.7 V To 5.5 V D1 IN C3 10pF R3 680 kW SW R4 100 kW VOUT, –5V SL02 C4 10 mF GND PowerPAD COMP C5 4x4.7 mF L1 4.7 mH C6 4.7 nF Figure 14. Circuit for –5 Volt Output TPS63700 C2 10 W C1 0.1 mF VIN VREF EN FB 0.22 mF OUT PS_GND VIN 2.7 V To 5.5 V C4 10 mF GND D1 SW IN PowerPAD COMP SL03 L1 10 mH R2 121 kW C3 10pF R3 1.2 MW R4 100 kW VOUT, –12V C5 4x4.7 mF C6 4.7 nF Figure 15. Circuit for –12 Volt Output Programming the Output Voltage Converter The output voltage of the TPS63700 converter can be adjusted with an external resistor divider connected to the FB pin. The reference point of the feedback divider is the reference voltage VREF with 1.213 V. The typical value of the voltage at the FB pin is 0 V. The minimum recommended output voltage at the converter is –15 V. The feedback divider current should be 10 µA. The voltage across R2 is 1.213 V. Based on those values, the recommended value for R2 should be 120 kΩ to 200 kΩ in order to set the divider current at the required value. The value of the resistor R3 can then be calculated using Equation 1, depending on the needed output voltage (VOUT): 13 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 APPLICATION INFORMATION (continued) V R3 R2 REF V V REF OUT 1 (1) For example, if an output voltage of –5 V is needed and a resistor of 150 kΩ has been chosen for R2, a 680-kΩ resistor is needed to program the desired output voltage. Inductor Selection An inductive converter normally requires two main passive components for storing energy during the conversion. An inductor and a storage capacitor at the output are required. The average inductor current depends on the output load, the input voltage (VIN), and the output voltage VOUT. It can be estimated with Equation 2, which shows the formula for the inverting converter. V V OUT I I IN Lavg OUT V 0.8 IN (2) with: ILavg= average inductor current An important parameter for choosing the inductor is the desired current ripple in the inductor. A ripple current value between 20% and 80% of the average inductor current can be considered as reasonable, depending on the application requirements. A smaller ripple reduces the losses in the inductor, as well as output voltage ripple and EMI. But in the same way, the inductor becomes larger and more expensive. Keeping those parameters in mind, the possible inductor value can be calculated using Equation 3. V V IN OUT L V f I V OUT IN L (3) with: ∆IL = peak-to-peak ripple current f = switching frequency L = inductor value With the known inductor current ripple, the peak inductor value can be approximated with Equation 4. The peak current through the switch and the inductor depends also on the output load, the input voltage (VIN), and the output voltage (VOUT). To select the right inductor, it is recommended to keep the possible peak inductor current below the current-limit threshold of the power switch. For example, the current-limit threshold of the TPS63700 switch for the inverting converter is nominally 1000 mA. V V I OUT I I IN L Lmax OUT 2 V 0.8 IN (4) with: ILMAX = peak inductor current With Equation 5, the inductor current ripple at a given inductor can be approximated. V V IN OUT I L V f L V OUT IN (5) Care has to be taken for the possibility that load transients and losses in the circuit can lead to higher currents as estimated in Equation 4. Also, the losses caused by magnetic hysteresis losses and copper losses are a major parameter for total circuit efficiency. 14 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 APPLICATION INFORMATION (continued) The following inductor series from different suppliers have been tested with the TPS63700 converter: List of Inductors Output Voltage Vendor SUGGESTED INDUCTOR VLF4012 4.7 µH –5V TDK –5V Coilcraft –12V Sumida CDRH5D18 10 µH –12V Coilcraft MOS6020 10 µH SLF6025-4.7 µH LPS4018 4.7 µH LPS3015 4.7 µH Capacitor Selection Input Capacitor At least a 10-µF ceramic input capacitor is recommended for a good transient behavior of the regulator, and EMI behavior of the total power supply circuit. Output Capacitors One of the major parameters necessary to define the capacitance value of the output capacitor is the maximum allowed output voltage ripple of the converter. This ripple is determined by two parameters of the capacitor, the capacitance and the ESR. It is possible to calculate the minimum capacitance needed for the defined ripple, supposing that the ESR is zero, by using Equation 6 for the inverting converter output capacitor. I V OUT OUT C min V f V V OUT IN S (6) Parameter f is the switching frequency and ∆V is the maximum allowed ripple. With a chosen ripple voltage in the range of 10 mV, a minimum capacitance of 12 µF is needed. The total ripple is larger due to the ESR of the output capacitor. This additional component of the ripple can be calculated using Equation 7 . V I R ESR OUT ESR (7) An additional ripple of 2 mV is the result of using a typical ceramic capacitor with an ESR in a 10-mΩ range. The total ripple is the sum of the ripple caused by the capacitance, and the ripple caused by the ESR of the capacitor. In this example, the total ripple is 12 mV. Additional ripple is caused by load transients. When the load current increases rapidly, the output capacitor must provide the additional current until the inductor current has been increased by the control loop by setting a higher on-time at the main switch (duty cycle). The higher duty cycle results in longer inductor charging periods. But the rate of increase of the inductor current is also limited by the inductance itself. When the load current decreases rapidly, the output capacitor needs to store the excessive energy (stored in the inductor) until the regulator has decreased the inductor current by reducing the duty cycle. The recommendation is to use higher capacitance values, as the previous calculations show. Stabilizing the Control Loop Feedback Divider To speed up the control loop, a feedforward capacitor of 10 pF is recommended in the feedback divider, parallel to R3. To avoid coupling noise into the control loop from the feedforward capacitor, the feedforward effect can be bandwidth-limited by adding series resistor R4. A value in the range of 100 kΩ is suitable. The higher the resistance, the lower the noise coupled into the control loop system. 15 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 Compensation Capacitor The control loop of the converter is completely compensated internally. However the internal feedforward system requires an external capacitor. A 4.7-nF capacitor at the COMP pin of the converter is recommended. Layout Considerations For all switching power supplies the layout is an important step in the design, especially at high peak currents and high switching frequencies. If the layout is not carefully done, the regulator could show stability problems as well as EMI problems. Therefore, use wide and short traces for the main current paths, and for the power-ground tracks. The input capacitors, output capacitors, the inductors, and the rectifying diodes should be placed as close as possible to the IC to keep parasitic inductances low. The feedback divider should be placed as close as possible to the VREF pin of the IC. Use short traces when laying out the control ground. Figure 18 is an example layout circuit. Figure 16. Layout Considerations, Top View 16 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 GND V OUT Sense Signal Figure 17. Layout Considerations, Bottom View TPS63700 10Ω C1 0.1 mF VIN VREF EN FB OUT PS_GND VIN 2.7 V to 5.5 V IN C4 10 mF C2 0.22 mF D1 R3 1.2 MW C3 10 pF R4 100 kW VOUT, –12 V SW GND R2 121 kW SL03 COMP PowerPAD C6 4.7nF L1 10 mH C5 4 x 4.7 mF Figure 18. Layout Circuit 17 TPS63700 www.ti.com SLVS530 – SEPTEMBER 2005 THERMAL INFORMATION Implementation of integrated circuits in low-profile and fine-pitch surface-mount packages typically requires special attention to power dissipation. Many system-dependent issues, such as thermal coupling, airflow, added heatsinks and convection surfaces, and the presence of heat-generating components affect the power-dissipation limits of a given component. Three basic approaches for enhancing thermal performance are: • Improving the power dissipation capability of the PCB design • Improving the thermal coupling of the component to the PCB • Introducing airflow to the system The maximum recommended junction temperature (TJ) of the TPS63700 device is 125°C. The thermal resistance of the 10-pin SON, 3x3-mm package (DRC) is RJA = 48.7°C/W. Specified regulator operation is ensured to a maximum ambient temperature TA of 85°C. Therefore, the maximum power dissipation is about 821 mW. More power can be dissipated if the maximum ambient temperature of the application is lower. T T A P JMAX DMAX R JA (8) 18 PACKAGE OPTION ADDENDUM www.ti.com 17-Nov-2005 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty TPS63700DRCR ACTIVE SON DRC 10 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS63700DRCRG4 ACTIVE SON DRC 10 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS63700DRCT ACTIVE SON DRC 10 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR TPS63700DRCTG4 ACTIVE SON DRC 10 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. 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