LM2641
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SNVS040B – JANUARY 2000 – REVISED APRIL 2013
LM2641 Dual Adjustable Step-Down Switching Power Supply Controller
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FEATURES
DESCRIPTION
•
•
The LM2641 is a dual step-down power supply
controller intended for application in notebook
personal computers and other battery-powered
equipment.
1
2
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•
•
•
•
•
•
•
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300 kHz Fixed-Frequency Switching
Switching Synchronization with an External
Signal up to 400 kHz
Optional Pulse-Skipping Mode
Adjustable Secondary Feedback
Input Undervoltage Lockout
Output Undervoltage Shutdown Protection
Output Overvoltage Shutdown Protection
Programmable Soft-Start (Each Controller)
5V, 50 mA Linear Regulator Output
Precision 2.5V Reference Output
28-pin TSSOP
APPLICATIONS
•
•
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Notebook and Subnotebook Computers
Wireless Data Terminals
Battery-Powered Instruments
KEY SPECIFICATIONS
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96 % Efficient
5.5 – 30 V Input Range
Dual Outputs Adjustable from 2.2 – 8 V
0.5 % Typical Load Regulation Error
0.002 %/V Typical Line Regulation Error
Fixed-frequency synchronous drive of logic-level Nchannel power MOSFETs is combined with an
optional pulse-skipping mode to achieve ultra efficient
power conversion over a 1000:1 load current range.
The pulse-skipping mode can be disabled in favor of
fixed-frequency operation regardless of the load
current level.
High DC gain and current-mode feedback control
assure excellent line and load regulation and a wide
loop bandwidth for fast response to dynamic loads.
An internal oscillator fixes the switching frequency at
300 kHz. Optionally, switching can be synchronized
to an external clock running as fast as 400 kHz.
An optional soft-start feature limits current surges
from the input power supply at start up and provides
a simple means of start-up sequencing.
Logic-level inputs allow the controllers to be turned
ON and OFF separately.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM2641
SNVS040B – JANUARY 2000 – REVISED APRIL 2013
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Connection Diagram
Figure 1. 28-Lead TSSOP (PW)
Top View
See Package Number PW0028A
Pin Descriptions (1)
Pin #
(1)
2
Name
Function
1
CSH2
The sense point for the positive side of the voltage across the current sense resistor (R13) placed in
series with output #2.
2
FB2
The regulated output voltage appearing at output #2 is sensed using this pin by connecting it to the
center of the output resistive divider (R15 and R16).
3
COMP2
An R-C network made up of R11, C10, and C12 is connected to this pin which provides loop
compensation for regulated output #2.
4
SS2
This provides programmable soft-start for the #2 output along with capacitor C15.
5
ON/OFF2
This pin turns off only output #2.
6
SD
The part can be put into “sleep” mode using this pin, where both outputs are off and the internal chip
functions are shut down.
7
SYNC
The internal oscillator may be synchronized to an external clock via this pin.
8
GND
Connect this pin to circuit Signal Ground.
9
REF
Internal 2.5V reference voltage. This voltage is turned off by the SD pin, but remains on if either or both
ON/OFF pins are pulled low, which turns off the regulated output(s).
10
2NDFB/FPWM
A 12V supply can be generated using an auxiliary winding on the 5V output inductor. Feedback to
control this 12V output is brought in through this pin. If the 12V supply is not required, this pin can also
force the chip to operate at fixed frequency at light loads by pulling the pin low (this is the “forced-PWM”
mode of operation). This will prevent the converter from operating in pulse-skipping mode.
11
ON/OFF1
This pin turns off only output #1.
12
SS1
This provides programmable soft-start for the #1 output along with capacitor C3.
13
COMP1
An R-C network made up of R6, C5, and C7 is connected to this pin which provides loop compensation
for regulated output #1.
14
FB1
The regulated output voltage appearing at output #1 is sensed using this pin by connecting it to the
center of the output resistive divider (R1 and R2).
15
CSH1
The sense point for the positive side of the voltage across the current sense resistor (R4) placed in
series with output #1.
16
HDRV1
The drive for the gate of the high-side switching FET used for output #1.
17
SW1
This is the switching output drive point of the two power FETs which produce output #1.
18
CBOOT1
The bootstrap capacitor (C8) for output #1 is returned to this point.
19
LDRV1
The drive for the gate of the low-side switching FET (synchronous rectifier) used for output #1.
20
PGND
Connect this pin to circuit Power Ground.
(Refer to Typical Application Circuits)
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Pin Descriptions(1) (continued)
Pin #
Name
Function
21
CSL1
The sense point for the negative side of the voltage across the current sense resistor (R4) placed in
series with output #1.
22
LIN
This pin provides a low-current (50 mA max) 5V output. This output is always on, and can not be turned
off by either the SD or ON/OFF pins.
23
IN
This is the connection for the main input power.
24
LDRV2
The drive for the gate of the low-side switching FET (synchronous rectifier) used for output #2.
25
CBOOT2
The bootstrap capacitor (C9) for output #2 is returned to this point.
26
SW2
This is the switching output drive point of the two power FETs which produce output #2.
27
HDRV2
The drive for the gate of the high-side switching FET used for output #2.
28
CSL2
The sense point for the negative side of the voltage across the current sense resistor (R13) placed in
series with output #2.
Typical Application Circuits
Figure 2. Application With 5V/3A and 3.3V/4A Outputs
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Figure 3. Application With 5V/3A, 3.3V/4A, and 12V/0.3A Outputs
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
4
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Absolute Maximum Ratings (1) (2) (3)
−0.3 to 31V
IN, SW1, and SW2
−0.3 to 3V
FB1 and FB2
−0.3 to (VLIN +0.3)V
SD, ON/OFF1, ON/OFF2,
2NDFB/FPWM, SYNC, REF,
SS1, SS2, COMP1,
COMP2 and CSL1
−0.3 to 6V
LIN
CSH1, CSH2, and CSL2
(4)
−0.3 to 9V
−0.3 to 5V
Voltage from CBOOT1 to SW1
and from CBOOT2 to SW2
Voltage from HDRV1 to SW1
and from HDRV2 to SW2
−0.3V
Voltage from CBOOT1 to HDRV1
and from CBOOT2 to HDRV2
−0.3V
Junction Temp.
Power Dissipation
+150°C
(5)
883 mW
−65 to +150°C
Ambient Storage Temp. (TJ)
Soldering Dwell Time, Temp.
ESD Rating
(1)
(2)
(3)
(4)
(5)
(6)
(7)
(6)
Wave
4 sec, 260°C
Infrared
10 sec, 240°C
Vapor Phase
75 sec, 219°C
(7)
2 kV
Unless otherwise specified, all voltages are with respect to the voltage at the GND and PGND pins.
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is specified. Operating Ratings do not imply specified performance limits. For specified performance limits and
associated test conditions, see the Electrical Characteristics tables.
If Military/Aerospace specified devices are required, please contact the TI Sales Office/ Distributors for availability and specifications.
In applications where the output voltage can exceed the absolute maximum rating, a 100Ω resistor must be placed in series with the
CSH and CSL inputs.
The Absolute Maximum power dissipation depends on the ambient temperature. The 883 mW rating results from substituting 150°C,
70°C, and 90.6°C/W for TJmax, TA, and θJA respectively into the formula Pmax = (TJmax - TA)/θJA, where Pmax is the Absolute Maximum
power dissipation, TJmax is the Absolute Maximum junction temperature, TA is the ambient temperature, and θJA is the junction-toambient thermal resistance of the package. A θJA of 90.6°C/W represents the worst-case condition of no heat sinking of the 28-pin
TSSOP. Heat sinking allows the safe dissipation of more power. The Absolute Maximum power dissipation must be derated by 11.04
mW per °C above 70°C ambient. The LM2641 actively limits its junction temperature to about 150°C.
For detailed information on soldering plastic small-outline packages, refer to (SNOA549) available from TI.
For testing purposes, ESD was applied using the human-body model, a 100 pF capacitor discharged through a 1.5 kΩ resistor.
Operating Ratings
(1) (2)
VIN
5.5 to 30V
Junction Temp. (TJ)
(1)
(2)
0 to +125°C
Unless otherwise specified, all voltages are with respect to the voltage at the GND and PGND pins.
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is specified. Operating Ratings do not imply specified performance limits. For specified performance limits and
associated test conditions, see the Electrical Characteristics tables.
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Electrical Characteristics
Typicals and limits appearing in regular type apply for TJ = 25°C. Limits appearing in boldface type apply over the entire
junction temperature range for operation, 0 to +125°C. Unless otherwise specified under the Parameter or Conditions
columns, VIN = 10V, and VSD = VON/OFF1 = VON/OFF2 = 5V. (1) (2) (3)
Symbol
Parameter
Conditions
Typical
Limit
Units
System
VIN
Input Supply Voltage Range
5.5
V(min)
30
V(max)
VOUT1
Output Voltage Adjustment
Range
2.2
V(min)
6.0
V(max)
Output Voltage Adjustment
Range
2.2
V(min)
8.0
V(max)
VOUT2
ΔVOUT/VOUT
Load Regulation
0 mV ≤ (CSH1-CSL1) ≤ 80 mV,
0 mV ≤ (CSH2-CSL2) ≤ 80 mV
ΔVOUT/ΔVIN
Line Regulation
5.5V ≤ VIN ≤ 30V
IIN
Input Supply Current
ON
(4)
0.5
%
0.002
%/V
0.6
VFB1 = VFB2 = 1.4V,
mA
1
mA(max)
150
µA(max)
60
µA(max)
VCSH1 = 5.2V, VCSL1 = 5V,
VCSH2 = 3.5V, VCSL2 = 3.3V
Standing By
(5)
80
VON/OFF1 = VON/OFF2 = 0V
Shut Down
(6)
25
VSD = 0V
ISS1, ISS2
VPCL
VNCL
Soft-Start Source Current
VSS1 = VSS2 = 1V
10
Positive Current Limit Voltage
(Voltage from CSH1 to CSL1
and from CSH2 to CSL2)
100
V2NDFB/FPWM = 0.8V
VOUT Undervoltage Shutdown
Latch Threshold
(2)
(3)
(4)
(5)
(6)
6
µA
2.0
µA(min)
7.0
µA(max)
µA
mV
80
mV(min)
140
mV(max)
−100
mV
−80
mV(min)
−140
mV(max)
70
VOUT Overvoltage Shutdown
Latch Threshold
(1)
µA
4.75
Soft-Start Sink Current
Negative Current Limit Voltage
(Voltage from CSH1 to CSL1
and from CSH2 to CSL2)
µA
%
60
%(min)
80
%(max)
150
%
135
%(min)
165
%(max)
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is specified. Operating Ratings do not imply specified performance limits. For specified performance limits and
associated test conditions, see the Electrical Characteristics tables.
A typical is the center of characterization data taken with TA = TJ = 25°C. Typicals are not specified.
All limits are specified. All electrical characteristics having room-temperature limits are tested during production with TA = 25°C. All hot
and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical
process control.
Both controllers are ON but not switching. Currents entering the IC at IN, CSL1, CSH1, CSL2, and CSH2 are measured. Those entering
at CSL1 and CSH1 are multiplied by 0.50 to emulate the effect of a switching conversion from 10V down to 5V. Those entering at CSL2
and CSH2 are multiplied by 0.33 to emulate the effect of a switching conversion from 10V down to 3.3V. After multiplication, all five
currents are added. Because the voltage at the CSL1 input is greater than the LIN-to-VOUT switchover threshold, most of the input
supply current enters the IC via the CSL1 input.
Both switching controllers are OFF. The 5V, 50 mA linear regulator (output at LIN) and the precision 2.5V reference (output at REF)
remain ON.
Both switching controllers and the 2.5V precision reference are OFF. The 5V, 50 mA linear regulator remains ON.
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Electrical Characteristics (continued)
Typicals and limits appearing in regular type apply for TJ = 25°C. Limits appearing in boldface type apply over the entire
junction temperature range for operation, 0 to +125°C. Unless otherwise specified under the Parameter or Conditions
columns, VIN = 10V, and VSD = VON/OFF1 = VON/OFF2 = 5V. (1)(2)(3)
Symbol
Parameter
Conditions
Secondary Feedback Threshold
Voltage (2NDFB/FPWM)
(2NDFB/FPWM) Pin
Pull-Up Current
Typical
Limit
Units
2.4
V(min)
2.6
V(max)
80
µA(max)
2.5
VSFB = 2.4V
VON/OFF1 = 0V
VON/OFF2 = 5V
(2NDFB/FPWM) Pin
Input Leakage Current
40
V
±0.1
µA
Gate Drive
VBOOT
Bootstrap Voltage (Voltage from
CBOOT1 to SW1 and from
CBOOT2 to SW2)
CBOOT1 and CBOOT2 Source 1µA
Each
4.5
V
4.3
V(min)
HDRV1 and HDRV2 Sink and
Source Current
0.35
A
LDRV1 and LDRV2 Sink and
Source Current
0.35
A
HDRV1 and HDRV2 High-Side
On-Resistance
VCBOOT1 = VCBOOT2 = 5V, VSW1 =
VSW2 = 0V
6
Ω
HDRV1 and HDRV2 Low-Side
On-Resistance
VCBOOT1 = VCBOOT2 = 5V, VSW1 =
VSW2 = 0V
4
Ω
LDRV1 and LDRV2 High-Side
On-Resistance
VLIN = 5V
8
Ω
LDRV1 and LDRV2 Low-Side
On-Resistance
VLIN = 5V
4
Ω
Oscillator
FOSC
Oscillator Frequency
Minimum OFF-Time
300
VFB1 =1V, Measured at HDRV1
SYNC Pulses are Low-Going
Feedback Input Bias Current
VFB1 = VFB2 = 1.4V
kHz(min)
345
kHz(max)
350
ns(max)
400
kHz(min)
200
ns(min)
250
nA(max)
250
Maximum Frequency of
Synchronization
Minimum Width of
Synchronization Pulses
kHz
255
ns
Error Amplifier
IFB1, IFB2
100
ICOMP1, ICOMP2
COMP Output Source Current
VFB1 = VFB2 = 1V, VCOMP1 = VCOMP2
= 1V
90
ICOMP1, ICOMP2
COMP Output Sink Current
VFB1 = VFB2 = 1.4V, VCOMP1 =
VCOMP2 = 0.2V
60
nA
µA
40
µA(min)
40
µA(min)
µA
Voltage References and Linear Voltage Regulator
VBG
Bandgap Voltage
VREF
Reference Voltage
VLIN
Output Voltage of the Linear
Voltage Regulator
0.01 mA ≤ IREF ≤ 5 mA Source, VLIN
≤ 6V
6V ≤ VIN ≤ 30V,
0 mA ≤ ILIN ≤ 25 mA
1.238
V
2.5
V
2.45
V(min)
2.55
V(max)
4.6
V(min)
5.4
V(max)
5
V
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Electrical Characteristics (continued)
Typicals and limits appearing in regular type apply for TJ = 25°C. Limits appearing in boldface type apply over the entire
junction temperature range for operation, 0 to +125°C. Unless otherwise specified under the Parameter or Conditions
columns, VIN = 10V, and VSD = VON/OFF1 = VON/OFF2 = 5V. (1)(2)(3)
Symbol
VUVLO
Parameter
Undervoltage Lockout Threshold
LIN-to-VOUT Switch-Over
Threshold
Conditions
See
(7)
VOUT taken at CSL1
Typical
Limit
Units
3.6
V(min)
4.4
V(max)
4.0
V
4.8
V
Logic Inputs
VIH
VIL
Minimum High Level Input
Voltage (SD, ON/OFF1,
ON/OFF2, and SYNC)
2.4
V(min)
Minimum High Level Input
Voltage (2NDFB/FPWM)
2.6
V(min)
Maximum Low Level Input
Voltage (SD, ON/OFF1,
ON/OFF2, SYNC, and
2NDFB/FPWM)
0.8
V(max)
Maximum Input Leakage Current Logic Input Voltage 0 or 5V
(SD, ON/OFF1, ON/OFF2, and
SYNC)
(7)
8
±0.1
µA
The controllers remain OFF until the voltage of the 5V, 50 mA linear regulator (output at LIN) reaches this threshold.
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Block Diagram
Figure 4. LM2641 Block Diagram
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Typical Performance Charateristics
10
Efficiency
vs
Load Current
Efficiency
vs
Load Current
Figure 5.
Figure 6.
Efficiency
vs
Load Current
Efficiency
vs
Load Current
Figure 7.
Figure 8.
Efficiency
vs
Load Current
Efficiency
vs
Load Current
Figure 9.
Figure 10.
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Typical Performance Charateristics (continued)
Efficiency
vs
Load Current
Efficiency
vs
Load Current
Figure 11.
Figure 12.
Efficiency
vs
Load Current
Ref Output Voltage
Figure 13.
Figure 14.
Ref Output Voltage
Normalized Switching Output Voltage vs
Junction Temperature
Figure 15.
Figure 16.
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Typical Performance Charateristics (continued)
12
Normalized Error Amplifier Voltage vs
Junction Temperature
Error Amplifier Gm vs
Junction Temperature
Figure 17.
Figure 18.
Normalized Oscillator Frequency vs
Junction Temperature
Shutdown Quiescent Current
And Standby Quiescent Current vs
Supply Voltage
Figure 19.
Figure 20.
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Theory of Operation
Basic Operation of the Current-Mode Controller
The output voltage is held at a constant value by the main control loop, which is made up of the error amplifier,
the current sense amplifier, and the PWM comparator (refer to the Block Diagram, Figure 4).
The LM2641 controller has two primary modes of operation: Forced Pulse-Width Modulation (FPWM) where the
controller always operates at a fixed frequency, and Pulse-Skipping mode where the controller frequency
decreases at reduced output loads to improve light-load efficiency.
FPWM Mode of Operation
Pulling the FPWM pin low initiates a mode of operation called Forced Pulse-Width Modulation (FPWM). This
means that the LM2641 will always operate at a fixed frequency, regardless of output load. The cycle of
operation is:
The high-side FET switch turns ON at the beginning of every clock cycle, causing current to flow through the
inductor. The inductor current ramps up, causing a voltage drop across the sense resistor, and this voltage is
amplified by the current sense amplifier.
The voltage signal from the current sense amplifier is applied to the input of the PWM comparator, where it is
compared to the control level set by the error amplifier. Once the current sense signal reaches this control
voltage, the PWM comparator resets the driver logic which turns OFF the high-side FET switch.
The low-side FET switch turns on after a delay time which is the lesser of either:
(a) The time it takes the SW pin voltage to reach zero (this voltage is sensed by the shoot-through protection
circuitry).
(b) 100 ns, which is the pre-set value for maximum delay.
When operating at very light loads (in FPWM mode), the inductor current must flow in a negative direction
through the low-side FET switch in order to maintain the fixed-frequency mode of operation. For this reason, the
built-in zero cross detector is disabled when ever FPWM mode is activated (that is, when ever the FPWM pin is
pulled to a low state).
It should be noted that if the FPWM pin is high (operation described in the FPWM Mode of Operation section),
the zero cross detector will turn OFF the low-side FET switch anytime the inductor current drops to zero (which
prevents negative inductor current).
Pulse-Skipping Mode of Operation
Pulling the FPWM pin high allows the LM2641 to operate in pulse-skipping mode at light loads, where the
switching frequency decreases as the output load is reduced. The controller will operate in fixed-frequency mode,
as described in the Pulse-Skipping Mode of Operation section, if the output load current is sufficiently high.
Pulse-skipping results in higher efficiency at light loads, as decreasing the switching frequency reduces switching
losses. The load current value where the transition from fixed-frequency to pulse-skipping operation occurs is the
point where the inductor current goes low enough to cause the voltage measured across the current sense
resistor (R4 or R13) to drop below 25 mV.
In pulse-skipping mode, the high-side FET switch will turn ON at the beginning of the first clock cycle which
occurs after the voltage at the feedback pin falls below the reference voltage. The high-side FET switch remains
ON until the voltage across the current sense resistor rises to 25 mV (and then it turns OFF).
Ramp Compensation
All current-mode controllers require the use of ramp compensation to prevent subharmonic oscillations, and this
compensation is built into the LM2641. The internal compensation assumes an RSENSE value of 25 mΩ, inductor
value of 6.8µH, and a maximum output voltage of 6V.
To prevent oscillations, the slope M of the compensation ramp must be equal to the maximum downward slope
of the voltage waveform at the output of the current sense amplifier. The relationship of the slope M to the
external components is given by:
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MCOMP = MCS AMP (max) = N X RSENSE X VOUT (max) / L
Where:
MCOMP is the slope of the compensation ramp.
MCS AMP (max) is the maximum downward slope of the voltage at the output of the current sense amplifier.
N is the gain of the current sense amplifier.
RSENSE is the value of the current sense resistor.
VOUT (max) is the maximum output voltage.
L is the inductance of the output inductor.
It is important to note that since the value RSENSE appears in the numerator and L is in the denominator, these
two values may be increased or decreased at the same ratio without changing the slope.
At higher values of load current, a lower value RSENSE will be selected. The inductance value for the output
inductor should be decreased by the same percentage to maintain correct ramp compensation.
APPLICATION INFORMATION
Improved Transient Response
If the output voltage falls below 97% of the nominal value, the low-voltage regulation (LREG) comparator will
activate logic which turns ON the high-side FET switch continuously until the output returns to nominal. The lowside FET switch is held OFF during this time.
This action will improve transient response since it bypasses the error amplifier and PWM comparator, forcing
the high-side switch ON until the output returns to nominal. This feature is disabled during start-up.
Boost High-Side Gate Drive
A “flying” bootstrap capacitor is used to generate the gate drive voltage used for the high-side FET switch. This
bootstrap capacitor is charged up to about 5V using an internal supply rail and diode when ever the low-side FET
switch is ON. When the high-side FET switch turns ON, the Source is pulled up near the input voltage. The
voltage across the bootstrap capacitor boosts up the gate drive voltage, ensuring that the Gate is driven at least
4.3V higher than the Source.
Reference
The internal bandgap reference is used to generate a 2.5V reference voltage which is connected to the REF pin.
The specified tolerance of the REF voltage is ±2% over the full operating temperature range, as long as the
current drawn is ≤ 5 mA.
A bypass capacitor on the REF pin is not required, but may be used to reduce noise.
5V LIN Output
The LM2641 contains a built-in 5V/50 mA LDO regulator whose output is connected to the LIN pin. Since this is
an LDO regulator, it does require an external capacitor to maintain stability. The minimum amount of capacitance
required for stability is 4.7 μF, with ESR in the range of about 100 mΩ to 3Ω. A good quality solid Tantalum
capacitor is recommended (ceramics can not be used because the ESR is too low). If cold temperature operation
is required, a capacitor must be selected which has an ESR that is in the stable range over the entire operating
temperature range of the application.
Since the current limit for this LDO regulator is set at about 85 mA, it can be used at load currents up to about 50
mA (assuming total IC power dissipation does not exceed the maximum value).
Ensured specifications are provided for worst-case values of VLIN over the full operating temperature range for
load currents up to 25mA (see Electrical Characteristics). To estimate how the VLIN output voltage changes when
going from ILIN = 25mA to ILIN = 50mA, a change in VLIN of about −30mV should be expected due to loading
(typical value only, not specified). This decrease in VLIN is linear with increasing load current.
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It must be understood that the maximum allowable current of 50mA must include the current drawn by the gate
drive circuitry. This means that the maximum current available for use at the LIN pin is 50 mA minus whatever is
being used internally for gate drive.
The amount of current used for gate drive by each switching output can be calculated using the formula:
IGD = 2 X Q X FOSC
(1)
Where:
IGD is the gate drive current supplied by VLIN.
Q is the gate charge required by the selected FET (see FET data sheet: Gate Charge Characteristics).
FOSC is the switching frequency.
Example: As shown in the typical application, if the FET NDS8410 is used with the LM2641, the turn-on gate
voltage (VGS) is 5V − VDIODE = 4.3V. Referring to the NDS8410 data sheet, the curve Gate Charge
Characteristics shows that the gate charge for this value of VGS is about 24 nC.
Assuming 300 kHz switching frequency, the gate drive current used by each switching output is:
IGD = 2 X Q X FOSC = 2 X (24 X 10−9) X (3 X 105) = 14.4 mA
(2)
If both outputs are switching, the total gate drive current drawn would be twice (28.8 mA).
Note that in cases where the voltage at switching output #1 is 4.8V or higher, the internal gate drive current is
obtained from that output (which means the full 50 mA is available for external use at the LIN pin).
SYNC Pin
The basic operating frequency of 300kHz can be increased to up to 400kHz by using the SYNC pin and an
external CMOS or TTL clock. The synchronizing pulses must have a minimum pulse width of 200 ns.
If the sync function is not used, the SYNC pin must be connected to the LIN pin or to ground to prevent false
triggering.
Current Limit Circuitry
The LM2641 is protected from damage due to excessive output current by an internal current limit comparator,
which monitors output current on a cycle-by-cycle basis. The current limiter activates when ever the absolute
magnitude of the voltage developed across the output sense resistor exceeds 100 mV (positive or negative
value).
If the sensed voltage exceeds 100 mV, the high-side FET switch is turned OFF. If the sensed voltage goes below
-100 mV, the low-side FET switch is turned OFF. It should be noted that drawing sufficient output current to
activate the current limit circuits can cause the output voltage to drop, which could result in a under-voltage latchOFF condition (see Under-voltage/Over-voltage Protection section).
Under-voltage/Over-voltage Protection
The LM2641 contains protection circuitry which activates if the output voltage is too low (UV) or too high (OV). In
the event of either a UV or OV fault, the LM2641 is latched off and the high-side FET is turned off, while the lowside FET is turned on.
If the output voltage drops below 70% of nominal value, the under-voltage comparator will latch OFF the
LM2641. To restore operation, power to the device must be shut off and then restored.
It should be noted that the UV latch provides protection in cases where excessive output current forces the
output voltage down. The UV latch circuitry is disabled during start-up.
If the output voltage exceeds 150% of nominal, the over-voltage comparator latches off the LM2641. As stated
before, power must be cycled OFF and then ON to restore operation.
It must be noted that the OV latch can not protect the load from damage in the event of a high-side FET switch
failure (where the FET shorts out and connects the input voltage to the load).
Protection for the load in the event of such a failure can be implemented using a fuse in the power lead. Since
the low-side FET switch turns ON whenever the OV latch activates, this would blow a series fuse if the FET and
fuse are correctly sized.
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Soft-Start
An internal 5 µA current source connected to the soft-start pins allows the user to program the turn-on time of the
LM2641. If a capacitor is connected to the SS pin, the voltage at that pin will ramp up linearly at turn ON. This
voltage is used to control the pulse widths of the FET switches.
The pulse widths start at a very narrow value and linearly increase up to the point where the SS pin voltage is
about 1.3V. At that time, the pulse-to-pulse current limiter controls the pulse widths until the output reaches its
nominal value (and the PWM current-mode control loop takes over).
The LM2641 contains a digital counter (referenced to the oscillator frequency) that times the soft-start interval.
The maximum allotted SS time period is 4096 counts of the oscillator clock, which means the time period varies
with oscillator frequency:
max. allowable SS interval = 4096 / FOSC
(3)
If the output voltage does not move to within −1% of nominal in the period of 4096 counts, the device will latch
OFF. To restore operation, the power must be cycled OFF to ON.
Minimum Pulse Width
As the input voltage is increased, the pulse widths of the switching FET's decreases. If the pulse widths become
narrower than 350ns, pulse jitter may occur as the pulses alternate with slightly different pulse widths. This is
does not affect regulator stability or output voltage accuracy.
Start-Up Issues
The LM2641 contains an output undervoltage protection circuit which is made up of a digital counter and a
comparator which monitors VOUT. During turn-on, the counter begins counting clock cycles when the input
voltage reaches approximately 3V. If the counter reaches 4096 cycles before the output voltage rises to within
1% of nominal value, the IC will be latched off in an undervoltage fault condition.
The function of this protection is to shut the regulator off if the output is overloaded (such as a short to ground).
However, the UV latch can cause start-up problems if the circuit is not properly designed. The following Input
Voltage Rise Time and Input Capacitance sections explain how to avoid these types of problems:
Input Voltage Rise Time
If the input voltage rises too slowly, the LM2641 will latch off in an undervoltage condition. To avoid this problem,
the input voltage must rise quickly enough to allow the output to get into regulation before the 4096 count time
interval elapses. For a switching frequency of 300 kHz, 4096 cycles will be completed in 13.6 milliseconds.
In reality, the total rise time of VIN should not approach the 4096 clock cycle limit if reliable start-up is to be
assured. It should be noted that the total rise time of VIN is also affected by current loading when the power
converter begins switching (which draws power from the input capacitors) causing their voltage to sag (details of
input capacitor requirements are outlined in the Input Capacitance section).
It is also important to note that this type of start-up problem is more likely to occur at higher values of output
voltage, since the input voltage must rise to a higher voltage to allow the output voltage to regulate (which means
the input dV/dt rate has to be faster). The recommended output voltage limit of 6V should not be exceeded.
Input Capacitance
The amount and type of input capacitance present is directly related to how well the regulator can start up. The
reason is that the input capacitors serve as the source of energy for the power converter when the regulator
begins switching. Typically, the input voltage (which is the voltage across the input capacitors) will sag as the
power converter starts drawing current which will cause a dip in VIN as it is ramping up. If the input capacitors are
too small or have excessive ESR, the input volatge may not be able to come up fast enough to allow the output
voltage to get into regulation before the digital clock counts off 4096 cycles and the part will latch off as an
undervoltage fault.
To prevent this type of start-up problem:
1. The input capacitors must provide sufficient bulk capacitance and have low impedance. Solid Tantalum
capacitors designed for high-frequency switching applications are recommended as they generally provide
the best cost/performance characteristics and maintain a very low ESR even at cold temperatures. Ceramic
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capacitors also have very low ESR over the full temperature range, but X5R/X7R dielectric types should be
used to assure sufficient capacitance will be provided (Z5U or Y5F types are not suitable).
– Some of the newer electrolytic types such as POSCAP, OSCON, and polymer electrolytic may also be
usable as input capacitors. However, care must be taken if the application will be used at low
temperatures as the ESR of these capacitors may increase significantly at temperatures below 0°C. Most
aluminum electrolytes are not usable with this IC at temperatures below this limit. Check the ESR
specifications of the selected capacitor carefully if low temperature operation will be required.
2. The input capacitors must be physically located not more than one centimeter away from the switching
FET's, as trace inductance in the switching current path can cause problems.
Loop Compensation
The LM2641 must be properly compensated to assure stable operation and good transient response. As with any
control loop, best performance is achieved when the compensation is optimized so that maximum bandwidth is
obtained while still maintaining sufficient phase margin for good stability.
Best performance for the LM2641 is typically obtained when the loop bandwidth (defined as the frequency where
the loop gain equals unity) is in the range of FOSC/10 to FOSC/5.
In the discussion of loop stability, it should be noted that there is a high-frequency pole fp(HF), whose frequency
can be approximated by:
fp(HF) ∼ FOSC/2 X QS (Assumes QS < 0.5)
Where:
(4)
As can be seen in the approximation for QS, the highest frequency for fp(HF) occurs at the maximum value of
VIN. The lowest frequency for fp(HF) is about FOSC/10 (when VIN = 4.5V and VOUT = 1.8V).
As noted above, the location of the pole fp(HF) is typically in the range of about FOSC/10 to FOSC/4. This pole will
often be near the unity-gain crossover frequency, and it can significantly reduce phase margin if left
uncompensated. Fortunately, the ESR of the output capacitor(s) forms a zero which is usually very near the
frequency of fp(HF), and provides cancellation of the negative phase shift it would otherwise cause. For this
reason, the output capacitor must be carefully selected.
Most of the loop compensation for the LM2641 is set by an R-C network from the output of the error amplifier to
ground (see Figure 21). Since this is a transconductance amplifier, it has a very high output impedance (160 kΩ).
Figure 21. Typical Compensation Network
The components shown will add poles and zeros to the loop gain as given by the following equations:
C10 adds a pole whose frequency is given by:
fp(C10) = 1 / [2π X C10 (R11 + 160k) ]
C12 adds a pole whose frequency is given by:
fp(C12) = 1 / [2π X C12 (R11 || 160k) ]
R11 adds a zero whose frequency is given by:
fz(R11) = 1 / [2π X R11 (C10 + C12) ]
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The output capacitor adds both a pole and a zero to the loop:
fp(COUT) = 1 / [2π X RL X COUT]
fz(ESR) = 1 / [2π X ESR X COUT]
Where RL is the load resistance, and ESR is the equivalent series resistance of the output capacitor(s).
The function of the compensation components will be explained in a qualitative discussion of a typical loop gain
plot for an LM2641 application, as illustrated in Figure 22.
Figure 22. Typical Loop Gain Plot
C10 and R11 form a pole and a zero. Changing the value of C10 moves the frequency of both the pole and the
zero. Changing R11 moves the zero without significantly affecting the pole.
The C10 pole is typically referred to as the dominant pole, and its primary function is to roll off loop gain and
reduce the bandwidth.
The R11 zero is required to add some positive phase shift to offset some of the negative phase shift from the two
low-frequency poles. Without this zero, these two poles would cause −180° of phase shift at the unity-gain
crossover, which is clearly unstable. Best results are typically obtained if R11 is selected such that the frequency
of fz(R11) is in the range of fc/4 to fc where fc is the unity-gain crossover frequency.
The output capacitor (along with the load resistance RL) forms a pole shown as fp(COUT). Although the frequency
of this pole varies with RL, the loop gain also varies proportionally which means the unity-gain crossover
frequency stays essentially constant regardless of RL value.
C12 can be used to create an additional pole most often used for bypassing high-frequency switching noise on
the COMP pin. In many applications, this capacitor is unnecessary.
If C12 is used, best results are obtained if the frequency of the pole is set in the range FOSC/2 to 2FOSC. This will
provide bypassing for the high-frequency noise caused by switching transitions, but add only a small amount of
negative phase shift at the unity-gain crossover frequency.
The ESR of COUT (as well as the capacitance of COUT) form the zero fz(ESR), which typically falls somewhere
between 10kHz and 50kHz. This zero is very important, as it cancels phase shift caused by the high-frequency
pole fp(HF). It is important to select COUT with the correct value of capacitance and ESR to place this zero near fc
(typical range fc/2 to fc).
As an example, we will present an analysis of the loop gain plot for a 3.3V design. Values used for calculations
are:
VIN = 12V
VOUT = 3.3V @ 4A
COUT = C14 + C16 = 200 µF
ESR = 60 mΩ(each) = 30mΩ total
FOSC = 300kHz
fp(HF) ∼ 40kHz
R13 = 20mΩ
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L2 = 6.8 µH
RL = 0.825Ω
DC gain = 55dB
The values of compensation components will be: C10 = 2200 pF, R11 = 8.2k, and C12 will not be used. Using
this data, the poles and zeros are calculated:
fp(C10) = 1 / [2π X C10 (R11 + 160k) ] = 430Hz
fz(R11) = 1 / [2π X R11 (C10 + C12) ] = 8.8kHz
fp(COUT) = 1 / [2π X RL X COUT] = 960Hz
fz(ESR) = 1 / [2π X ESR X COUT] = 27kHz
fp(HF) ∼ 40kHz
Using these values, the calculated gain plot is shown in Figure 23.
Figure 23. Calculated Gain Plot for 3.3V/4A Application
Looking at the plot, it can be seen that the unity-gain crossover frequency fc is expected to be about 25kHz.
Using this value, the phase margin at the point is calculated to be about 84°.
To verify the accuracy of these calculations, the circuit was bench tested using a network analyzer. The
measured gain and phase are shown plotted in Figure 24.
Figure 24. Measured Gain/Phase Plot for 3.3V/4A Application
The measured gain plot agrees very closely to the predicted values. The phase margin at 0dB is slightly less
than predicted (71° vs. 84°), which is to be expected due to the negative phase shift contributions of high
frequency poles not included in this simplified analysis.
It should be noted that 70° phase margin with 25kHz bandwidth is excellent, and represents the optimal
compensation for this set of values for VIN, VOUT, inductor and RL.
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Optimizing Stability
The best tool for measuring both bandwidth and phase margin is a network analyzer. If this is not available, a
simple method which gives a good measure of loop stability is to apply a minimum to maximum step of output
load current and observe the resulting output voltage transient. A design which has good phase margin (>50°)
will typically show no ringing after the output voltage transient returns to its nominal value.
It should be noted that the stability (phase margin) does not have to be optimal for the regulator to be stable. The
design analyzed in the Loop Compensation section was re-compensated by changing R11 and C10 to
intentionally reduce the phase margin to about 35° and re-tested for step response. The output waveform
displayed slight ringing after the initial return to nominal, but was completely stable otherwise.
In most cases, the compensation components shown in the Typical Application Circuits will give good
performance. To assist in optimizing phase margin, the following guidelines show the effects of changing various
components.
COUT: Increasing the capacitance of COUT moves the frequency of the pole fp(COUT) to a lower value and reduces
loop bandwidth. Increasing COUT can be beneficial (increasing the phase margin) if the loop bandwidth is too
wide (>FOSC/5) which places the high-frequency poles too close to the unity-gain crossover frequency.
ESR of COUT: The ESR forms a zero fz(ESR), which is needed to cancel negative phase shift near the unity-gain
frequency. High-ESR capacitors can not be used, since the zero will be too low in frequency which will make the
loop bandwidth too wide.
R11/C10: These form a pole and a zero. Changing the value of C10 changes the frequency of both the pole and
zero. Note that since this causes the frequency of both the pole and zero to move up or down together, adjusting
the value of C10 does not significantly affect loop bandwidth.
Changing the value of R11 moves the frequency location of the zero fz(R11), but does not significantly shift the
C10 pole (since the value of R11 is much less than the 160kΩ output impedance of the Gm amplifier). Since only
the zero is moved, this affects both bandwidth and phase margin. This means adjusting R11 is an easy way to
maximize the positive phase shift provided by the zero. Best results are typically obtained if fz(R11) is in the
frequency range of fc/4 to fc (where fc is the unity-gain crossover frequency).
Design Procedure
This section presents guidelines for selecting external components.
INDUCTOR SELECTION
In selecting an inductor, the parameters which are most important are inductance, current rating, and DC
resistance.
Inductance
It is important to understand that all inductors are not created equal, as the method of specifying inductance
varies widely.
It must also be noted that the inductance of every inductor decreases with current. The core material, size, and
construction type all contribute the the inductor's dependence on current loading. Some inductors exhibit
inductance curves which are relatively flat, while others may vary more than 2:1 from minimum to maximum
current. In the latter case, the manufacturer's specified inductance value is usually the maximum value, which
means the actual inductance in your application will be much less.
An inductor with a flatter inductance curve is preferable, since the loop characteristics of any switching converter
are affected somewhat by inductance value. An inductor which has a more constant inductance value will give
more consistent loop bandwidth when the load current is varied.
The data sheet for the inductor must be reviewed carefully to verify that the selected component will have the
desired inductance at the frequency and current for the application.
Current Rating
This specification may be the most confusing of all when picking an inductor, as manufacturers use different
methods for specifying an inductor's current rating.
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The current rating specified for an inductor is typically given in RMS current, although in some cases a peak
current rating will also be given (usually as a multiple of the RMS rating) which gives the user some indication of
how well the inductance operates in the saturation region.
Other things being equal, a higher peak current rating is preferred, as this allows the inductor to tolerate high
values of ripple current without significant loss of inductance.
In the some cases where the inductance vs. current curve is relatively flat, the given current rating is the point
where the inductance drops 10% below the nominal value. If the inductance varies a lot with current, the current
rating listed by the manufacturer may be the “center point” of the curve. This means if that value of current is
used in your application, the amount of inductance will be less than the specified value.
DC Resistance
The DC resistance of the wire used in an inductor dissipates power which reduces overall efficiency. Thicker wire
decreases resistance, but increases size, weight, and cost. A good tradeoff is achieved when the inductor's
copper wire losses are about 2% of the maximum output power.
Selecting An Inductor
Determining the amount of inductance required for an application can be done using the formula:
(5)
Where:
VIN is the maximum input voltage.
VOUT is the output voltage.
F is the switching frequency, FOSC
IRIPPLE is the inductor ripple current. In general, a good value for this is about 30% of the DC output current.
It can be seen from the above equation, that increasing the switching frequency reduces the amount of required
inductance proportionally. Of course, higher frequency operation is typically less efficient because switching
losses become more predominant as a percentage of total power losses.
It should also be noted that reducing the inductance will increase inductor ripple current (other terms held
constant). This is a good point to remember when selecting an inductor: increased ripple current increases the
FET conduction losses, inductor core losses, and requires a larger output capacitor to maintain a given amount
of output ripple voltage. This means that a cheaper inductor (with less inductance at the operating current of the
application) will cost money in other places.
INPUT CAPACITORS
The switching action of the high-side FET requires that high peak currents be available to the switch or large
voltage transients will appear on the VIN line. To supply these peak currents, a low ESR capacitor must be
connected between the drain of the high-side FET and ground. The capacitor must be located as close as
possible to the FET (maximum distance = 0.5 cm).
A solid Tantalum or low ESR aluminum electrolytic can be used for this capacitor. If a Tantalum is used, it must
be able to withstand the turn-ON surge current when the input power is applied. To assure this, the capacitor
must be surge tested by the manufacturer and specified to work in such applications.
Caution: If a typical off-the-shelf Tantalum is used that has not been surge tested, it can be blown during powerup and will then be a dead short. This can cause the capacitor to catch fire if the input source continues to supply
current.
Voltage Rating
For an aluminum electrolytic, the voltage rating must be at least 25% higher than the maximum input voltage for
the application.
Tantalum capacitors require more derating, so it is recommended that the selected capacitor be rated to work at
a voltage that is about twice the maximum input voltage.
Current Rating
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Capacitors are specified with an RMS current rating. To determine the requirement for an application, the
following formula can be used:
(6)
It is also recommended that a 0.1µF ceramic capacitor be placed from VIN to ground for high frequency
bypassing, located as close as possible to the VIN pin.
OUTPUT CAPACITORS
The output capacitor(s) are critical in loop stability (covered in INPUT CAPACITORS section) and also output
voltage ripple.
The types best suited for use as output capacitors are aluminum electrolytics and solid Tantalum.
Aluminum Electrolytics
The primary advantage of aluminum electrolytics is that they typically give the maximum capacitance-to-size
ratio, and they are reasonably priced. However, it must be noted that aluminum electrolytics used in highperformance switching regulator designs must be high frequency, low ESR types such as Sanyo OSCON or
Panasonic HFQ which are specifically designed for switching applications. Capacitors such as these with good
high frequency (≥ 100kHz) specifications are not cheap.
Aluminum electrolytic capacitors should generally not be used in switching regulator applications where the
ambient temperature goes below 0°C. A typical low-voltage aluminum electrolytic has an ESR vs. Temperature
curve that is fairly flat from 25°C to 125°C. However, a temperature change from 25°C to 0°C will approximately
double the ESR, and it will double again going from 0°C down to −20°C.
Tantalum
Solid Tantalum capacitors are best in applications which must operate over a wide temperature range. A good
quality Tantalum will typically exhibit less than 2:1 change in ESR over the temperature range of +125°C to
−40°C. Recommended types are Sprague 593D, Sprague 594D, and AVX TPS series.
Selecting An Output Capacitor
The required value of output capacitance is directly related to the specification for the maximum amount of output
voltage ripple allowed in the application. Since ESR effects the ripple voltage, it is important to have a guideline
for ESR. The maximum allowed ESR can be calculated as follows.
VRIPPLE = IRIPPLE *ESR(max)
Using V = Ldi/dt
VOUT = L *IRIPPLE/{(1−D)TS} = L *IRIPPLE *FS/(1−D)
IRIPPLE = VOUT*(1−D)/)L *FS)
ESR(max) = VRIPPLE/IRIPPLE
A reasonable value for COUT can be obtained by choosing capacitors with net ESR less than ½ of ESR(max).
Hence,
ESR(max) = VRIPPLE*L* FS/ {VOUT(1−D)}
(7)
The value of COUT necessary to meet the voltage ripple specification can be found using the approximation:
(8)
Where:
IRIPPLE is the inductor ripple current.
VRIPPLE is the output ripple voltage.
ESR is the equivalent series resistance of the output capacitor.
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F is the switching frequency, FS.
TS = 1/FS.
D = Duty Cycle.
The ESR term predominates in determining output ripple voltage. Good quality Tantalum capacitors have
specified maximum specifications for ESR, but the typical values for ESR are usually considerably lower than the
maximum limit.
POWER MOSFETs
Two N-channel logic-level MOSFETs are required for each output. The voltage rating should be at least 1.2 times
the maximum input voltage.
Maximizing efficiency for a design requires selecting the right FET. The ON-resistance of the FET determines the
ON-state (conduction) losses, while gate charge defines the losses during switch transitions. These two
parameters require a trade-off, since reducing ON-resistance typically requires increasing gate capacitance
(which increases the charge required to switch the FET). Improved FETs are currently being released which are
designed specifically for optimized ON-resistance and gate charge characteristics.
The VIN and VOUT for a specific application determines the ON time of each switch. In some cases where one
FET is on most of the time, efficiency may be improved slightly by selecting a low ON-resistance FET for one of
the FET switches and a different type with lower gate charge requirement for the other FET switch. However, for
most applications this would give no measurable improvement.
CURRENT SENSE RESISTOR
A sense resistor is placed between the inductor and the output capacitor to measure the inductor current. The
value of this resistor is set by the current limit voltage of the LM2641 (see Electrical Characteristics) and the
maximum (peak) inductor current. The value of the sense resistor can be calculated from:
(9)
Where:
VCL(MIN) is the minimum specified current limit voltage (see Electrical Characteristics).
IMAX is the maximum output current for the application.
IRIPPLE is the inductor ripple current for the application.
TOL is the tolerance (in %) of the sense resistor.
The physical placement of the sense resistors should be as close as possible to the LM2641 to minimize the lead
length of the connections to the CSH and CSL pins. Keeping short leads on these connections reduces the
amount of switching noise conducted into the current sense circuitry of the LM2641.
EXTERNAL DIODES
FET Diodes
Both of the low-side MOSFET switches have an external Schottky diode connected from drain to source. These
diodes are electrically in parallel with the intrinsic body diode present inside the FET. These diodes conduct
during the dead time when both FETs are off and the inductor current must be supplied by the catch diode
(which is either the body diode or the Schottky diode).
Converter efficiency is improved by using external Schottky diodes. Since they have much faster turn-off
recovery than the FET body diodes, switching losses are reduced.
The voltage rating of the Schottky must be at least 25% higher than the maximum input voltage. The average
current rating of the diode needs to be only about 30% of the output current, because the duty cycle is low.
The physical placement of the Schottky diode must be as close as possible to the FET, since any parasitic (lead)
inductance in series with the Schottky will slow its turn-ON and cause current to flow through the FET body
diode.
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Bootstrap Diodes
As shown in the block diagram for the LM2641, the CBOOT pin has an internal diode which is connected to the
5V internal rail (which is also connected to the LIN pin). This diode charges up the bootstrap capacitor to about
5V when the low-side FET switch turns ON and pulls its drain down to ground. The internal diode works well until
the pulse widths get extremely narrow, and then the charge applied to the bootstrap capacitor can become
insufficient to fully turn ON the gate of the FET.
For this reason, an external diode should be used which connects directly between the bootstrap capacitor and
the external capacitor connected to the LIN pin (C17). A fast-recovery silicon diode should be used which has an
average current rating ≥ 50 mA, with voltage rating > 30V.
Output Diodes
It is recommended that diodes be placed between the regulated outputs and ground to prevent the outputs from
swinging below ground. The diode used may be a Schottky or silicon type, and should have a current rating of
1A or more. If the outputs are allowed to swing below ground more than a Vbe, the substrate of the LM2641 will
become forward biased which will cause the part to operate incorrectly. Another potential problem which could be
caused by negative output transients is damage to the output capacitors, since tantalum capacitors can be
damaged if a reverse voltage is forced across them
The operating conditions where this can occur are not typical: it can happen if one or both of the outputs are very
lightly loaded, and an undervoltage (or overvoltage) condition is detected. When this happens, the LM2641 turns
off the switching oscillator and turns on both of the low-side FET's which abruptly grounds one end of the
inductor. When this happens, the other end of the inductor (which is connected to the regulated output) will
experience a transient ringing voltage as the energy stored in the inductor is discharged. The amplitude and
duration of the ringing is a function of the R-L-C tank circuit made up the output capacitance, inductor, and
resistance of the inductor windings.
Because of this, the choice of inductor influences how large in amplitude the ringing will be. In tests performed on
the Typical Application Circuits, the Sumida inductor showed less ringing than the Pulse inductor, but both
showed a voltage transient that would go slightly below ground. For this reason, the output diodes are
recommended.
24
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Copyright © 2000–2013, Texas Instruments Incorporated
Product Folder Links: LM2641
LM2641
www.ti.com
SNVS040B – JANUARY 2000 – REVISED APRIL 2013
REVISION HISTORY
Changes from Revision A (April 2013) to Revision B
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 24
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Copyright © 2000–2013, Texas Instruments Incorporated
Product Folder Links: LM2641
25
PACKAGE OPTION ADDENDUM
www.ti.com
18-Oct-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LM2641MTC-ADJ/NOPB
ACTIVE
TSSOP
PW
28
48
Green (RoHS
& no Sb/Br)
SN | CU SN
Level-3-260C-168 HR
0 to 125
LM2641M
TC-ADJ
LM2641MTCX-ADJ/NOPB
ACTIVE
TSSOP
PW
28
2500
Green (RoHS
& no Sb/Br)
CU SN | Call TI
Level-3-260C-168 HR
0 to 125
LM2641M
TC-ADJ
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
18-Oct-2013
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
23-Sep-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
LM2641MTCX-ADJ/NOPB TSSOP
PW
28
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
2500
330.0
16.4
Pack Materials-Page 1
6.8
B0
(mm)
K0
(mm)
P1
(mm)
10.2
1.6
8.0
W
Pin1
(mm) Quadrant
16.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
23-Sep-2013
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM2641MTCX-ADJ/NOPB
TSSOP
PW
28
2500
367.0
367.0
38.0
Pack Materials-Page 2
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