OP
A8
90
OPA890
SBOS369 – MAY 2007
Low-Power, Wideband, Voltage-Feedback
OPERATIONAL AMPLIFIER with Disable
FEATURES
•
•
•
•
•
•
•
DESCRIPTION
FLEXIBLE SUPPLY RANGE:
+3V to +12V Single Supply
±1.5V to ±6V Dual Supplies
UNITY-GAIN STABLE
WIDEBAND +5V OPERATION: 115MHz
(G = +2V/V)
OUTPUT VOLTAGE SWING: ±4V
HIGH SLEW RATE: 500V/µs
LOW QUIESCENT CURRENT: 1.1mA
LOW DISABLE CURRENT: 30µA
The OPA890 represents a major step forward in
unity-gain stable, voltage-feedback op amps. A new
internal architecture provides slew rate and
full-power bandwidth previously found only in
wideband, current-feedback op amps. These
capabilities provide exceptional full power bandwidth.
Using a single +5V supply, the OPA890 can deliver a
1V to 4V output swing with over 35mA drive current
and 220MHz bandwidth. This combination of
features makes the OPA890 an ideal RGB line driver
or single-supply analog-to-digital converter (ADC)
input driver.
APPLICATIONS
•
•
•
•
•
•
•
The low 1.1mA supply current of the OPA890 is
precisely trimmed at +25°C. This trim, along with low
temperature drift, ensures lower maximum supply
current than competing products. System power may
be reduced further using the optional disable control
pin. Leaving this disable pin open, or holding it
HIGH, operates the OPA890 normally. If pulled
LOW, the OPA890 supply current drops to less than
30µA while the output goes into a high-impedance
state.
VIDEO LINE DRIVING
xDSL LINE DRIVERS/RECEIVERS
HIGH-SPEED IMAGING CHANNELS
ADC BUFFERS
PORTABLE INSTRUMENTS
TRANSIMPEDANCE AMPLIFIERS
ACTIVE FILTERS
RELATED
OPERATIONAL AMPLIFIER
PRODUCTS
Multiplying DAC Transimpedance Amplifier
+5V
DESCRIPTION
SINGLES
DUALS
TRIPLES
—
OPA2890
—
Voltage-Feedback Amplifier
with Disable (1800V/µs)
OPA690
OPA2690
OPA3690
Current-Feedback Amplifier
with Disable (2100V/µs)
OPA691
OPA2691
OPA3691
Fixed Gain
OPA692
—
OPA3692
Low-Power Voltage-Feedback
with Disable
VDD
GND
DB0
DB1
VREF
DB2
½
DB3
R1
DAC7822
DB4
RFB
DB5
DB6
IOUT1
DB7
IOUT2
DB8
DB9
R2
DB10
R2_3
DB11
R3
-5V
2.5pF
+7.5V
OPA890
VOUT
0V £ VOUT £ 5V
5.56kW
0.1mF -2.5V
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007, Texas Instruments Incorporated
OPA890
www.ti.com
SBOS369 – MAY 2007
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
ORDERING INFORMATION (1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
OPA890
SO-8
D
–40°C to +85°C
OPA890
OPA890
SOT23-6
DBV
–40°C to +85°C
BRI
(1)
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA890ID
Rail, 75
OPA890IDR
Tape and Reel, 2500
OPA890IDBVT
Tape and Reel, 250
OPA890IDBVR
Tape and Reel, 3000
For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range (unless otherwise noted).
OPA890
UNIT
±6.5
V
Power Supply
Internal Power Dissipation
See Thermal Characteristics
±VS
V
–40 to +125
°C
Lead Temperature (soldering, 10s)
+260
°C
Maximum Junction Temperature (TJ)
+150
°C
Maximum Junction Temperature, Continuous Operation, Long-Term Reliability
+140
°C
Human Body Model (HBM)
2000
V
Charge Device Model (CDM)
1500
V
Machine Model (MM)
200
V
Input Voltage Range
Storage Temperature Range
ESD Rating:
(1)
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond
those specified is not implied.
PIN CONFIGURATIONS
TOP VIEW
SO
NC
1
8
DIS
Inverting Input
2
7
+VS
Noninverting Input
3
6
Output
-VS
4
5
NC
TOP VIEW
SOT23
Output
1
6
+VS
-VS
2
5
DIS
Noninverting Input
3
4
Inverting Input
6
5
4
BRI
NC = No Connection
1
2
3
Pin Orientation/Package Marking
2
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OPA890
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SBOS369 – MAY 2007
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
At RF = 750Ω, G = +2V/V, and RL = 100Ω, unless otherwise noted.
OPA890ID, IDBV
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
+25°C (2)
0°C to
+70°C (3)
–40°C to
+85°C (3)
115
75
65
60
G = +10V/V, VO = 100mVPP
13
9
8
G > +20V/V
130
100
90
G = +2V/V, VO = 100mVPP
20
CONDITIONS
+25°C
G = +1V/V, VO = 100mVPP, RF = 0Ω
260
G = +2V/V, VO = 100mVPP
MIN/
MAX
TEST
LEVEL (1)
MHz
typ
C
MHz
min
B
7.5
MHz
min
B
85
MHz
min
B
MHz
typ
C
UNITS
AC PERFORMANCE
Small-Signal Bandwidth
Gain Bandwidth Product
Bandwidth for 0.1dB Flatness
Peaking at a Gain of +1V/V
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
VO < 100mVPP
1
dB
typ
C
G = +2V/V, VO = 2VPP
170
MHz
typ
C
G = +2V/V, VO = 2V Step
500
V/µs
min
B
0.2V Step
3.5
ns
typ
C
G = +1V/V, VO = 2V Step
16
ns
typ
C
10
ns
typ
C
Settling Time to 0.1%
Harmonic Distortion
2nd-Harmonic
325
300
275
G = +2V/V, f = 1MHz, VO = 2VPP
RL = 200Ω
-88
-78
-76
-75
dBc
max
B
RL ≥ 500Ω
-102
-84
-82
-80
dBc
max
B
RL = 200Ω
-89
-84
-81
-80
dBc
max
B
RL ≥ 500Ω
-94
-90
-87
-86
dBc
max
B
Input Voltage Noise
f > 100kHz
8
9
10
11
nV/√Hz
max
B
Input Current Noise
f > 100kHz
1
1.3
1.7
1.9
pA/√Hz
max
B
Differential Gain
G = +2V/V, VO = 1.4VPP, RL = 150Ω
0.05
%
typ
C
Differential Phase
G = +2V/V, VO = 1.4VPP, RL = 150Ω
0.03
°
typ
C
f = 5MHz, Input-Referred
–68
dB
typ
C
VO = 0V, RL = 100Ω
62
57
56
54
dB
min
A
VCM = 0V
±1
±5
±5.7
±6
mV
max
A
±15
±15
µV/°C
max
B
±1.8
±2
µA
max
A
±5
±6
nA/°C
max
B
±450
±500
nA
max
A
±2.5
±2.5
nA/°C
max
B
3rd-Harmonic
Channel-to-Channel Crosstalk
DC PERFORMANCE (4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Input Bias Current Drift
Input Offset Current
Average Input Offset Current Drift
VCM = 0V
VCM = 0V
±0.1
±1.6
VCM = 0V
VCM = 0V
±70
±350
VCM = 0V
INPUT
±3.9
±3.8
±3.7
±3.6
V
min
A
VCM = 0V, Input-Referred
67
61
58
57
dB
min
A
Differential
VCM = 0V
190 || 0.6
kΩ || pF
typ
C
Common-Mode
VCM = 0V
3.2 || 0.9
MΩ || pF
typ
C
Common-Mode Input Range (CMIR) (5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance
OUTPUT
Output Voltage Swing
Output Current, Sourcing, Sinking
Peak Output Current
Closed-Loop Output Impedance
(1)
(2)
(3)
(4)
(5)
No Load
±4.0
±3.9
±3.8
±3.7
V
min
A
RL = 100Ω
±3.5
±3.1
±3.05
±2.9
V
min
A
VO = 0V
±40
±35
±33
±30
mA
min
A
Output Shorted to Ground
±75
mA
typ
C
G = +2V/V, f = 100kHz
0.04
Ω
typ
C
Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for +25°C tested specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient +2°C at high temperature limit for over
temperature specifications.
Current is considered positive out-of-node. VCM is the input common-mode voltage.
Tested < 3dB below minimum specified CMRR at ±CMIR limits
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3
OPA890
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SBOS369 – MAY 2007
ELECTRICAL CHARACTERISTICS: VS = ±5V (continued)
Boldface limits are tested at +25°C.
At RF = 750Ω, G = +2V/V, and RL = 100Ω, unless otherwise noted.
OPA890ID, IDBV
TYP
PARAMETER
CONDITIONS
DISABLE
Disable LOW
Power-Down Supply Current (+VS)
MIN/MAX OVER TEMPERATURE
+25°C
+25°C (2)
0°C to
+70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
55
60
75
µA
max
A
µs
typ
C
VDIS = 0
30
VIN = 1VDC
7
Enable Time
VIN = 1VDC
200
ns
typ
C
Off Isolation
G = +2V/V, f = 5MHz
70
dB
typ
C
4
pF
typ
C
Disable Time
Output Capacitance in Disable
Enable Voltage
3.0
3.2
3.4
3.8
V
min
A
Disable Voltage
1.4
1.1
1.0
0.8
V
max
A
15
30
35
40
µA
max
A
Control Pin Input Bias Current (VDIS)
VDIS = 0V, Each Channel
POWER SUPPLY
Specified Operating Voltage
±5
V
typ
C
Minimum Operating Voltage
±1.5
V
typ
C
Maximum Operating Voltage
±6.0
±6.0
±6.0
V
max
A
Maximum Quiescent Current
VS = ±5V
1.1
1.2
1.22
1.25
mA
max
A
Minimum Quiescent Current
VS = ±5V
1.1
1.05
1.02
1
mA
min
A
+VS = 4.5V to 5.5V
74
66
62
60
dB
min
A
–40 to +85
°C
typ
C
Power-Supply Rejection Ratio
(+PSRR)
THERMAL CHARACTERISTICS
Specified Operating Range
Thermal Resistance θJA
4
Junction-to-Ambient
D
SO-8
105
°C/W
typ
C
DBV
SOT23-6
110
°C/W
typ
C
Submit Documentation Feedback
OPA890
www.ti.com
SBOS369 – MAY 2007
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
At RF = 750Ω, G = +2V/V, and RL = 100Ω, unless otherwise noted.
OPA890ID, IDBV
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
+25°C (2)
0°C to
+70°C (3)
–40°C to
+85°C (3)
105
70
60
55
G = +10V/V, VO = 100mVPP
12
8
6.8
G > +20V/V
125
90
75
G = +2V/V, VO = 100mVPP
16
CONDITIONS
+25°C
G = +1V/V, VO = 100mVPP, RF = 0Ω
220
G = +2V/V, VO = 100mVPP
MIN/
MAX
TEST
LEVEL (1)
MHz
typ
C
MHz
min
B
6.3
MHz
min
B
70
MHz
min
B
MHz
typ
C
UNITS
AC PERFORMANCE
Small-Signal Bandwidth
Gain Bandwidth Product
Bandwidth for 0.1dB Flatness
Peaking at a Gain of +1V/V
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
VO < 100mVPP
2
dB
typ
C
G = +2V/V, VO = 2VPP
130
MHz
typ
C
G = +2V/V, VO = 2V Step
350
V/µs
min
B
0.2V Step
3.8
ns
typ
C
G = +1V/V, VO = 2V Step
18
ns
typ
C
12
ns
typ
C
Settling Time to 0.1%
Harmonic Distortion
2nd-Harmonic
250
200
175
G = +2V/V, f = 1MHz, VO = 2VPP
RL = 200Ω
-85
-76
-73
-72
dBc
max
B
RL ≥ 500Ω
-90
-78
-74
-73
dBc
max
B
RL = 200Ω
-85
-81
-79
-78
dBc
max
B
RL ≥ 500Ω
-87
-84
-82
-81
dBc
max
B
Input Voltage Noise
f > 100kHz
8.1
9.1
10.1
11.1
nV/√Hz
max
B
Input Current Noise
f > 100kHz
1.1
1.4
1.7
2.0
pA/√Hz
max
B
Differential Gain
G = +2V/V, VO = 1.4VPP, RL = 150Ω
0.06
%
typ
C
Differential Phase
G = +2V/V, VO = 1.4VPP, RL = 150Ω
0.04
°
typ
C
f = 5MHz, Input-Referred
-68
dB
typ
C
VO = VS/2, RL = 100Ω
60
55
54
52
dB
min
A
VCM = VS/2
±1
±5
±5.7
±6
mV
max
A
±15
±15
µV/°C
max
B
±1.9
±2.1
µA
max
A
±5
±6
nA/°C
max
B
±500
±550
nA
max
A
±2.5
±2.5
nA/°C
max
B
3rd-Harmonic
Channel-to-Channel Crosstalk
DC PERFORMANCE (4)
Open-Loop Voltage Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Input Bias Current Drift
Input Offset Current
Average Input Offset Current Drift
VCM = VS/2
VCM = VS/2
±0.1
±1.7
VCM = VS/2
VCM = VS/2
±70
±400
VCM = VS/2
INPUT
Most Positive Input Voltage (5)
+4
+3.8
+3.75
+3.7
V
min
A
Least Positive Input Voltage (5)
+1
+1.2
+1.2
+1.3
V
max
A
VCM = VS/2, Input-Referred
65
59
56
55
dB
min
A
Differential
VCM = VS/2
190 || 0.6
kΩ || pF
typ
C
Common-Mode
VCM = VS/2
3.2 || 0.9
MΩ || pF
typ
C
Common-Mode Rejection Ratio (CMRR)
Input Impedance
OUTPUT
Most Positive Output Voltage
Least Positive Output Voltage
Output Current: Sourcing, Sinking
Short-Circuit Output Current
Closed-Loop Output Impedance
(1)
(2)
(3)
(4)
(5)
No Load
+4.0
+3.9
+3.85
+3.8
V
min
A
RL = 100Ω
+3.9
+3.75
+3.7
+3.65
V
min
A
No Load
+1.0
+1.1
+1.15
+1.2
V
max
A
RL = 100Ω
+1.1
+1.35
+1.4
+1.45
V
max
A
VO = VS/2
±35
±30
±28
±25
mA
min
A
Output Shorted to Ground
±65
mA
typ
C
G = +2V/V, f = 100kHz
0.04
Ω
typ
C
Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for +25°C tested specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient +2°C at high temperature limit for over
temperature specifications.
Current is considered positive out-of-node. VCM is the input common-mode voltage.
Tested < 3dB below minimum specified CMRR at ±CMIR limits
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OPA890
www.ti.com
SBOS369 – MAY 2007
ELECTRICAL CHARACTERISTICS: VS = +5V (continued)
Boldface limits are tested at +25°C.
At RF = 750Ω, G = +2V/V, and RL = 100Ω, unless otherwise noted.
OPA890ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
+25°C
+25°C (2)
0°C to
+70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
VDIS = 0V, both channels
18
45
50
65
µA
max
A
VOUT = 1VDC
7
ns
typ
C
Enable Time
VOUT = 1VDC
200
ns
typ
C
Off Isolation
G = +2V/V, f = 5MHz
70
dB
typ
C
4
pF
typ
C
PARAMETER
CONDITIONS
DISABLE
TEST
LEVEL (1)
Disable LOW
Power-Down Supply Current (+VS)
Disable Time
Output Capacitance in Disable
Enable Voltage
3.0
3.2
3.4
3.8
V
min
A
Disable Voltage
1.4
1.1
1.0
0.8
V
max
A
15
30
35
40
µA
max
A
Control Pin Input Bias Current (VDIS)
VDIS = 0V, Each Channel
POWER SUPPLY
Specified Operating Voltage
+5
V
typ
C
Minimum Operating Voltage
+3
V
typ
C
Maximum Operating Voltage
+12
+12
+12
V
max
A
A
Maximum Quiescent Current
VS = +5V
1.06
1.18
1.20
1.25
mA
max
Minimum Quiescent Current
VS = +5V
1.06
0.92
0.90
0.87
mA
min
A
+VS = 4.5V to 5.5V
65
dB
typ
C
–40 to +85
°C
typ
C
Power-Supply Rejection Ratio
(+PSRR)
THERMAL CHARACTERISTICS
Specified Operating Range
Thermal Resistance θJA
6
Junction-to-Ambient
D
SO-8
105
°C/W
typ
C
DBV
SOT23-6
110
°C/W
typ
C
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OPA890
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SBOS369 – MAY 2007
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, G = +2V/V, RF = 750Ω, and RL = 200Ω, unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
3
G = +1V/V
R F = 0W
0
1VPP
6
-3
2VPP
3
Gain (dB)
Normalized Gain (dB)
LARGE-SIGNAL FREQUENCY RESPONSE
9
-6
-9
0
4VPP
G = +2V/V
-3
-12
7VPP
G = +5V/V
-15
VO = 0.1VPP
-6
G = +10V/V
-18
1
10
RL = 200W
G = +2V/V
-9
100
1
600
10
Figure 1.
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
3
VO = 0.5VPP
G = +2V/V
VO = 5VPP
G = +2V/V
2
200
Output Voltage (V)
Output Voltage (mV)
400
Figure 2.
400
300
100
Frequency (MHz)
Frequency (MHz)
100
0
-100
1
0
-1
-200
-2
-300
-3
-400
Time (10ns/div)
Time (10ns/div)
Figure 3.
Figure 4.
VIDEO DIFFERENTIAL GAIN/DIFFERENTIAL PHASE
-dP
0.18
-45
0.36
-50
0.32
-dG
0.14
0.28
0.12
0.24
0.10
0.20
+dG
0.08
0.16
+dP
0.06
0.12
0.04
0.08
0.02
0.04
0
0
1
2
3
Number of 150W Loads
4
Differential Phase (°)
Differential Gain (%)
0.16
DISABLE FEEDTHROUGH
0.40
Disable Feedthrough (dB)
0.20
VDIS = 0V
Input Referred
-55
-60
-65
-70
-75
-80
-85
-90
1
10
100
Frequency (MHz)
Figure 5.
Figure 6.
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OPA890
www.ti.com
SBOS369 – MAY 2007
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2V/V, RF = 750Ω, and RL = 200Ω, unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
1MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
-80
VO = 2VPP
f = 1MHz
G = +2V/V
-85
3rd Harmonic
-90
-95
2nd Harmonic
-100
-105
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
-80
VO = 2VPP
RL = 200W
G = +2V/V
-85
3rd Harmonic
-90
2nd Harmonic
-95
-100
-110
100
2.5
1k
3.0
Load Resistance (W)
5.0
5.5
6.0
HARMONIC DISTORTION vs OUTPUT VOLTAGE
-70
VO = 2VPP
RL = 200W
G = +2V/V
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
4.5
Figure 8.
HARMONIC DISTORTION vs FREQUENCY
-60
4.0
Supply Voltage (±VS)
Figure 7.
-50
3.5
-70
3rd Harmonic
-80
-90
2nd Harmonic
-100
RL = 200W
f = 1MHz
G = +2V/V
-75
-80
3rd Harmonic
-85
-90
-95
2nd Harmonic
-100
-110
0.1
1
0.1
10
Frequency (MHz)
Figure 9.
Harmonic Distortion (dBc)
3rd Harmonic
-80
-85
HARMONIC DISTORTION vs INVERTING GAIN
-70
Harmonic Distortion (dBc)
VO = 2VPP
RL = 200W
f = 1MHz
-75
10
Figure 10.
HARMONIC DISTORTION vs NONINVERTING GAIN
-70
1
Output Voltage Swing (VPP)
2nd Harmonic
-90
-95
-75
VO = 2VPP
RL = 200W
f = 1MHz
3rd Harmonic
2nd Harmonic
-80
-85
-100
-105
-90
1
8
10
20
-1
-10
Gain (V/V)
Gain (V/V)
Figure 11.
Figure 12.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2V/V, RF = 750Ω, and RL = 200Ω, unless otherwise noted.
LOW-FREQUENCY INVERTING HARMONIC DISTORTION
TWO-TONE, 3RD-ORDER INTERMODULATION SPURIOUS
-40
VO = 2VPP
RL = 500W
G = -1V/V
-95
Load Power at Matched 50W Load
-50
Spurious Point (dBc)
Harmonic Distortion (dBc)
-90
-100
2nd Harmonic
-105
-110
3rd Harmonic
10MHz
-60
5MHz
-70
-80
-90
1MHz
-115
-100
-120
-110
1k
10k
100k
1M
-8
-6
-4
-2
0
2
4
Frequency (Hz)
Single-Tone Load Power (dBm)
Figure 13.
Figure 14.
RECOMMENDED RS vs CAPACITIVE LOAD
6
8
FREQUENCY RESPONSE vs CAPACITIVE LOAD
9
100
G = +2V/V
6
Gain (dB)
RS (W)
CL = 10pF
3
10
CL = 100pF
CL = 22pF
0
CL = 47pF
-3
RS
VIN
VOUT
OPA890
CL
1kW(1)
750W
-6
750W
NOTE: (1) 1kW is optional.
-9
1
1
10
100
0
1000
20
40
60
80
Figure 15.
Figure 16.
COMMON-MODE REJECTION RATIO AND
POWER-SUPPLY REJECTION RATIO vs FREQUENCY
INPUT VOLTAGE AND CURRENT NOISE
80
100
Voltage Noise Density (nV/ÖHz)
Current Noise Density (pA/ÖHz)
-PSRR
70
CMRR and PSRR (dB)
100 120 140 160 180 200
Frequency (MHz)
Capacitive Load (pF)
CMRR
60
+PSRR
50
40
30
20
10
0
Voltage Noise Density (8nV/ÖHz)
10
Current Noise Density (1pA/ÖHz)
1
0.1
1k
10k
100k
1M
10M
100M
10
Frequency (Hz)
100
1k
10k
100k
1M
10M
Frequency (Hz)
Figure 17.
Figure 18.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, G = +2V/V, RF = 750Ω, and RL = 200Ω, unless otherwise noted.
50
1.14
49
Supply Current
Output Current,
Sourcing
47
46
1.10
45
1.09
44
43
Output Current,Sinking
2.00
1.95
100
1.90
50
1.85
42
0
Input Offset Voltage (VOS)
1.80
-50
41
1.06
1.75
40
1.05
-50
0
-25
25
50
75
100
-100
-50
125
-25
0
25
50
75
100
125
Ambient Temperature (°C)
Ambient Temperature (°C)
Figure 19.
Figure 20.
LARGE-SIGNAL DISABLE/ENABLE RESPONSE
6
3
-2
3
2
Output Voltage (V)
4
4
VDIS (V)
8
0
4
NONINVERTING OVERDRIVE RECOVERY
6
2
Output Voltage (V)
150
Input Offset Current (IOS)
4
2
2
Output Voltage
Left Scale
1
0
0
Input Voltage
Right Scale
-2
-1
1
-4
-3
0
-6
-3
-8
-1
-4
Time (5ns/div)
Time (10ns/div)
Figure 21.
Figure 22.
CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY
OPEN-LOOP GAIN AND PHASE
100
80
180
10
324W
ZO
OPA890
Open-Loop Gain (dB)
Output Impedance (W)
70
750W
1
160
Open-Loop Gain
60
120
40
100
Open-Loop Phase
30
80
20
60
0.01
10
40
0
20
0.001
-10
750W
0.1
1k
10k
100k
1M
10M
100M
0
100
1k
Frequency (Hz)
10k
100k
1M
10M
Frequency (Hz)
Figure 23.
10
140
50
Figure 24.
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100M
1G
Open-Loop Phase (°)
1.07
200
Input Voltage (V)
1.11
1.08
2.05
48
1.12
250
Input Bias Current (IB)
Output Current (mA)
Supply Current (mA)
1.13
2.10
Input Offset Voltage (V)
1.15
TYPICAL DC DRIFT vs TEMPERATURE
Input Bias and Input Offset Currents (nA)
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
OPA890
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TYPICAL CHARACTERISTICS: VS = ±5V, Differential
At TA = +25°C, Differential Gain = +2V/V, RF = 750Ω, and RL = 400Ω, unless otherwise noted.
DIFFERENTIAL SMALL-SIGNAL FREQUENCY RESPONSE
DIFFERENTIAL LARGE-SIGNAL FREQUENCY RESPONSE
3
9
GD = 1V/V
6
GD = 5VPP
-3
3
Gain (dB)
Normalized Gain (dB)
0
-6
GD = 2V/V
-9
GD = 5V/V
-12
0
GD = 14VPP
-3
GD = 10V/V
-15
-6
RF = 750W
RL = 400W
-18
GD = 8VPP
-9
1
10
100
300
1
10
Figure 25.
DIFFERENTIAL DISTORTION vs FREQUENCY
-30
-75
-40
Harmonic Distortion (dBc)
3rd Harmonic
-85
-90
-95
2nd Harmonic
-105
-110
-115
-120
VO = 4VPP
f = 1MHz
GD = 2V/V
RL = 400W
GD = 2V/V
3rd Harmonic
-50
-60
-70
-80
2nd Harmonic
-90
-100
-110
-120
100
1
1k
10
Load Resistance (W)
Frequency (MHz)
Figure 27.
Figure 28.
20
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
-75
-80
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
-100
300
Figure 26.
-70
-80
100
Frequency (MHz)
Frequency (MHz)
RL = 400W
f = 1MHz
GD = 2V/V
3rd Harmonic
-85
-90
-95
-100
2nd Harmonic
-105
-110
0.1
1
10
Output Voltage (VPP)
Figure 29.
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TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, G = +2V/V, RF = 750Ω, and RL = 200Ω, unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
3
G = +1V/V
RF = 0W
0
6
1VPP
-3
3
Gain (dB)
Normalized Gain (dB)
LARGE-SIGNAL FREQUENCY RESPONSE
9
-6
-9
G = +2V/V
2VPP
-3
-12
3VPP
G = +5V/V
-6
-15
VO = 100mVPP
0
RL = 200W
G = +2V/V
G = +10V/V
-18
-9
1
10
100
1
500
10
Figure 30.
300
Figure 31.
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
2.9
4.1
VO = 0.5VPP
G = +2V/V
2.8
VO = 0.5VPP
G = +2V/V
3.7
2.7
Output Voltage (V)
Output Voltage (V)
100
Frequency (MHz)
Frequency (MHz)
2.6
2.5
2.4
2.3
2.2
3.3
2.9
2.5
2.1
1.7
1.3
2.1
0.9
Time (10ns/div)
Time (10ns/div)
Figure 32.
Figure 33.
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
9
200
RS
VIN
VOUT
OPA890
100
CL
6
1kW(1)
750W
750W
NOTE: (1) 1kW is optional.
Gain (dB)
RS (W)
3
10
0
CL = 10pF
CL = 22pF
-3
CL = 47pF
-6
1
CL = 100pF
-9
1
10
100
1000
0
20
Figure 34.
12
40
60
80
100 120 140 160 180 200
Frequency (MHz)
Capacitive Load (pF)
Figure 35.
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TYPICAL CHARACTERISTICS: VS = +5V (continued)
At TA = +25°C, G = +2V/V, RF = 750Ω, and RL = 200Ω, unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
4.0
4.5
3.5
3.5
3.0
Output Voltage
Left Scale
2.5
2.5
Input Voltage
Right Scale
1.5
2.0
0.5
1.5
-0.5
1.0
Harmonic Distortion (dBc)
4.5
5.5
-75
Input Voltage (1V/div)
6.5
-80
VO = 2VPP
f = 1MHz
GD = +2V/V
-85
3rd Harmonic
-90
2nd Harmonic
-95
0.5
-1.5
Time (10ns/div)
100
1k
Load Resistance (W)
Figure 36.
Figure 37.
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs OUTPUT VOLTAGE
-45
-50
3rd Harmonic
-70
Harmonic Distortion (dBc)
-60
VO = 2VPP
RL = 200W to VS/2
G = +2V/V
2nd Harmonic
-80
-90
-55
f = 1MHz
G = +2V/V
RL = 200W to VS/2
-65
2nd Harmonic
-75
3rd Harmonic
-85
-95
-100
0.1
0.1
10
1
1
Frequency (MHz)
Output Voltage Swing (VPP)
Figure 38.
Figure 39.
10
TWO-TONE, 3RD-ORDER INTERMODULATION SPURIOUS
-40
Load Power at Matched 50W Load
10MHz
-50
Spurious Point (dBc)
Harmonic Distortion (dBc)
Output Voltage (1V/div)
NONINVERTING OVERDRIVE RECOVERY
-60
5MHz
-70
-80
1MHz
-90
-100
-8
-7
-6
-5
-4
-3
-2
-1
0
1
2
Single-Tone Load Power (dBm)
Figure 40.
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TYPICAL CHARACTERISTICS: VS = +5V, Differential
At TA = +25°C, Differential Gain = +2V/V, RF = 750Ω, and RL = 400Ω, unless otherwise noted.
DIFFERENTIAL SMALL-SIGNAL FREQUENCY RESPONSE
DIFFERENTIAL LARGE-SIGNAL FREQUENCY RESPONSE
6
9
GD = 2V/V
6
0
3
-3
Gain (dB)
Normalized Gain (dB)
3
GD = 1V/V
R F = 0W
-6
-9
-15
-18
1VPP
-6
RF = 750W
RL = 400W
GD = 10V/V
10
1
0
-3
GD = 5V/V
-12
4VPP
-9
100
200
Figure 41.
Figure 42.
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
Harmonic Distortion (dBc)
-85
VO = 4VPP
f = 1MHz
GD = 2V/V
-90
-95
-100
-105
-110
2nd Harmonic
-115
RL = 400W
f = 1MHz
GD = 2V/V
-50
3rd Harmonic
-80
-60
3rd Harmonic
-70
-80
-90
2nd Harmonic
-100
-110
-120
-125
-120
100
1k
1
10
Load Resistance (W)
Frequency (MHz)
Figure 43.
Figure 44.
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
Harmonic Distortion (dBc)
-60
-70
-80
3rd Harmonic
-90
-100
-110
2nd Harmonic
-120
-130
0.1
1
Output Voltage Swing (VPP)
Figure 45.
14
300
DIFFERENTIAL DISTORTION vs FREQUENCY
-40
-75
100
Frequency (MHz)
-70
Harmonic Distortion (dBc)
10
1
Frequency (MHz)
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APPLICATION INFORMATION
WIDEBAND VOLTAGE-FEEDBACK
OPERATION
+5V
The OPA890 provides an exceptional combination of
low quiescent current with a wideband, unity-gain
stable, voltage-feedback op amp using a new high
slew rate input stage. Typical differential input stages
used for voltage-feedback op amps are designed to
steer a fixed-bias current to the compensation
capacitor, setting a limit to the achievable slew rate.
The OPA890 uses an input stage that places the
transconductance element between two input
buffers, using the combined output currents as the
forward signal. As the error voltage increases across
the two inputs, an increasing current is delivered to
the compensation capacitor. This increasing current
provides very high slew rate (500V/µs) while
consuming relatively low quiescent current (1.1mA).
This exceptional full-power performance comes at
the price of a slightly higher input noise voltage than
alternative architectures. The 8nV/√Hz input voltage
noise for the OPA890 is low for this combination of
input stage and low quiescent current.
Figure 46 shows the dc-coupled, gain of +2, dual
power-supply circuit configuration used as the basis
of the ±5V Electrical Characteristics and Typical
Characteristics. For test purposes, the input
impedance is set to 50Ω with a resistor to ground
and the output impedance is set to 50Ω with a series
output resistor. Voltage swings reported in the
Typical Characteristics are taken directly at the input
and output pins, while output powers (dBm) are at
the matched 50Ω load. For the circuit of Figure 46,
the total effective load will be 100Ω 1.5kΩ. The
disable control line is typically left open to ensure
normal amplifier operation. Two optional components
are included in Figure 46. An additional resistor
(324Ω) is included in series with the noninverting
input. Combined with the 25Ω dc source resistance
looking back towards the signal generator, this
configuration gives an input bias current cancelling
resistance that matches the 375Ω source resistance
seen at the inverting input (see the DC Accuracy and
Offset Control section). In addition to the usual
power-supply decoupling capacitors to ground, a
0.1µF capacitor is included between the two
power-supply pins. In practical printed circuit board
(PCB) layouts, this optional-added capacitor typically
improves the 2nd-harmonic distortion performance
by 3dB to 6dB.
0.1mF
50W Source
VI
6.8mF
+
324W
50W
DIS
VO
50W
50W Load
OPA890
0.1mF
RF
750W
RG
750W
+
6.8mF
0.1mF
-5V
Figure 46. DC-Coupled, G = +2, Bipolar Supply,
Specification and Test Circuit
Figure 47 shows the ac-coupled, gain of +2,
single-supply circuit configuration used as the basis
of the +5V Electrical Characteristics and Typical
Characteristics. Though not a rail-to-rail design, the
OPA890 requires minimal input and output voltage
headroom compared to other very wideband
voltage-feedback op amps. It delivers a 2VPP output
swing on a single +5V supply with > 100MHz
bandwidth. The key requirement of broadband
single-supply operation is to maintain input and
output signal swings within the usable voltage ranges
at both the input and the output. The circuit of
Figure 47 establishes an input midpoint bias using a
simple resistive divider from the +5V supply (two
698Ω resistors). The input signal is then ac-coupled
into the midpoint voltage bias. The input voltage can
swing to within 1.5V of either supply pin, giving a
2VPP input signal range centered between the supply
pins. The input impedance matching resistor (59Ω)
used for testing is adjusted to give a 50Ω input load
when the parallel combination of the biasing divider
network is included.
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MULTIPLYING DAC SINGLE-ENDED OUTPUT
TRANSIMPEDANCE AMPLIFIER
+5V
+VS
0.1mF
50W Source
0.1mF
VI
59W
Multiplyings digital-to-analog converters (DACs),
such as the DAC7822, can make good use of the
low-power, high slew rate amplifier, OPA890.
6.8mF
698W
50W
698W
The frequency response of the schematic shown in
Figure 48 is shown in Figure 49.
DIS
VO
OPA890
100W
VS/2
RF
750W
RG
750W
0.1mF
Figure 47. AC-Coupled, G = +2, Single-Supply,
Specification and Test Circuit
Again, an additional resistor (50Ω, in this case) is
included directly in series with the noninverting input.
This minimum recommended value provides part of
the dc source resistance matching for the
noninverting input bias current. It is also used to form
a simple parasitic pole to roll off the frequency
response at very high frequencies ( > 500MHz) using
the input parasitic capacitance to form a bandlimiting
pole. The gain resistor (RG) is ac-coupled, giving the
circuit a dc gain of +1, which puts the input dc bias
voltage (2.5V) at the output as well. The voltage can
swing to within 1.35V of either supply pin. Driving a
demanding 100Ω load to a midpoint bias is used in
this characterization circuit. Higher swings are
possible using a lighter load.
+5V
VDD
GND
DB0
DB1
VREF
DB2
½
DB3
R1
DAC7822
DB4
RFB
DB5
DB6
IOUT1
DB7
IOUT2
DB8
DB9
R2
DB10
R2_3
DB11
R3
-5V
+7.5V
2.5pF
OPA890
VOUT
0V £ VOUT £ 5V
5.56kW
0.1mF -2.5V
Figure 48. DAC Transimpedance Amplifier
83
77
71
Gain (dB)
+
65
59
53
47
41
100k
1M
10M
100M
Frequency (Hz)
Figure 49. OPA2890 (as DAC Transimpedance
Amplifier) Frequency Response
16
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Driving a light load, the OPA890 can output ±4V over
±5V supplies. Setting the reference voltage to –5V
results in an output voltage swing from 0V to 5V. In
order to optimize the OPA2890 operation for this
application, the supply voltages have been adjusted
so that the output voltage swing is balanced around
mid-supply of the amplifier. Note that as a result of
the internal architecture of the multiplying DAC, the
IOUT1 output is not high impedance. The IOUT1 output
resistance is between 4.5kΩ and 22.1kΩ (excluding
code 000h) for a 10kΩ nominal VREF input
resistance. IOUT1 output resistance changes are
directly related to the code change. This low
impedance has multiple effects when a bipolar
technology amplifier is used.
Some of these effects are:
• The noise gain of the amplifier changes for each
code.
• The output offset voltage of the amplifier changes
for each code, because of the input offset
voltage.
• The input bias current cannot be cancelled. The
effects of the input bias current can be reduced,
but not eliminated, thereby affecting the total
output offset voltage of the amplifier with each
code.
• The noninverting pin of the amplifier must be tied
to ground and cannot be used to create a dc
offset on the output amplifier, as is the case for
the transimpedance amplifier.
The following analysis excludes the input offset
current.
The total output offset voltage variations because of
code changing in the DAC can be expressed as:
∆VOSO = +∆NG {[(RF
ROUT1) – RS] + VOS}
Where:
4.5kΩ ≤ ROUT1 ≤ 22.1kΩ
RF = 10kΩ
Notice that most of the error occurs mainly at the first
codes (0, 1, 2); excluding these codes from the
analysis yields the following results, shown in
Table 1.
Table 1. DC Accuracy vs Code
CODES
TOTAL ERROR DUE TO
VOS and IB
All codes
3.9LSB
Excluding code 0
2.5LSB
Excluding codes 0 and 1
2LSB
Excluding codes 0, 1, and 2
1.83LSB
Note that 1LSB = 1.221mV in the example shown in
Figure 48
If more precision is required while maintaining the ac
performance, a FET-input amplifier (such as the
OPA656 or the THS4631) is a good alternative.
Figure 48 shows a single-ended output drive
implementation. In this circuit, only one side of the
complementary output drive signal is used. A dual
amplifier, such as the OPA2890, provides both
output drivers for the DAC7822. If even lower
quiescent current is needed, the OPA2889 can be
used instead, with minor modifications. The diagram
shows the signal output current connected into the
virtual ground summing junction of the OPA890,
which is set up as a transimpedance stage or I-V
converter. The unused current output of the DAC is
connected to ground. The dc gain for this circuit is
equal to RF. At high frequencies, the DAC output
capacitance produces a zero in the noise gain for the
OPA890 that may cause peaking in the closed-loop
frequency response. CF is added across RF to
compensate for this noise gain peaking. To achieve
a flat transimpedance frequency response, the pole
in the feedback network should be set to:
1
+
2pR FCF
Using the previous values, the variation of the
parallel combination of RF and ROUT1 can be
constrained to: 4.19kΩ≤ (RF ROUT1) ≤ 6.88kΩ. In
order to optimize the bias current cancellation, we
select RS to be the average of those limiting
numbers, or RS = (6.88kΩ + 4.19kΩ)/2 = 5.56kΩ.
Looking at the variation for each code, the total error
(when including all codes) is ~3.9LSB for the
OPA890.
GBP
Ǹ4pR
C
F
D
which
gives
a
closed-loop
bandwidth, f–3dB, of approximately:
f *3dB +
GBP
Ǹ2pR
C
F
D
(2)
transimpedance
(3)
Using the DAC7822 internal output capacitance of
25pF gives a feedback capacitance (CF) of 2.5pF
and an 8.8MHz bandwidth.
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SINGLE-SUPPLY ACTIVE FILTERS
The high bandwidth provided by the OPA890, while
operating on a single +5V supply, lends itself well to
high-frequency active filter designs. Again, the key
additional requirement is to establish the dc
operating point of the signal near the supply midpoint
for highest dynamic range. See Figure 50 for an
example design of a 5MHz low-pass Butterworth
filter using the Sallen-Key topology.
Both the input signal and the gain setting resistor are
ac-coupled using 0.1µF blocking capacitors (actually
giving band pass response with the low-frequency
pole set to 32kHz for the component values shown).
As discussed for Figure 47, this configuration allows
the midpoint bias formed by the two 1.87kΩ resistors
to appear at both the input and output pins. The
midband signal gain is set to +4 (12dB) in this case.
The capacitor to ground on the noninverting input is
intentionally set larger to dominate input parasitic
terms. At a gain of +4, the OPA890 on a single
supply shows ~30MHz small- and large-signal
bandwidth. The resistor values have been slightly
adjusted to account for this limited bandwidth in the
amplifier stage. Tests of this circuit show a precise
5MHz, –3dB point with a maximally flat passband
(above the 32kHz ac-coupling corner), and a
maximum stop band attenuation of 24dB at the
amplifier –3dB bandwidth of 30MHz.
Note that the dc impedance looking out of each input
for this circuit has been set to 1.5kΩ to reduce the
output offset voltage retaining maximum signal swing
for a mid supply nominal operating voltage at the
output.
+5V
15
12
100pF
1.87kW
DIS
432W
VI
4VI
OPA890
1.87kW
150pF
1.5kW
500W
0.1mF
5MHz,
2nd-Order,
Butterworth
Filter
Gain (dB)
9
0.1mF 137W
6
3
0
-3
-6
100k
1M
Frequency (Hz)
Figure 50. Single-Supply, High-Frequency Active Filter
18
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DESIGN-IN TOOLS
DEMONSTRATION FIXTURES
Two printed circuit boards (PCBs) are available to
assist in the initial evaluation of circuit performance
using the OPA890 in its two package options. Both
of these are offered free of charge as unpopulated
PCBs, delivered with a user's guide. The summary
information for these fixtures is shown in Table 2.
Table 2. Demonstration Board Summary
PRODUCT
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
OPA890ID
SO-8
DEM-OPA-SO-1A
SBOU009
OPA890IDBV
SOT23-6
DEM-OPA-SOT-1A
SBOU010
The demonstration fixtures can be requested at the
Texas Instruments web site (www.ti.com) through the
OPA890 product folder.
MACROMODELS AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This
practice is particularly true for video and RF amplifier
circuits where parasitic capacitance and inductance
can have a major effect on circuit performance. A
SPICE model for the OPA890 is available through
the Texas Instruments web page (www.ti.com).
These models do a good job of predicting
small-signal ac and transient performance under a
wide variety of operating conditions. They do not do
as well in predicting the harmonic distortion or dG/dP
characteristics. These models do not attempt to
distinguish between package types in the
small-signal ac performance.
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OPERATING SUGGESTIONS
OPTIMIZING RESISTOR VALUES
Because the OPA890 is a unity-gain stable,
voltage-feedback op amp, a wide range of resistor
values can be used for the feedback and gain setting
resistors. The primary limits on these values are set
by dynamic range (noise and distortion) and parasitic
capacitance considerations. Usually, for G > 1
applications, the feedback resistor value should be
between 200Ω and 1.5kΩ. Below 200Ω, the
feedback network presents additional output loading
that can degrade the harmonic distortion
performance of the OPA890. Above 1.5kΩ, the
typical parasitic capacitance (approximately 0.2pF)
across the feedback resistor may cause unintentional
band-limiting in the amplifier response.
The combined impedance of RF RG interacts with
the inverting input capacitance, placing an additional
pole in the feedback network and thus, a zero in the
forward response. Assuming a 2pF total parasitic on
the inverting node, having RF RG < 400Ω keeps
this pole above 250MHz. By itself, this constraint
implies that the feedback resistor RF can increase to
several kΩ at high gains. This increase is
acceptable, as long as the pole formed by RF and
any parasitic capacitance appearing in parallel is
kept out of the frequency range of interest.
BANDWIDTH VERSUS GAIN
Noninverting Amplifier Operation
Voltage-feedback op amps exhibit decreasing
closed-loop bandwidth as the signal gain is
increased. In theory, this relationship is described by
the gain bandwidth product (GBP) shown in the
Electrical Characteristics. Ideally, dividing GBP by
the noninverting signal gain (also called the noise
gain, or NG) predicts the closed-loop bandwidth. In
practice, this relationship only holds true when the
phase margin approaches 90°, as it does in
high-gain configurations. At low gains (increased
feedback factors), most amplifiers exhibit a more
complex response with lower phase margin. The
OPA890 is compensated to give a slightly peaked
response in a noninverting gain of 2V/V (see
Figure 46). This compensation results in a typical
gain of +2V/V bandwidth of 115MHz, far exceeding
that predicted by dividing the 130MHz GBP by 2.
Increasing the gain causes the phase margin to
approach 90° and the bandwidth to more closely
20
approach the predicted value of (GBP/NG). At a gain
of +10V/V, the 13MHz bandwidth shown in the
Electrical Characteristics agrees with that predicted
using the simple formula and the typical GBP of
130MHz.
The OPA890 exhibits minimal bandwidth reduction
going to single-supply (+5V) operation as compared
with ±5V. This difference in performance occurs
because the internal bias control circuitry retains
nearly constant quiescent current as the total supply
voltage between the supply pins is changed.
Inverting Amplifier Operation
The OPA890 is a general-purpose, wideband
voltage-feedback op amp; therefore, all of the
familiar op amp application circuits are available to
the designer. Inverting operation is one of the more
common
requirements
and
offers
several
performance benefits. Figure 51 shows a typical
inverting configuration where the I/O impedances
and signal gain from Figure 46 are retained in an
inverting circuit configuration.
In the inverting configuration, three key design
considerations must be noted. First, the gain resistor
(RG) becomes part of the signal channel input
impedance. If input impedance matching is desired
(which is beneficial whenever the signal is coupled
through a cable, twisted-pair, long PCB trace, or
other transmission line conductor), RG may be set
equal to the required termination value and RF
adjusted to give the desired gain. This approach is
the simplest, and results in optimum bandwidth and
noise performance. However, at low inverting gains,
the resultant feedback resistor value can present a
significant load to the amplifier output. For an
inverting gain of –2V/V, setting RG to 50Ω for input
matching eliminates the need for RM but requires a
100Ω feedback resistor. This option has the
interesting advantage that the noise gain becomes
equal to 2V/V for a 50Ω source impedance—the
same as the noninverting circuits considered in the
previous section. The amplifier output, however, now
sees the 100Ω feedback resistor in parallel with the
external load. In general, the feedback resistor
should be limited to a range of 200Ω to 1.5kΩ. In this
case, it is preferable to increase both the RF and RG
values, as shown in Figure 51, and then achieve the
input matching impedance with a third resistor (RM)
to ground. The total input impedance becomes the
parallel combination of RG and RM.
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DRIVING CAPACITIVE LOADS
+5V
+
0.1mF
6.8mF
0.1mF
DIS
RB
240W
50W
Source
RO
50W
0.1mF OPA890
50W Load
RF
750W
RG
324W
RM
59W
0.1mF
+
6.8mF
-5V
Figure 51. Gain of –2V/V Example Circuit
The second major consideration, touched on in the
previous paragraph, is that the signal source
impedance becomes part of the noise gain equation
and influences the bandwidth. For the example in
Figure 51, the RM value combines in parallel with the
external 50Ω source impedance, yielding an effective
driving impedance of 50Ω
59Ω = 27Ω. This
impedance is added in series with RG for calculating
the noise gain (NG). The resulting NG is 3.14V/V for
Figure 51, as opposed to only 2 if RM could be
eliminated as discussed previously. The bandwidth is
therefore slightly lower for the gain of –2V/V circuit of
Figure 51 than for the gain of +2V/V circuit of
Figure 46.
The third important consideration in inverting
amplifier design is setting the bias current
cancellation resistor on the noninverting input (RB). If
this resistor is set equal to the total dc resistance
looking out of the inverting node, the output dc error
(because of the input bias currents) is reduced to
(Input Offset Current) × RF. If the 50Ω source
impedance is dc-coupled in Figure 51, the total
resistance to ground on the inverting input is 351Ω.
Combining this resistance in parallel with the
feedback resistor gives the value of RB = 240Ω used
in this example. To reduce the additional
high-frequency noise introduced by this resistor, it is
sometimes bypassed with a capacitor. As long as RB
< 350Ω, a capacitor is not required because the total
noise contribution of all other terms is less than that
of the op amp input noise voltage. As a minimum,
the OPA890 requires an RB value of 50Ω to damp
out parasitic-induced peaking—a direct short to
ground on the noninverting input runs the risk of a
very high-frequency instability in the input stage.
One of the most demanding and yet very common
load conditions for an op amp is capacitive loading.
Often, the capacitive load is the input of an
ADC—including additional external capacitance that
may be recommended to improve ADC linearity. A
high-speed, high open-loop gain amplifier such as
the OPA890 can be very susceptible to decreased
stability and closed-loop response peaking when a
capacitive load is placed directly on the output pin.
When the amplifier open-loop output resistance is
considered, this capacitive load introduces an
additional pole in the signal path that can decrease
the phase margin. Several external solutions to this
problem have been suggested. When the primary
considerations are frequency response flatness,
pulse response fidelity, and/or distortion, the simplest
and most effective solution is to isolate the capacitive
load from the feedback loop by inserting a
series-isolation resistor between the amplifier output
and the capacitive load. This solution does not
eliminate the pole from the loop response, but rather
shifts it and adds a zero at a higher frequency. The
additional zero acts to reduce the phase lag from the
capacitive load pole, thus increasing the phase
margin and improving stability.
The Typical Characteristics show the recommended
RS versus capacitive load and the resulting
frequency response at the load. Parasitic capacitive
loads greater than 2pF can begin to degrade the
performance of the OPA890. Long PCB traces,
unmatched cables, and connections to multiple
devices can easily exceed this value. Always
consider this effect carefully, and add the
recommended series resistor as close as possible to
the OPA890 output pin (see the Board Layout
Guidelines section).
NOISE PERFORMANCE
The input-referred voltage noise, and the two
input-referred current noise terms, combine to give
low output noise under a wide variety of operating
conditions. Figure 52 shows the op amp noise
analysis model with all the noise terms included. In
this model, all noise terms are taken to be noise
voltage or current density terms in either nV/√Hz or
pA/√Hz.
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DC ACCURACY AND OFFSET CONTROL
ENI
RS
EO
OPA890
IBN
RF
ERS
Ö 4kTRS
IBI
RG
4kT
RG
Ö 4kTRF
4kT = 1.6E - 20J
at 290°K
Figure 52. Op Amp Noise Analysis Model
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 4 shows the
general form for the output noise voltage using the
terms shown in Figure 52.
EO +
ǸǒE
2
NI
Ǔ
2
2
) ǒI BNR SǓ ) 4kTR S NG 2 ) (I BIR F) ) 4kTR FNG
(4)
Dividing this expression by the noise gain [NG = (1 +
RF/RG)] gives the equivalent input-referred spot noise
voltage at the noninverting input, as shown in
Equation 5.
EN +
Ǹ
2
2
E2NI ) ǒI BNR SǓ ) 4kTR S )
ǒINGR Ǔ ) 4kTR
NG
BI
F
F
(5)
Evaluating these two equations for the OPA890
circuit and component values (see Figure 46) gives a
total output spot noise voltage of 17.4nV/√Hz and a
total equivalent input spot noise voltage of
8.7nV/√Hz. This total includes the noise added by
the bias current cancellation resistor (175Ω) on the
noninverting input. This total input-referred spot
noise voltage is only slightly higher than the 8nV/√Hz
specification for the op amp voltage noise alone.
This result will be the case, as long as the
impedances appearing at each op amp input are
limited to the previously recommend maximum value
of 350Ω. Keeping both (RF
RG) and the
noninverting input source impedance less than 350Ω
satisfies both noise and frequency response flatness
considerations. Because the resistor-induced noise is
relatively negligible, additional capacitive decoupling
across the bias current cancellation resistor (RB) for
the inverting op amp configuration of Figure 51 is not
required.
22
The balanced input stage of a wideband
voltage-feedback op amp allows good output dc
accuracy in a wide variety of applications. The
power-supply current trim for the OPA890 gives even
tighter control than comparable amplifiers. Although
the high-speed input stage does require relatively
high input bias current (+25°C worst case, 1.6µA at
each input terminal), the close matching between
them may be used to reduce the output dc error
caused by this current. The total output offset voltage
may be considerably reduced by matching the dc
source resistances appearing at the two inputs. This
matching reduces the output dc error resulting from
the input bias currents to the offset current times the
feedback resistor. Evaluating the configuration of
Figure 46, and using worst-case +25°C input offset
voltage and current specifications, gives a
worst-case output offset voltage equal to:
±(NG × VOS(MAX)) ± (RF× IOS(MAX))
= ±(2 × 5mV) ± (750Ω× 0.35µA)
= ±11.3mV
with NG = noninverting signal gain
A fine-scale output offset null or dc operating point
adjustment is often required. Numerous techniques
are available for introducing dc offset control into an
op amp circuit. Most of these techniques eventually
reduce to adding a dc current through the feedback
resistor. In selecting an offset trim method, one key
consideration is the impact on the desired signal
path frequency response. If the signal path is
intended to be noninverting, the offset control is best
applied as an inverting summing signal to avoid
interaction with the signal source. If the signal path is
intended to be inverting, applying the offset control to
the noninverting input may be considered. However,
the dc offset voltage on the summing junction will set
up a dc current back into the source that must be
considered. Applying an offset adjustment to the
inverting op amp input can change the noise gain
and frequency response flatness. For a dc-coupled
inverting amplifier, see Figure 53 for one example of
an offset adjustment technique that has minimal
impact on the signal frequency response. In this
case, the dc offsetting current is brought into the
inverting input node through resistor values that are
much larger than the signal path resistors. This
configuration ensures that the adjustment circuit has
minimal effect on the loop gain and thus, the
frequency response.
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+5V
Power-supply decoupling
not shown.
0.1mF
226W
OPA890
RG
324W
+5V
5kW
-5V
VO
RF
750W
VI
20kW
±150mV Output Adjustment
10kW
VO
0.1mF
VI
5kW
=-
RF
RG
= -2
-5V
Figure 53. DC-Coupled, Inverting Gain of -2V/V,
with Offset Adjustment
collector current out of Q1, turning the amplifier off.
The supply current in the disable mode is only that
required to operate the circuit of Figure 54.
Additional circuitry ensures that turn-on time occurs
faster than turn-off time (make-before-break).
When disabled, the output and input nodes go to a
high-impedance state. If the OPA890 is operating at
a gain of +1V/V, it shows a very high impedance at
the output and exceptional signal isolation. If
operating at a gain greater than +1V/V, the total
feedback network resistance (RF + RG) appears as
the impedance looking back into the output, but the
circuit still shows very-high forward and reverse
isolation. If configured as an inverting amplifier, the
input and output are connected through the feedback
network resistance (RF + RG) and the isolation is
very poor, as a result.
THERMAL ANALYSIS
DISABLE OPERATION
The OPA890 provides an optional disable feature
that may be used either to reduce system power or
to implement a simple channel multiplexing
operation. If the DIS control pin is left unconnected,
the OPA890 operates normally. To disable the
OPA890, the control pin must be asserted low.
Figure 54 shows a simplified internal circuit for the
disable control feature.
+VS
80kW
Operating junction temperature (TJ) is given by TA +
PD × θJA. The total internal power dissipation (PD) is
the sum of quiescent power (PDQ) and additional
power dissipated in the output stage (PDL) to deliver
load power. Quiescent power is simply the specified
no-load supply current times the total supply voltage
across the part. PDL depends on the required output
signal and load, but for a grounded resistive load is
at a maximum when the output is fixed at a voltage
equal to 1/2 of either supply voltage (for equal
bipolar supplies). Under this condition, PDL = VS2/(4 ×
RL) where RL includes feedback network loading.
Note that it is the power in the output stage and not
into the load that determines internal power
dissipation.
Q1
200kW
2MW
VDIS
IS
Control
Maximum desired junction temperature sets the
maximum allowed internal power dissipation, as
described below. In no case should the maximum
junction temperature be allowed to exceed +150°C.
-VS
Figure 54. Simplified Disable Control Circuit
In normal operation, base current to Q1 is provided
through the 2MΩ resistor, while the emitter current
through the 80kΩ resistor sets up a voltage drop that
is inadequate to turn on the two diodes in the Q1
emitter. As VDIS is pulled low, additional current is
pulled through the 80kΩ resistor, eventually turning
on those two diodes (≈15µA). At this point, any
further current pulled out of VDIS goes through those
diodes, holding the emitter-base voltage of Q1 at
approximately 0V. This process shuts off the
As a worst-case example, compute the maximum TJ
using an OPA890IDBV (SOT23-6 package) in the
circuit of Figure 46 operating at the maximum
specified ambient temperature of +85°C and driving
a grounded 100Ω load.
PD = 10V × 1.25mA + 52/(4 × (100Ω
79mW
1.5kΩ)) =
Maximum TJ = +85°C + (79W × 150°C/W) = +97°C.
Although this result is still well below the specified
maximum junction temperature, system reliability
considerations may require lower operating junction
temperatures. The highest possible internal
dissipation occurs if the load requires current to be
forced into the output for positive output voltages, or
sourced from the output for negative output voltages.
This configuration puts a high current through a large
internal voltage drop in the output transistors.
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BOARD LAYOUT GUIDELINES
Achieving
optimum
performance
with
a
high-frequency amplifier such as the OPA890
requires careful attention to board layout parasitics
and external component types. Recommendations
that optimize performance include the following:
a. Minimize parasitic capacitance to any ac
ground for all of the signal I/O pins. Parasitic
capacitance on the output and inverting input
pins can cause instability; on the noninverting
input, it can react with the source impedance
to cause unintentional bandlimiting. To reduce
unwanted capacitance, a window around the
signal I/O pins should be opened in all of the
ground and power planes around those pins.
Otherwise, ground and power planes should
be unbroken elsewhere on the board.
b. Minimize the distance (< 0.25") from the
power-supply pins to high-frequency 0.1µF
decoupling capacitors. At the device pins, the
ground and power-plane layout should not be
in close proximity to the signal I/O pins. Avoid
narrow power and ground traces to minimize
inductance between the pins and the
decoupling capacitors. The power-supply
connections should always be decoupled with
these capacitors. An optional supply
decoupling capacitor (0.1µF) across the two
power supplies (for bipolar operation) will
improve 2nd-harmonic distortion performance.
Larger (2.2µF to 6.8µF) decoupling capacitors,
effective at lower frequencies, should also be
used on the main supply pins. These
capacitors may be placed somewhat farther
from the device and may be shared among
several devices in the same area of the PCB.
c. Careful selection and placement of
external
components
preserves
the
high-frequency
performance
of
the
OPA890. Resistors should be a very low
reactance type. Surface-mount resistors work
best and allow a tighter overall layout. Metal
film or carbon composition axially-leaded
resistors
can
also
provide
good
high-frequency performance. Again, keep the
leads and PCB traces as short as possible.
Never use wirewound type resistors in a
high-frequency application. Because the
output pin and inverting input pin are the most
sensitive to parasitic capacitance, always
position the feedback and series output
resistor, if any, as close as possible to the
output pin. Other network components, such
as noninverting input termination resistors,
should also be placed close to the package.
Where double-side component mounting is
24
allowed, place the feedback resistor directly
under the package on the other side of the
board between the output and inverting input
pins. Even with a low parasitic capacitance
shunting the external resistors, excessively
high resistor values can create significant time
constants that can degrade performance.
Good axial metal film or surface-mount
resistors have approximately 0.2pF in shunt
with the resistor. For resistor values > 1.5kΩ,
this parasitic capacitance can add a pole
and/or zero below 500MHz that can effect
circuit operation. Keep resistor values as low
as possible consistent with load driving
considerations. The 750Ω feedback used in
the Typical Characteristics is a good starting
point for design. Note that a direct short is
suggested for the unity-gain follower
application.
d. Connections to other wideband devices on
the board may be made with short, direct
traces or through onboard transmission lines.
For short connections, consider the trace and
the input to the next device as a lumped
capacitive load. Relatively wide traces (50mils
to 100mils) should be used, preferably with
ground and power planes opened up around
them. Estimate the total capacitive load and
set RS from the plot of Recommended RS vs
Capacitive Load. Low parasitic capacitive
loads (< 5pF) may not need an RS because
the OPA890 is nominally compensated to
operate with a 2pF parasitic load. Higher
parasitic capacitive loads without an RS are
allowed as the signal gain increases
(increasing the unloaded phase margin). If a
long trace is required, and the 6dB signal loss
intrinsic to a doubly-terminated transmission
line is acceptable, implement a matched
impedance transmission line using microstrip
or stripline techniques (consult an ECL design
handbook for microstrip and stripline layout
techniques). A 50Ω environment is normally
not necessary on the board, and in fact, a
higher impedance environment will improve
distortion as shown in the distortion versus
load plots. With a characteristic board trace
impedance defined (based on board material
and trace dimensions), a matching series
resistor into the trace from the output of the
OPA890 is used as well as a terminating
shunt resistor at the input of the destination
device. Remember also that the terminating
impedance is the parallel combination of the
shunt resistor and the input impedance of the
destination device; this total effective
impedance should be set to match the trace
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impedance. The high output voltage and
current capability of the OPA890 allows
multiple destination devices to be handled as
separate transmission lines, each with its
respective series and shunt terminations. If
the 6dB attenuation of a doubly-terminated
transmission line is unacceptable, a long trace
can be series-terminated at the source end
only. Treat the trace as a capacitive load in
this case, and set the series resistor value as
shown in the plot of Recommended RS vs
Capacitive Load. This configuration does not
preserve signal integrity as well as a
doubly-terminated line. If the input impedance
of the destination device is low, there will be
some signal attenuation because of the
voltage divider formed by the series output
into the terminating impedance.
e. Socketing a high-speed part such as the
OPA890 is not recommended. The
additional
lead
length and
pin-to-pin
capacitance introduced by the socket can
create an extremely troublesome parasitic
network that can make it almost impossible to
achieve a smooth, stable frequency response.
Best results are obtained by soldering the
OPA890 directly onto the board.
INPUT AND ESD PROTECTION
The OPA890 is built using a very high-speed,
complementary, bipolar process. The internal
junction breakdown voltages are relatively low for
these very small geometry devices. These
breakdowns are reflected in the Absolute Maximum
Ratings table. All device pins are protected with
internal ESD protection diodes to the power supplies,
as shown in Figure 55.
+VCC
External
Pin
Internal
Circuitry
-VCC
Figure 55. Internal ESD Protection
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA
continuous current. Where higher currents are
possible (for example, in systems with ±15V supply
parts driving into the OPA890), current-limiting series
resistors should be added into the two inputs. Keep
these resistor values as low as possible, because
high values degrade both noise performance and
frequency response.
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25
PACKAGE OPTION ADDENDUM
www.ti.com
8-Jun-2007
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
OPA890ID
ACTIVE
SOIC
D
8
OPA890IDBVR
ACTIVE
SOT-23
DBV
OPA890IDBVT
ACTIVE
SOT-23
OPA890IDR
ACTIVE
SOIC
75
Lead/Ball Finish
MSL Peak Temp (3)
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
6
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
DBV
6
250
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
D
8
2500 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Jun-2007
TAPE AND REEL INFORMATION
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
Device
6-Jun-2007
Package Pins
Site
Reel
Diameter
(mm)
Reel
Width
(mm)
A0 (mm)
B0 (mm)
K0 (mm)
P1
(mm)
W
Pin1
(mm) Quadrant
OPA890IDBVR
DBV
6
MLA
180
8
6.83
7.42
1.88
8
12
Q3
OPA890IDBVT
DBV
6
MLA
180
8
6.83
7.42
1.88
8
12
Q3
OPA890IDR
D
8
MLA
330
12
6.9
5.4
2.0
8
12
Q1
TAPE AND REEL BOX INFORMATION
Device
Package
Pins
Site
Length (mm)
Width (mm)
OPA890IDBVR
DBV
6
MLA
0.0
0.0
0.0
OPA890IDBVT
DBV
6
MLA
190.0
212.7
31.75
OPA890IDR
D
8
MLA
342.9
336.6
28.58
Pack Materials-Page 2
Height (mm)
PACKAGE MATERIALS INFORMATION
www.ti.com
6-Jun-2007
Pack Materials-Page 3
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