TM MP2106 1.5A, 15V, 800KHz Synchronous Buck Converter The Future of Analog IC Technology TM DESCRIPTION FEATURES The MP2106 is a 1.5A, 800KHz synchronous buck converter designed for low voltage applications requiring high efficiency. It is capable of providing output voltages as low as 0.9V, and integrates top and bottom switches to minimize power loss and component count. The 800KHz switching frequency reduces the size of filtering components, further reducing the solution size. • • • • • • • • • • • The MP2106 includes cycle-by-cycle current limiting and under voltage lockout. The internal power switches, combined with the tiny 10-pin MSOP and QFN packages, provide a solution requiring a minimum of surface area. EVALUATION BOARD REFERENCE Board Number Dimensions EV2106DQ/DK-00A 2.5”X x 2.0”Y x 0.5”Z 1.5A Output Current Synchronous Rectification Internal 210mΩ and 255mΩ Power Switches Input Range of 2.6V to 15V >90% Efficiency Zero Current Shutdown Mode Under Voltage Lockout Protection Soft-Start Operation Thermal Shutdown Internal Current Limit (Source & Sink) Tiny 10-Pin MSOP or QFN Package APPLICATIONS • • • • • • DC/DC Regulation from Wall Adapters Portable Entertainment Systems Set Top Boxes Digital Video Cameras, DECT Networking Equipment Wireless Modems “MPS” and “The Future of Analog IC Technology” are Trademarks of Monolithic Power Systems, Inc. TYPICAL APPLICATION Efficiency vs. Load Current C7 R4 5 OFF ON 1 3 C5 10nF C3 3.3nF R3 100 10nF C1 7 6 VIN RUN BST LX MP2106 SS COMP VREF 4 FB SGND PGND 10 C6 10nF 9 90 VIN=3.3V 80 L1 OUTPUT 1.8V / 1.5A 8 2 R2 C2 EFFICIENCY (%) INPUT 2.6V to 15V 70 60 VIN=5V 50 40 30 20 R1 10 0 0.01 MP2106_TAC_S01 0.1 1 LOAD CURRENT (A) 10 MP2106_TAC_EC02 MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 1 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER PACKAGE REFERENCE TOP VIEW TOP VIEW SS 1 10 SGND FB 2 9 PGND LX COMP 3 8 LX VIN VREF 4 7 VIN RUN 5 6 BST SS 1 10 SGND FB 2 9 PGND COMP 3 8 VREF 4 7 RUN 5 6 BST MP2106_PD01-MSOP10 EXPOSED PAD ON BACKSIDE MP2106_PD02-QFN10 Part Number* Package Temperature Part Number** Package Temperature MP2106DK MSOP10 –40°C to +85°C MP2106DQ QFN10 (3mm x 3mm) –40°C to +85°C * For Tape & Reel, add suffix –Z (eg. MP2106DK–Z) For Lead Free, add suffix –LF (eg. MP2106DK–LF–Z) ** For Tape & Reel, add suffix –Z (eg. MP2106DQ–Z) For Lead Free, add suffix –LF (eg. MP2106DQ–LF–Z) ABSOLUTE MAXIMUM RATINGS (1) Thermal Resistance Input Supply Voltage VIN .............................. 16V LX Voltage VLX ..................... –0.3V to VIN + 0.3V BST to LX Voltage ......................... –0.3V to +6V Voltage on All Other Pins............... –0.3V to +6V Storage Temperature............... –55°C to +150°C MSOP10 ................................ 150 ..... 65... °C/W QFN10 .................................... 50 ...... 12... °C/W Recommended Operating Conditions (2) (3) θJA θJC Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1” square of 1 oz copper. Input Supply Voltage VIN ..................2.6V to 15V Output Voltage VOUT ........................0.9V to 5.5V Operating Temperature.............. –40°C to +85°C ELECTRICAL CHARACTERISTICS VIN = 5.0V, TA = +25°C, unless otherwise noted. Parameter Input Voltage Range Input Under Voltage Lockout Input Under Voltage Lockout Hysteresis Shutdown Supply Current Operating Supply Current VREF Voltage RUN Input Low Voltage RUN Input High Voltage RUN Hysteresis RUN Input Bias Current Oscillator Switching Frequency Maximum Duty Cycle Minimum On Time MP2106 Rev. 1.6 2/22/2006 Symbol Condition VIN VREF VIL VHL Min 2.6 VRUN ≤ 0.3V VRUN > 2V, VFB = 1.1V VIN = 2.6V to 15V Typ 2.2 Units V V 100 mV 0.5 1.2 2.4 Max 15 1.0 1.8 0.4 1.5 100 1 fSW DMAX tON VFB = 0.7V 700 85 800 200 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 900 µA mA V V V mV µA KHz % ns 2 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER ELECTRICAL CHARACTERISTICS (continued) VIN = 5.0V, TA = +25°C, unless otherwise noted. Parameter Error Amplifier Voltage Gain Transconductance COMP Maximum Output Current FB Regulation Voltage FB Input Bias Current Soft-Start Soft-Start Current Output Switch On-Resistance Switch On Resistance Synchronous Rectifier On Resistance Switch Current Limit (Source) Synchronous Rectifier Current Limit (Sink) Thermal Shutdown Symbol Condition Min AVEA GEA VFB IFB 875 VFB = 0.895V ISS VIN = 5V VIN = 3V VIN = 5V VIN = 3V Typ 400 300 ±30 895 –100 Max 915 Units V/V µA/V µA mV nA 2 µA 255 315 210 255 2.5 mΩ mΩ mΩ mΩ A 350 mA 160 °C PIN FUNCTIONS Pin # 1 2 3 4 5 6 7 8 9 10 Name Description Soft-Start Input. Place a capacitor from SS to SGND to set the soft-start period. The MP2106 SS sources 2µA from SS to the soft-start capacitor at startup. As the SS voltage rises, the feedback threshold voltage increases to limit inrush current during startup. Feedback Input. FB is the inverting input of the internal error amplifier. Connect a resistive FB voltage divider from the output voltage to FB to set the output voltage value. Compensation Node. COMP is the output of the error amplifier. Connect a series RC network COMP to compensate the regulation control loop. Internal 2.4V Regulator Bypass. Connect a 10nF capacitor between VREF and SGND to VREF bypass the internal regulator. Do not apply any load to VREF. On/Off Control Input. Drive RUN high to turn on the MP2106; low to turn it off. For automatic RUN startup, connect RUN to VIN via a pullup resistor. Power Switch Boost. BST powers the gate of the high-side N-Channel power MOSFET switch. BST Connect a 10nF or greater capacitor between BST and LX. Internal Power Input. VIN supplies the power to the MP2106 through the internal LDO VIN regulator. Bypass VIN to PGND with a 10µF or greater capacitor. Connect VIN to the input source voltage. Output Switching Node. LX is the source of the high-side N-Channel switch and the drain of the LX low-side N-Channel switch. Connect the output LC filter between LX and the output. Power Ground. PGND is the source of the N-Channel MOSFET synchronous rectifier. Connect PGND PGND to SGND as close to the MP2106 as possible. SGND Signal Ground. MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 3 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS Circuit of Figure 2, VIN = 5V, VOUT = 1.8V, L1 = 5µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless otherwise noted. Steady State Operation Steady State Operation 1.5A Load No Load VSW 5V/div. Load Transient VSW 5V/div. VO AC Coupled 10mV/div. VIN AC Coupled 200mV/div. VOUT AC Coupled 200mV/div. VO AC Coupled 10mV/div. IL 1A/div. VIN AC Coupled 20mV/div. ILOAD 1A/div. IL 1A/div. IL 1A/div. MP2106-TPC01 MP2106-TPC02 MP2106-TPC03 Startup from Shutdown Startup from Shutdown 1.5A Resistive Load No Load VEN 2V/div. VEN 2V/div. VOUT 1V/div. VOUT 1V/div. IL 1A/div. IL 1A/div. VSW 5V/div. VSW 5V/div. 1ms/div. 1ms/div. MP2106-TPC04 MP2106-TPC05 Short Circuit Protection Short Circuit Recovery VOUT 1V/div. VOUT 1V/div. IL 1A/div. IL 1A/div. MP2106-TPC06 MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. MP2106-TPC07 4 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER OPERATION VIN OFF ON RUN 5 VREF 4 C6 ENABLE CKT & LDO REGULATOR VBP 2.4V GATE DRIVE REGULATOR Vdr CURRENT SENSE AMPLIFIER C1 7 VIN 2.6V to 15V + -BST Vdr PWM COMPARATOR 6 + -- C7 LX CONTROL LOGIC Vdr L1 VOUT 8 C2 800KHz OSCILLATOR RAMP VBP CURRENT LIMIT COMPARATOR + -- UVLO & THERMAL SHUTDOWN R2 + -- PGND SS 9 C5 1 -FB GM -ERROR AMPLIFIER VFB 0.895V CURRENT LIMIT THRESHOLD 10 R1 3 SGND C4 2 + COMP R3 C3 MP2106_BD01 Figure 1—Functional Block Diagram MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 5 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER switch, forcing the inductor current to decrease. The average inductor current is controlled by the voltage at COMP, which in turn is controlled by the output voltage. Thus the output voltage controls the inductor current to satisfy the load. The MP2106 measures the output voltage through an external resistive voltage divider and compares that voltage to the internal 0.9V reference in order to generate the error voltage at COMP. The current-mode regulator uses the voltage at COMP and compares it to the inductor current to regulate the output voltage. The use of current-mode regulation improves transient response and improves control loop stability. Since the high-side N-Channel MOSFET requires voltages above VIN to drive its gate, a bootstrap capacitor from LX to BST is required to drive the high-side MOSFET gate. When LX is driven low (through the low-side MOSFET), the BST capacitor is internally charged. The voltage at BST is applied to the high-side MOSFET gate to turn it on, and maintains that voltage until the high-side MOSFET is turned off and the low-side MOSFET is turned on, and the cycle repeats. Connect a 10nF or greater capacitor from BST to SW to drive the high-side MOSFET gate. At the beginning of each cycle, the high-side N-Channel MOSFET is turned on, forcing the inductor current to rise. The current at the drain of the high-side MOSFET is internally measured and converted to a voltage by the current sense amplifier. That voltage is compared to the error voltage at COMP. When the inductor current rises sufficiently, the PWM comparator turns off the high-side switch and turns on the low-side APPLICATION INFORMATION C7 10nF INPUT 2.6V to 15V 5 1 3 C5 10nF C4 OPEN C3 3.3nF 7 6 VIN BST RUN LX MP2106 SS COMP VREF C6 10nF 4 FB 8 OUTPUT 1.8V / 1.5A 2 SGND PGND 10 9 MP2106_TAC_F02 Figure 2—Typical Application Circuit Internal Low-Dropout Regulator The internal power to the MP2106 is supplied from the input voltage (VIN) through an internal 2.4V low-dropout linear regulator, whose output is VREF. Bypass VREF to SGND with a 10nF or greater capacitor for proper operation. The internal regulator can not supply more current than is required to operate the MP2106. Therefore, do not apply any external load to VREF. MP2106 Rev. 1.6 2/22/2006 Soft-Start The MP2106 includes a soft-start timer that slowly ramps the output voltage at startup to prevent excessive current at the input. When power is applied to the MP2106, and RUN is asserted, a 2µA internal current source charges the external capacitor at SS. As the capacitor charges, the voltage at SS rises. The MP2106 internally limits the feedback threshold voltage at FB to that of the voltage at SS. This forces the output voltage to rise at the same rate as the voltage at SS, forcing the output www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 6 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER voltage to ramp linearly from 0V to the desired regulation voltage during soft-start. The soft-start period is determined by the equation: t SS = 0.45 × C5 Where C5 (in nF) is the soft-start capacitor from SS to GND, and tSS (in ms) is the soft-start period. Determine the capacitor required for a given soft-start period by the equation: C5 = 2.22 × t SS Use values between 10nF and 22nF for C5 to set the soft-start period (between 4ms and 10ms). Setting the Output Voltage (see Figure 2) Set the output voltage by selecting the resistive voltage divider ratio. The voltage divider drops the output voltage to the 0.895V feedback voltage. Use 10kΩ for the low-side resistor of the voltage divider. Determine the high-side resistor by the equation: ⎞ ⎛ V R2 = ⎜⎜ OUT − 1⎟⎟ × R1 ⎠ ⎝ 0.895 V Where R2 is the high-side resistor, VOUT is the output voltage and R1 is the low-side resistor. Selecting the Input Capacitor The input current to the step-down converter is discontinuous, and so a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. A low ESR capacitor is required to keep the noise at the IC to a minimum. Ceramic capacitors are preferred, but tantalum or low ESR electrolytic capacitors may also suffice. The capacitor can be electrolytic, tantalum or ceramic. Because it absorbs the input switching current it must have an adequate ripple current rating. Use a capacitor with RMS current rating greater than 1/2 of the DC load current. For stable operation, place the input capacitor as close to the IC as possible. A smaller high quality 0.1µF ceramic capacitor may be placed closer to the IC with the larger capacitor placed further away. If using this technique, it is recommended that the larger capacitor be a tantalum or electrolytic type. All ceramic MP2106 Rev. 1.6 2/22/2006 capacitors should be placed close to the MP2106. For most applications, a 10µF ceramic capacitor will work. Selecting the Output Capacitor The output capacitor (C2) is required to maintain the DC output voltage. Low ESR capacitors are preferred to keep the output voltage ripple to a minimum. The characteristics of the output capacitor also affect the stability of the regulation control system. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. The output voltage ripple is: VRIPPLE = ⎛ VOUT V × ⎜⎜1 − OUT f SW × L ⎝ VIN ⎞ ⎞ ⎛ 1 ⎟ ⎟⎟ × ⎜ R ESR + ⎜ 8 × f SW × C2 ⎟⎠ ⎠ ⎝ Where VRIPPLE is the output voltage ripple, fSW is the switching frequency, VIN is the input voltage, RESR is the equivalent series resistance of the output capacitors and fSW is the switching frequency. Choose an output capacitor to satisfy the output ripple requirements of the design. A 22µF ceramic capacitor is suitable for most applications. Selecting the Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor results in less ripple current that will result in lower output ripple voltage. However, the larger value inductor is likely to have a larger physical size and higher series resistance. Choose an inductor that does not saturate under the worst-case load conditions. A good rule for determining the inductance is to allow the peak-to-peak ripple current to be approximately 30% to 40% of the maximum load current. Make sure that the peak inductor current (the load current plus half the peak-topeak inductor ripple current) is below 2.5A to prevent loss of regulation due to the current limit. www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 7 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER Calculate the required inductance value by the equation: L= VOUT × (VIN − VOUT ) VIN × f SW × ∆I Where ∆I is the peak-to-peak inductor ripple current. It is recommended to choose ∆I to be 30%~40% of the maximum load current. Compensation The system stability is controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC loop gain is: ⎛ V A VDC = ⎜⎜ FB ⎝ VOUT ⎞ ⎟ × A VEA × G CS × R LOAD ⎟ ⎠ Where VFB is the feedback voltage, AVEA is the transconductance error amplifier voltage gain, GCS is the current sense transconductance (roughly the output current divided by the voltage at COMP) and RLOAD is the load resistance: R LOAD V = OUT I OUT Where IOUT is the output load current. The system has 2 poles of importance, one is due to the compensation capacitor (C3), and the other is due to the load resistance and the output capacitor (C2), where: fP1 = GEA 2π × A VEA × C3 P1 is the first pole, and GEA is the error amplifier transconductance (300µA/V) and fP 2 = MP2106 Rev. 1.6 2/22/2006 1 2π × R LOAD × C2 The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). The zero is: f Z1 = 1 2π × R3 × C3 If large value capacitors with relatively high equivalent-series-resistance (ESR) are used, the zero due to the capacitance and ESR of the output capacitor can be compensated by a third pole set by R3 and C4. The pole is: fP3 = 1 2π × R3 × C4 The system crossover frequency (the frequency where the loop gain drops to 1, or 0dB, is important. Set the crossover frequency to below one tenth of the switching frequency to insure stable operation. Lower crossover frequencies result in slower response and worse transient load recovery. Higher crossover frequencies degrade the phase and/or gain margins and can result in instability. Table 1—Compensation Values for Typical Output Voltage/Capacitor Combinations VOUT C2 1.8V 22µF Ceramic 2.5V 22µF Ceramic 3.3V 22µF Ceramic 47µF Tantalum 1.8V (300mΩ) 47µF Tantalum 2.5V (300mΩ) 47µF Tantalum 3.3V (300mΩ) R3 C3 C4 6.8kΩ 9.1kΩ 12kΩ 3.3nF 2.2nF 1.8nF None None None 13kΩ 2nF 1nF 18kΩ 1.2nF 750pF 24kΩ 1nF 560pF Choosing the Compensation Components The values of the compensation components given in Table 1 yield a stable control loop for the given output voltage and capacitor. To optimize the compensation components for conditions not listed in Table 1, use the following procedure. www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 8 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER Choose the compensation resistor to set the desired crossover frequency. Determine the value by the following equation: R3 = 2π × C2 × VOUT × f C G EA × G CS × VFB External Boost Diode For input voltages less than or equal to 5V, it is recommended that an external boost diode be added. This will help improve the regulator efficiency. The diode can be a low cost diode such as an IN4148 or BAT54. Where fC is the desired crossover frequency (preferably 33KHz). Choose the compensation capacitor to set the zero below one fourth of the crossover frequency. Determine the value by the following equation: 2 C3 > π × R3 × f C Determine if the second compensation capacitor, C4 is required. It is required if the ESR zero of the output capacitor happens at less than half of the switching frequency. Or: π × C2 × R ESR × f SW > 1 If this is the case, then add the second compensation capacitor. 5V BST 6 10nF MP2106 LX BOOST DIODE 8 MP2106_F03 Figure 3—External Boost Diode This diode is also recommended for high duty cycle operation (when VOUT >65%) and high VIN applications. output voltage (VOUT>12V) However, do not exceed the absolute maximum voltage for these pins. Determine the value by the equation: C4 = C2 × R ESR(max) R3 Where RESR(MAX) is the maximum ESR of the output capacitor. MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 9 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER PACKAGE INFORMATION MSOP10 MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 10 TM MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER QFN10 (3mm x 3mm) NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications. MP2106 Rev. 1.6 2/22/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 11