LM4961 Ceramic Speaker Driver General Description Key Specifications The LM4961 is an audio power amplifier primarily designed for driving Ceramic Speaker for applications in Cell Phone and PDAs. It integrates a boost converter, with variable output voltage, with an audio power amplifier. It is capable of driving 15Vp-p in BTL mode to 2uF+ 30 ohms load, continuous average power, with less than 1% distortion (THD+N) from a 3.2VDC power supply. n Quiescent Power Supply Current n Voltage Swing in BTL at 1% THD n Shutdown current Boomer audio power amplifiers were designed specifically to provide high quality output power with a minimal number of external components. The LM4961 does not require bootstrap capacitors, or snubber circuits therefore it is ideally suited for portable applications requiring high voltage output to drive capacitive loads like Ceramic Speakers. The LM4961 features a low-power consumption shutdown mode. Additionally, the LM4961 features an internal thermal shutdown protection mechanism. The LM4961 contains advanced pop & click circuitry that eliminates noises which would otherwise occur during turn-on and turn-off transitions. The LM4961 is unity-gain stable and can be configured by external gain-setting resistors. 7mA (typ) 15Vp-p (typ) 0.1µA (typ) Features n Pop & click circuitry eliminates noise during turn-on and turn-off transitions n Low current shutdown mode n Low quiescent current n Mono 15Vp-p BTL output, RL = 2µF+30Ω, f = 1kHz n Thermal shutdown protection n Unity-gain stable n External gain configuration capability n Including Band exchange SW n Including Leakage cut SW Applications n Cellphone n PDA Connection Diagram LM4961LQ (5x5) 20094084 Top View Order Number LM4XXX See NS Package Number Boomer ® is a registered trademark of National Semiconductor Corporation. © 2004 National Semiconductor Corporation DS200940 www.national.com LM4961 Ceramic Speaker Driver October 2004 LM4961 Typical Application 20094083 * RC is needed for over/under voltage protection. If inputs are less than VDD +0.3V and greater than –0.3V, and if inputs are disabled when in shutdown mode, then RC can be shorted. FIGURE 1. Typical Audio Amplifier Application Circuit Shutdown 1 Shutdown 2 Band-SW Receiver Mode — high low Ringer Mode high high high Shutdown low low — www.national.com 2 Junction Temperature If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Thermal Resistance Supply Voltage (Vdd) 150˚C θJA (LLP) 66˚C/W See AN-1187 ’Leadless Leadframe Packaging (LLP).’ 6.0V Amplifier Supply Voltage (V1) 9.5V Storage Temperature Operating Ratings −65˚C to +150˚C Input Voltage Temperature Range −0.3V to VDD + 0.3V Power Dissipation (Note 3) Internally limited ESD Susceptibility (Note 4) 2000V ESD Susceptibility (Note 5) 200V TMIN ≤ TA ≤ TMAX −40˚C ≤ TA ≤ +85˚C Supply Voltage (VDD) 3.0V < VDD < 5.0V 2.7V < V1 < 9.0V Amplifier Supply Voltage (V1) Electrical Characteristics VDD = 4.2V The following specifications apply for VDD = 4.2V, AV-BTL = 26dB, RL = 2µF+30Ω, Cb = 1.0µF, Band-SW = VDD unless otherwise specified. Limits apply for TA = 25˚C. Symbol Parameter Conditions LM4961 Typical (Note 6) Limit (Notes 7, 8) Units (Limits) IDD Quiescent Power Supply Current VIN = 0V, No Load Band-SW = VDD 7 14 mA (max) Iddrcv Iq in receiver mode VIN = 0V, No Load Band-SW = GND 2 4 mA (max) ISD Shutdown Current VSHUTDOWN1 = VSHUTDOWN2 = GND Band-SW = GND (Note 9) 0.1 2.0 µA (max) VLH Logic High Threshold Voltage For Shutdown 1, Shutdown 2, and Band-SW 1.5 V (min) VLL Logic Low Threshold Voltage For Shutdown 1, Shutdown 2, and Band-SW 0.4 V (max) RPULLDOWN Pulldown Resistor For Shutdown 2 and Band-SW TSD Thermal Shutdown Temperature Vout Output Voltage Swing THD = 1%, f = 1kHz RL = 2µF+30Ω Mono BTL THD+N Total Harmomic Distortion + Noise eOS 50k Ω (min) 125 ˚C (min) 15 14 Vp-p (min) Vout = 14Vp-p, f = 1kHz 0.05 1.0 % (max) Output Noise A-Weighted Filter, VIN = 0V (Note 10) 115 PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 100Hz 80 65 dB (min) Ron-sw-out On Resistance on SW-Out Band SW “High” Isink = 100µA (Between pin 1 and pin 28) 170 220 Ω (max) 70k µV Electrical Characteristics VDD = 3.2V The following specifications apply for VDD = 3.2V, AV-BTL = 26dB, RL = 2µF+30Ω, Cb = 1.0µF, Band-SW = VDD unless otherwise specified. Limits apply for TA = 25˚C. Symbol Parameter Conditions LM4961 Typical (Note 6) Limit (Notes 7, 8) Units (Limits) IDD Quiescent Power Supply Current VIN = 0V, No Load Band-SW = VDD 9 15 mA (max) Iddrcv Iq in receiver mode VIN = 0V, No Load Band-SW = GND 2 4 mA (max) ISD Shutdown Current VSHUTDOWN1 = VSHUTDOWN2 = GND Band-SW = GND (Note 9) 0.1 2.0 µA (max) 3 www.national.com LM4961 Absolute Maximum Ratings (Notes 1, 2) LM4961 Electrical Characteristics VDD = 3.2V (Continued) The following specifications apply for VDD = 3.2V, AV-BTL = 26dB, RL = 2µF+30Ω, Cb = 1.0µF, Band-SW = VDD unless otherwise specified. Limits apply for TA = 25˚C. Symbol Parameter Conditions LM4961 Typical (Note 6) Limit (Notes 7, 8) Units (Limits) VLH Logic High Threshold Voltage For Shutdown 1, Shutdown 2, and Band-SW 1.5 V (min) VLL Logic Low Threshold Voltage For Shutdown 1, Shutdown 2, and Band-SW 0.4 V (max) RPULLDOWN Pulldown Resistor For Shutdown 2 and Band-SW 50k Ω (min) TSD Thermal Shutdown Temperature 125 ˚C (min) 15 14 Vp-p (min) 1.0 % (max) 70k Vout Output Voltage Swing THD = 1%, f = 1kHz RL = 2µF+30Ω Mono BTL THD+N Total Harmomic Distortion + Noise Vout = 14Vp-p, f = 1kHz 0.1 eOS Output Noise A-Weighted Filter, VIN = 0V (Note 10) 125 PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 100Hz 80 65 dB (min) Ron-sw-out On Resistance on SW-Out Band SW “High” Isink = 100µA (Between pin 1 and pin 28) 170 220 Ω (max) µV Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified. Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good indication of device performance. Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower. For the LM4961 typical application (shown in Figure 1) with VDD = 4.2V, RL = 2µF+30Ω mono BTL operation the maximum power dissipation is 232mW. θJA = 66˚C/W. Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor. Note 5: Machine Model, 220pF–240pF discharged through all pins. Note 6: Typicals are measured at 25˚C and represent the parametric norm. Note 7: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level). Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. Note 9: Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to GND for minimum shutdown current. Note 10: Noise measurements are dependent on the absolute values of closed loop gain setting resistors (input and feedback resistors). www.national.com 4 THD+N vs Frequency VDD = 3.2V, VO = 14VP-P, RL = 2µF+30Ω THD+N vs Frequency VDD = 4.2V, VO = 14VP-P, RL = 2µF+30Ω 20094087 200940C3 THD+N vs Output Voltage VDD = 3.2V, RL = 2µF + 30Ω THD+N vs Output Voltage VDD = 4.2V, RL = 2µF + 30Ω 200940C4 200940C5 PSRR vs Frequency VDD = 4.2V, RL = 8Ω, VRIPPLE = 200mVP-P PSRR vs Frequency VDD = 3.2V, RL = 8Ω, VRIPPLE = 20mVP-P 20094089 20094090 5 www.national.com LM4961 Typical Performance Characteristics LM4961 Typical Performance Characteristics (Continued) Power Dissipation vs Output Power VDD = 4.2V, RL = 2µF + 30Ω, f = 1kHz Power Dissipation vs Output Power VDD = 3.3V, RL = 2µF + 30Ω, f = 1kHz 20094091 20094092 Frequency Response vs Input Capacitor Size RL = 8Ω Supply Current vs Supply Voltage RL = 2µF + 30Ω, VIN = 0V, RSOURCE = 50Ω 20094093 20094094 Switch Current Limit vs Duty Cycle Oscillator Frequency vs Temperature 20094096 20094095 www.national.com 6 LM4961 Typical Performance Characteristics (Continued) Feedback Voltage vs Temperature Feedback Bias Current vs Temperature 20094097 20094098 Max. Duty Cycle vs Temperature - ”X” RDS (ON) vs Temperature 200940A1 200940A0 RDS (ON) vs VDD 200940A2 7 www.national.com LM4961 Application Information where DC is the duty cycle. BRIDGE CONFIGURATION EXPLANATION There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation. The Audio Amplifier portion of the LM4961 has two internal amplifiers allowing different amplifier configurations. The first amplifier’s gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain, inverting configuration. The closed-loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the second amplifier’s gain is fixed by the two internal 20kΩ resistors. Figure 1 shows that the output of amplifier one serves as the input to amplifier two. This results in both amplifiers producing signals identical in magnitude, but out of phase by 180˚. Consequently, the differential gain for the Audio Amplifier is TOTAL POWER DISSIPATION The total power dissipation for the LM4961 can be calculated by adding Equation 1 and Equation 2 together to establish Equation 3: PDMAX(TOTAL) = [4*(VDD)2/ 2π ZL]+[DCxIIND(AVE)2xRDS(ON)] 2 (3) The result from Equation 3 must not be greater than the power dissipation that results from Equation 4: AVD = 2 *(Rf/Ri) PDMAX = (TJMAX - TA) / θJA By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as “bridged mode” is established. Bridged mode operation is different from the classic single-ended amplifier configuration where one side of the load is connected to ground. A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides differential drive to the load, thus doubling the output swing for a specified supply voltage. Four times the output power is possible as compared to a single-ended amplifier under the same conditions. The bridge configuration also creates a second advantage over single-ended amplifiers. Since the differential outputs, Vo1 and Vo2, are biased at half-supply, no net DC voltage exists across the load. This eliminates the need for an output coupling capacitor which is required in a single supply, single-ended amplifier configuration. Without an output coupling capacitor, the half-supply bias across the load would result in both increased internal IC power dissipation and also possible loudspeaker damage. For the LQA28A, θJA = 66˚C/W. TJMAX = 125˚C for the LM4961. Depending on the ambient temperature, TA, of the system surroundings, Equation 4 can be used to find the maximum internal power dissipation supported by the IC packaging. If the result of Equation 3 is greater than that of Equation 4, then either the supply voltage must be increased, the load impedance increased or TA reduced. For the typical application of a 4.2V power supply, with a 2uF+30Ω load, the maximum ambient temperature possible without violating the maximum junction temperature is approximately 109˚C provided that device operation is around the maximum power dissipation point. Thus, for typical applications, power dissipation is not an issue. Power dissipation is a function of output power and thus, if typical operation is not around the maximum power dissipation point, the ambient temperature may be increased accordingly. Refer to the Typical Performance Characteristics curves for power dissipation information for lower output levels. EXPOSED-DAP PACKAGE PCB MOUNTING CONSIDERATIONS The LM4961’s exposed-DAP (die attach paddle) package (LD) provides a low thermal resistance between the die and the PCB to which the part is mounted and soldered. The low thermal resistance allows rapid heat transfer from the die to the surrounding PCB copper traces, ground plane, and surrounding air. The LD package should have its DAP soldered to a copper pad on the PCB. The DAP’s PCB copper pad may be connected to a large plane of continuous unbroken copper. This plane forms a thermal mass, heat sink, and radiation area. Further detailed and specific information concerning PCB layout, fabrication, and mounting an LD (LLP) package is found in National Semiconductor’s Package Engineering Group under application note AN1187. AMPLIFIER POWER DISSIPATION Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or single-ended. A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation. Since the amplifier portion of the LM4961 has two operational amplifiers, the maximum internal power dissipation is 4 times that of a single-ended amplifier. The maximum power dissipation for a given BTL application can be derived from Equation 1. (1) PDMAX(AMP) = 4(VDD)2 / (2π2ZL) where ZL = Ro1 + Ro2 +1/2πfc SHUTDOWN FUNCTION In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to provide a quick, smooth transition into shutdown. Another solution is to use a single-pole, single-throw switch connected between VDD and Shutdown pins. BOOST CONVERTER POWER DISSIPATION At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be determined by power dissipation within the LM4961 FET switch. The switch power dissipation from ON-time conduction is calculated by Equation 2. (2) PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON) www.national.com (4) BAND SWITCH FUNCTION The LM4961 features a Band Switch function which allows the user to use one amplifier for both receiver (earpiece) mode and ringer/loudspeaker mode. When a logic high 8 REDUCING TRANSIENT CURRENT SPIKE (Continued) Due to the quick turn-on time of the Boost Converter, a transient supply current spike is observed on shutdown release. To reduce the rise time of the output voltage (V1), thus reducing the value of the supply current spike, please refer to application circuit in Figure 2. Using this configuration will allow the user to reduce the transient supply current spike without the Boost Converter experiencing any stability issues. (VDD) is applied to the Band-SW pin (pin 19) the amplifier is in ringer mode. This enables the boost converter and sets the externally configurable closed loop gain selection to BW1. If the Band-SW pin has a logic low (GND) applied to its terminal then the device is in receiver mode. In this mode the boost converter is disabled and the gain selection is switched to BW2. This allows the amplifier to be powered directly from the battery minus the voltage drop across the Schottky diode. 200940A3 FIGURE 2. Transient Current Spike Reduction Configuration PROPER SELECTION OF EXTERNAL COMPONENTS Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC converters, is critical for optimizing device and system performance. Consideration to component values must be used to maximize overall system quality. The best capacitors for use with the switching converter portion of the LM4961 are multi-layer ceramic capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency, which makes them optimum for high frequency switching converters. When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as Z5U 9 www.national.com LM4961 Application Information LM4961 Application Information This feedback capacitor creates a low pass filter that eliminates possible high frequency noise. Care should be taken when calculating the -3dB frequency because an incorrect combination of Rf and Cf2 will cause rolloff before the desired frequency (Continued) and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from Taiyo-Yuden. SELECTING OUTPUT CAPACITOR (CO) FOR BOOST CONVERTER A single 4.7µF to 10µF ceramic capacitor will provide sufficient output capacitance for most applications. If larger amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500 kHz because of significant ringing and temperature rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability. In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce ringing, switching losses, and output voltage ripple. POWER SUPPLY BYPASSING As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply rejection. The capacitor location on both V1 and VDD pins should be as close to the device as possible. SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is dictated by the choice of external components shown in Figure 1. The input coupling capacitor, Ci, forms a first order high pass filter which limits low frequency response. This value should be chosen based on needed frequency response for a few distinct reasons. High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value capacitor is needed to couple in low frequencies without severe attenuation. But ceramic speakers used in portable systems, whether internal or external, have little ability to reproduce signals below 100Hz to 150Hz. Thus, using a high value input capacitor may not increase actual system performance. In addition to system cost and size, click and pop performance is affected by the value of the input coupling capacitor, Ci. A high value input coupling capacitor requires more charge to reach its quiescent DC voltage (nominally 1/2 VDD). This charge comes from the output via the feedback and is apt to create pops upon device enable. Thus, by minimizing the capacitor value based on desired low frequency response, turn-on pops can be minimized. SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nominal value of 4.7µF, but larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other circuitry. SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER The output voltage is set using the external resistors R1 and R2 (see Figure 1). A value of approximately 13.3kΩ is recommended for R2 to establish a divider current of approximately 92µA. R1 is calculated using the formula: SELECTING BYPASS CAPACITOR FOR AUDIO AMPLIFIER Besides minimizing the input capacitor value, careful consideration should be paid to the bypass capacitor value. Bypass capacitor, CB, is the most critical component to minimize turn-on pops since it determines how fast the amplifer turns on. The slower the amplifier’s outputs ramp to their quiescent DC voltage (nominally 1/2 VDD), the smaller the turn-on pop. Choosing CB equal to 1.0µF along with a small value of Ci (in the range of 0.039µF to 0.39µF), should produce a virtually clickless and popless shutdown function. Although the device will function properly, (no oscillations or motorboating), with CB equal to 0.1µF, the device will be much more susceptible to turn-on clicks and pops. Thus, a value of CB equal to 1.0µF is recommended in all but the most cost sensitive designs. R1 = R2 X (V2/1.23 − 1) FEED-FORWARD COMPENSATION FOR BOOST CONVERTER Although the LM4961’s internal Boost converter is internally compensated, the external feed-forward capacitor Cf is required for stability (see Figure 1). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz should be approximately 6kHz. Cf1 can be calculated using the formula: Cf1 = 1 / (2 X R1 X fz) (6) SELECTING DIODES The external diode used in Figure 1 should be a Schottky diode. A 20V diode such as the MBR0520 from Fairchild Semiconductor is recommended. The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used. SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER The LM4961 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic speakers, a typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor (Cf2) will be needed as shown in Figure 1 to bandwidth limit the amplifier. www.national.com (5) 10 LM4961 Application Information (Continued) DUTY CYCLE The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined as: Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW This applies for continuous mode operation. 20094099 INDUCTANCE VALUE FIGURE 3. 10µH Inductor Current 5V - 12V Boost (LM4961X) The first question we are usually asked is: “How small can I make the inductor.” (because they are the largest sized component and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be delivered because the energy stored during each switching cycle is: During the 0.390µs ON-time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFFtime. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to about 33mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and continuous operation will be maintained at the typical load current values. Taiyo-Yudens NR4012 inductor series is recommended. E = L/2 X (lp)2 Where “lp” is the peak inductor current. An important point to observe is that the LM4961 will limit its switch current based on peak current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load. Conversely, using too little inductance may limit the amount of load current which can be drawn from the output. Best performance is usually obtained when the converter is operated in “continuous” mode at the load current range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous” over a wider load current range. To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10µH inductor) will be analyzed. We will assume: MAXIMUM SWITCH CURRENT The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in a graph in the typical performance characterization section which shows typical values of switch current as a function of effective (actual) duty cycle. CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP) As shown in Figure 2 which depicts inductor current, the load current is related to the average inductor current by the relation: ILOAD = IIND(AVG) x (1 - DC) (7) Where "DC" is the duty cycle of the application. The switch current can be found by: ISW = IIND(AVG) + 1/2 (IRIPPLE) (8) VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency: Since the frequency is 1.6MHz (nominal), the period is approximately 0.625µs. The duty cycle will be 62.5%, which means the ON-time of the switch is 0.390µs. It should be noted that when the switch is ON, the voltage across the inductor is approximately 4.5V. Using the equation: IRIPPLE = DC x (VIN-VSW) / (f x L) (9) combining all terms, we can develop an expression which allows the maximum available load current to be calculated: V = L (di/dt) ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/2FL(10) We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON-time. Using these facts, we can then show what the inductor current will look like during operation: The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses of the FET and diode. 11 www.national.com LM4961 Application Information GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION (Continued) DESIGN PARAMETERS VSW AND ISW The value of the FET "ON" voltage (referred to as VSW in equations 7 thru 10) is dependent on load current. A good approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current. FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V. This section provides practical guidelines for mixed signal PCB layout that involves various digital/analog power and ground traces. Designers should note that these are only "rule-of-thumb" recommendations and the actual results will depend heavily on the final layout. Power and Ground Circuits For 2 layer mixed signal design, it is important to isolate the digital power and ground trace paths from the analog power and ground trace paths. Star trace routing techniques (bringing individual traces back to a central point rather than daisy chaining traces together in a serial manner) can have a major impact on low level signal performance. Star trace routing refers to using individual traces to feed power and ground to each circuit or even device. This technique will take require a greater amount of design time but will not increase the final price of the board. The only extra parts required may be some jumpers. The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical Performance Characteristics curves. INDUCTOR SUPPLIERS The recommended inductors for the LM4961 is the TaiyoYuden NR4012. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation at peak currents. A suitable core type must be used to minimize core (switching) losses, and wire power losses must be considered when selecting the current rating. Single-Point Power / Ground Connection The analog power traces should be connected to the digital traces through a single point (link). A "Pi-filter" can be helpful in minimizing high frequency noise coupling between the analog and digital sections. It is further recommended to place digital and analog power traces over the corresponding digital and analog ground traces to minimize noise coupling. PCB LAYOUT GUIDELINES High frequency boost converters require very careful layout of components in order to get stable operation and low noise. All components must be as close as possible to the LM4961 device. It is recommended that a 4-layer PCB be used so that internal ground planes are available. See Figures 4–7 for demo board reference schematic and layout. Some additional guidelines to be observed: 1. Keep the path between L1, D1, and Co extremely short. Parasitic trace inductance in series with D1 and Co will increase noise and ringing. 2. The feedback components R1, R2 and Cf 1 must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace. 3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative sides of capacitors Cs1 and Co. www.national.com Placement of Digital and Analog Components All digital components and high-speed digital signals traces should be located as far away as possible from analog components and circuit traces. Avoiding Typical Design / Layout Problems Avoid ground loops or running digital and analog traces parallel to each other (side-by-side) on the same PCB layer. When traces must cross over each other do it at 90 degrees. Running digital and analog traces at 90 degrees to each other from the top to the bottom side as much as possible will minimize capacitive noise coupling and crosstalk. 12 LM4961 Schematic Board Layout 200940A4 FIGURE 4. Demo Board Schematic 13 www.national.com LM4961 Demonstration Board Layout 200940C1 FIGURE 5. Recommended TS SE PCB Layout: Top Silkscreen 200940C0 FIGURE 6. Recommended TS SE PCB Layout: Top Layer www.national.com 14 LM4961 Demonstration Board Layout (Continued) 200940C2 FIGURE 7. Recommended TS SE PCB Layout: Bottom Layer 15 www.national.com LM4961 Ceramic Speaker Driver Physical Dimensions inches (millimeters) unless otherwise noted LQ Package Order Number LM4961LQ NS Package Number LQA28A National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. 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