LM3448 LM3448 Phase Dimmable Offline LED Driver with Integrated FET Literature Number: SNOSB51B LM3448 Phase Dimmable Offline LED Driver with Integrated FET General Description Features The LM3448 is an adaptive constant off-time AC/DC buck (step-down) constant current LED regulator designed to be compatible with TRIAC dimmers. The LM3448 provides a constant current for illuminating high power LEDs and includes a phase angle dim decoder. The dim decoder allows wide range LED dimming using standard forward and reverse phase TRIAC dimmers. The integrated high-voltage and low Rdson MOSFET reduces design complexity while improving LED driver efficiency. The integrated and patented architecture facilitates implementation of small form factor LED drivers suitable for integrated LED lamps with very low external component count. The LM3448 also provides the flexibility required to implement both isolated and non-isolated solutions based on the Flyback, Buck or Buck-Boost topology using either active or passive power factor correction (ValleyFill) circuits. Additional features include thermal shutdown, current limit and VCC under-voltage lockout. ■ Input phase angle dim decoder circuit for LED dimming ■ Integrated, vertical 600V MOSFET with superior avalanche energy capability ■ Application voltage range 85VAC – 265VAC ■ Adjustable switching frequency ■ Adaptive programmable off-time allows for constant ripple ■ ■ ■ ■ ■ current No 120Hz flicker possible Low quiescent current Thermal shutdown Low profile 16-pin Narrow SOIC package Wave solder capable Applications ■ ■ ■ ■ Retrofit TRIAC Dimming Solid State Lighting Industrial and Commercial Lighting Residential Lighting Typical LM3448 LED Driver Application Circuit 301258a0 TRI-STATE® is a registered trademark of National Semiconductor Corporation. © 2011 Texas Instruments Incorporated 301258 www.ti.com LM3448 Phase Dimmable Offline LED Driver with Integrated FET November 8, 2011 LM3448 Connection Diagram Top View 30125873 16-Lead Narrow SOIC Package NS Package Drawing M16A Ordering Information Order Number Spec. Package Type NSC Package Drawing LM3448MA NOPB Narrow SOIC-16 M16A 48 Units, Rails LM3448MAX NOPB Narrow SOIC-16 M16A 2500 Units, Tape and Reel Supplied As Pin Descriptions Pin(s) Name 1, 2, 15, 16 SW Drain connection of internal 600V MOSFET. 3, 14 NC No connect. Provides clearance between high voltage and low voltage pins. Do not tie to GND. 4 BLDR Bleeder pin. Provides the input signal to the angle detect circuitry. A 230Ω internal resistor ensures BLDR is pulled down for proper angle sense detection. 5, 12 GND Circuit ground connection. 6 VCC Input voltage pin. This pin provides the power for the internal control circuitry and gate driver. Connect a 22uF (minimum) bypass capacitor to ground. 7 ASNS PWM output of the TRIAC dim decoder circuit. Outputs a 0 to 4V PWM signal with a duty cycle proportional to the TRIAC dimmer on-time. 8 FLTR1 First filter input. The 120Hz PWM signal from ASNS is filtered to a DC signal and compared to a 1 to 3V, 5.85 kHz ramp to generate a higher frequency PWM signal with a duty cycle proportional to the TRIAC dimmer firing angle. Pull above 4.9V (typical) to TRI-STATE® DIM. 9 DIM 10 COFF OFF time setting pin. A user set current and capacitor connected from the output to this pin sets the constant OFF time of the switching controller. 11 FLTR2 Second filter input. A capacitor tied to this pin filters the PWM dimming signal to supply a DC voltage to control the LED current. Could also be used as an analog dimming input. 13 ISNS LED current sense pin (internally connected to MOSFET source). Connect a resistor from ISNS to GND to set the maximum LED current. www.ti.com Description Input/output dual function dim pin. This pin can be driven with an external PWM signal to dim the LEDs. It may also be used as an output signal and connected to the DIM pin of other LM3448/LM3445 devices or LED drivers to dim multiple LED circuits simultaneously. 2 If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. SW to GND BLDR to GND VCC, FLTR1 to GND ISNS to GND ASNS, DIM, FLTR2, COFF to GND SW FET Drain Current: Peak Continuous -0.3V to +600V -0.3V to +17V -0.3V to +14V -0.3V to +2.5V Operating Conditions -0.3V to +7.0V VCC Junction Temperature Range 1.2A Limited by TJ-MAX Internally Limited 2 kV 125°C -65°C to +150°C 260°C (Note 1) 8V to 12V −40°C to +125°C Electrical Characteristics (Note 1) VCC = 12V unless otherwise noted. Limits in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40°C to +125°C). Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = +25ºC and are provided for reference purposes only. Typ (Note 5) Max (Note 4) Units Bleeder resistance to GND IBLDR = 10mA 230 325 Ω IVCC Operating supply current 2.00 2.85 mA VCC-UVLO Rising threshold 7.4 7.7 V 1.276 1.327 V 60 Ω Symbol Parameter Min (Note 4) Conditions BLEEDER RBLDR VCC SUPPLY Non-switching Falling threshold 6.0 Hysterisis 6.4 1 COFF VCOFF Time out threshold RCOFF Off timer sinking impedance 33 tCOFF Restart timer 180 1.225 µs CURRENT LIMIT VISNS ISNS limit threshold tISNS Leading edge blanking time 125 ns Current limit reset delay 180 µs 5.85 kHz 1.174 1.269 1.364 V INTERNAL PWM RAMP fRAMP Frequency VRAMP Valley voltage 0.96 1.00 1.04 Peak voltage 2.85 3.00 3.08 Maximum duty cycle 96.5 98.0 6.79 7.21 DRAMP V % DIM DECODER VANG_DET Angle detect rising threshold VASNS ASNS filter delay Observed on BLDR pin 4 ASNS VMAX IASNS IDIM 7.81 3.81 3.96 ASNS drive capability sink VASNS = 2V -7.6 ASNS drive capability source VASNS = 2V 4.3 DIM low sink current VDIM = 1V DIM high source current VDIM = 4V -2.80 3.00 3 V µs 4.11 V mA -1.65 4.00 www.ti.com LM3448 Continuous Power Dissipation (Note 2) ESD Susceptibility: HBM (Note 3) Junction Temperature (TJ-MAX) Storage Temperature Range Maximum Lead Temperature (Solder and Reflow) Absolute Maximum Ratings (Note 1) LM3448 Symbol Parameter Conditions VDIM DIM low voltage PWM input voltage threshold Min (Note 4) Typ (Note 5) 0.9 1.33 DIM high voltage VTSTH TRI-STATE threshold voltage RDIM DIM comparator TRISTATE impedance Apply to FLTR1 pin Max (Note 4) Units V 2.33 3.15 4.87 5.25 10 V MΩ CURRENT SENSE COMPARATOR VFLTR2 FLTR2 open circuit voltage RFLTR2 FLTR2 impedance 720 750 780 mV 420 kΩ 660 V OUTPUT MOSFET (SW FET) VBVDS SW to ISNS breakdown voltage IDS SW to ISNS leakage current (Note 8) RON SW to ISNS switch on resistance 600 SW - ISNS = 600V 1 µA 3.6 Ω 165 °C THERMAL SHUTDOWN TSD Thermal shutdown temperature (Note 6) Thermal shutdown hysteresis 20 THERMAL RESISTANCE RθJA Junction to Ambient (Note 6, Note 7) 95 °C/W Note 1: Absolute Maximum Ratings are limits beyond which damage to the component may occur. Operating Ratings are conditions under which operation of the device is guaranteed and do not imply guaranteed performance limits. For guaranteed performance limits and associated test conditions, see the Electrical Characteristics table. All voltages are with respect to the potential at the GND pin unless otherwise specified. Note 2: Internal thermal shutdown circuitry protects the device from permanent damage. Thermal shutdown engages at approximately TJ = 165°C (typ.) and disengages at approximately TJ = 145°C (typ). Note 3: Human Body Model, applicable std. JESD22-A114-C. Note 4: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100% production tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL). Note 5: Typical numbers are at 25°C and represent the most likely norm. Note 6: These electrical parameters are guaranteed by design and are not verified by test. Note 7: This RθJA typical value determined using JEDEC specifications JESD51-1 to JESD51-11. However junction-to-ambient thermal resistance is highly boardlayout dependent. In applications where high maximum power dissipation exists, special care must be paid to thermal dissipation issues during board design. In high-power dissipation applications, the maximum ambient temperature may have to be derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating junction temperature (TJ-MAX-OP = 125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to ambient thermal resistance of the part/package in the application (RθJA), as given by the following equation: TA-MAX = TJ-MAX-OP – (RθJA × PD-MAX). Note 8: High voltage devices such as the LM3448 are susceptible to increased leakage currents when exposed to high humidity and high pressure operating environments. Users of this device are cautioned to satisfy themselves as to the suitability of this product in the intended end application and take any necessary precautions (e.g. system level HAST/HALT testing, conformal coating, potting, etc.) to ensure proper device operation. Note 9: Data used for this plot taken from Design #3. Note 10: Data used for this plot taken from Design #2. www.ti.com 4 TJ = 25°C and VCC = 12V unless otherwise specified. Efficiency vs. Input Line Voltage (Note 9) Power Factor vs. Input Line Voltage (Note 9) 84 0.98 7 LEDs 0.97 POWER FACTOR EFFICIENCY (%) 82 80 78 9 LEDs 76 9 LEDs 0.96 0.95 7 LEDs 0.94 74 80 90 100 110 120 130 INPUT VOLTAGE (VRMS) 0.93 80 140 90 100 110 120 130 INPUT VOLTAGE (VRMS) 30125881 fSW vs. Input Line Voltage (Note 10) 90 220 SWITCHING FREQUENCY (kHz) LED CURRENT (mA) 240 8 LEDs 200 180 10 LEDs 160 12 LEDs 140 80 140 30125882 LED Current vs. Input Line Voltage (Note 10) 90 100 110 120 130 INPUT VOLTAGE (VRMS) 8 LEDs 85 80 75 70 10 LEDs 65 12 LEDs 60 140 80 90 100 110 120 130 INPUT VOLTAGE (VRMS) 30125884 BLDR Resistor vs. Temperature 300 190 280 BLDR RESISTOR (Ω) 200 180 170 160 260 240 220 200 150 -50 -25 140 30125883 Min On-Time (tON) vs. Temperature MIN ON-TIME (ns) LM3448 Typical Performance Characteristics -50 -25 0 25 50 75 100 125 150 TEMPERATURE (°C) 0 25 50 75 100 125 150 TEMPERATURE (°C) 30125802 30125833 5 www.ti.com LM3448 VCC UVLO vs. Temperature VCOFF Threshold vs. Temperature 1.30 UVLO (VCC) Rising VCOFF THRESHOLD (V) UVLO THRESHOLD (V) 8.0 7.5 7.0 UVLO (VCC) Falling 6.5 6.0 -50 -25 1.29 1.28 1.27 1.26 1.25 0 25 50 75 100 125 150 TEMPERATURE (°C) -50 -25 30125814 0 25 50 75 100 125 150 TEMPERATURE (°C) 30125837 Angle Detect Threshold vs. Temperature Leading Edge Blanking Variation Over Temperature ANGLE DETECT THRESHOLD (V) 7.8 7.6 7.4 7.2 7.0 6.8 6.6 -50 -30 -10 10 30 50 70 90 110 130 150 TEMPERATURE (°C) 30125842 30125872 DIM Pin Duty Cycle vs. FLTR1 Voltage (Note 9) DIM PIN DUTY CYCLE (%) 100 80 60 40 20 0 1.0 1.3 1.5 1.8 2.0 2.3 2.5 2.8 3.0 FLTR1 VOLTAGE (V) 30125862 www.ti.com 6 LM3448 Simplified Internal Block Diagram 30125811 Theory of Operation The LM3448 contains all the necessary circuitry to build a linepowered (mains powered) constant current LED driver whose output current can be controlled with a conventional TRIAC dimmer. OVERVIEW OF PHASE CONTROL DIMMING A basic "phase controlled" TRIAC dimmer circuit is shown in Figure 1. 30125812 FIGURE 1. Basic TRIAC Dimmer 30125813 An RC network consisting of R1, R2, and C1 delay the turn on of the TRIAC until the voltage on C1 reaches the trigger voltage of the diac. Increasing the resistance of the potentiometer (wiper moving downward) increases the turn-on delay which decreases the on-time or "conduction angle" of the TRIAC (θ). This reduces the average power delivered to the load. FIGURE 2. Line Voltage and Dimming Waveforms Voltage waveforms for a simple TRIAC dimmer are shown in Figure 2. Figure 2(a) shows the full sinusoid of the input voltage. Even when set to full brightness, few dimmers will provide 100% on-time (i.e. the full sinusoid). Figure 2(b) shows a theoretical waveform from a dimmer. The on-time is often referred to as the "conduction angle" and may be stated in 7 www.ti.com LM3448 degrees or radians. The off-time represents the delay caused by the RC circuit feeding the TRIAC. The off-time can be referred to as the "firing angle" and is simply (180° - θ). Figure 2(c) shows a waveform from a reverse phase dimmer, sometimes referred to as an electronic dimmer. These typically are more expensive, microcontroller based dimmers that use switching elements other than TRIACs. Note that the conduction starts from the zero-crossing and terminates some time later. This method of control reduces the noise spike at the transition. Since the LM3448 has been designed to assess the relative on-time and control the LED current accordingly, most phase control dimmers both forward and reverse phase may be used with success. A bridge rectifier converts the line (mains) voltage of (b) into a series of half-sines as shown in (a). 30125815 FIGURE 3. Voltage Waveforms After Bridge Rectifier Without TRIAC Dimming 30125817 (b) and (a) show typical TRIAC dimmed voltage waveforms before and after the bridge rectifier. FIGURE 5. AC Line Sense Circuitry D1 is typically a 15V zener diode which forces transistor Q1 to “stand-off” most of the rectified line voltage. Having no capacitance on the source of Q1 allows the voltage on the BLDR pin to rise and fall with the rectified line voltage as the line voltage drops below zener voltage D1 (see the section on Angle Detect). A diode-capacitor network (D2, C5) is used to maintain the voltage on the VCC pin while the voltage on the BLDR pin goes low. This provides the supply voltage to operate the LM3448. Resistor R5 is used to bleed charge out of any stray capacitance on the BLDR node and may be used to provide the necessary holding current for the dimmer when operating at light output currents. ANGLE DETECT The Angle Detect circuit uses a comparator with a fixed threshold voltage of 7.21V to monitor the BLDR pin to determine whether the TRIAC is on or off. The output of the comparator drives the ASNS buffer and also controls the bleeder circuit. A 4s delay line on the output is used to filter out noise that could be present on this signal. The output of the Angle Detect circuit is limited to a 0V to 4.0V swing by the buffer and presented to the ASNS pin. R1 and C3 comprise a low-pass filter with a bandwidth on the order of 1.0Hz. 30125816 FIGURE 4. Voltage Waveforms After Bridge Rectifier With TRIAC Dimming SENSING THE RECTIFIED TRIAC WAVEFORM An external series pass regulator (R2, D1, and Q1) translates the rectified line voltage to a level where it can be sensed by the BLDR pin on the LM3448 as shown in Figure 5. www.ti.com 8 The transition from dimming with the DIM decoder to headroom or minimum on-time dimming is seamless. LED currents from full load to as low as 0.5mA can be easily achieved. COFF AND CONSTANT OFF-TIME CONTROL OVERVIEW The LM3448 is a buck regulator that uses a proprietary constant off-time method to maintain constant current through a string of LEDs as shown in Figure 6. BLEEDER While the BLDR pin is below the 7.21V threshold, the internal bleeder MOSFET is on to place a small load (230Ω) on the series pass regulator. This additional load is necessary to complete the circuit through the TRIAC dimmer so that the dimmer delay circuit can operate correctly. Above 7.21V, the bleeder resistor is removed to increase efficiency. FLTR1 PIN The FLTR1 pin has two functions. Normally it is fed by ASNS through filter components R1 and C3 and drives the dim decoder. However if the FLTR1 pin is tied above 4.9V ( e.g., to VCC) the ramp comparator is at TRI-STATE disabling the dim decoder. DIM DECODER The ramp generator produces a 5.85 kHz saw tooth wave with a minimum of 1.0V and a maximum of 3.0V. The filtered ASNS signal enters pin FLTR1 where it is compared against the output of the Ramp Generator. The output of the ramp comparator will have an on-time which is inversely proportional to the average voltage level at pin FLTR1. However since the FLTR1 signal can vary between 0V and 4.0V (the limits of the ASNS pin), and the ramp generator signal only varies between 1.0V and 3.0V, the output of the ramp comparator will be on continuously for VFLTR1 < 1.0V and off continuously for VFLTR1 > 3.0V. This allows a decoding range from 45° to 135° to provide a 0 – 100% dimming range. The output of the ramp comparator drives both a common source N-channel MOSFET through a Schmitt trigger and the DIM pin. The MOSFET drain is pulled up to 750 mV by a 50kΩ resistor. Since the MOSFET inverts the output of the ramp comparator, the drain voltage of the MOSFET is proportional to the duty cycle of the line voltage that comes through the TRIAC dimmer. The amplitude of the ramp generator causes this proportionality to "hard limit" for duty cycles above 75% and below 25%. 30125823 FLTR2 The MOSFET drain signal next passes through an RC filter comprised of an internal 370kΩ resistor and an external capacitor on pin FLTR2. This forms a second low pass filter to further reduce the ripple in this signal which is used as a reference by the PWM comparator. This RC filter is generally set to 10Hz. The net effect is that the output of the dim decoder is a DC voltage whose amplitude varies from near 0V to 750 mV as the duty cycle of the dimmer varies from 25% to 75%. This corresponds to conduction angles of 45° to 135°. The output voltage of the dim decoder directly controls the peak current that will be delivered by the internal SW FET. As the TRIAC fires beyond 135°, the DIM decoder no longer controls the dimming. At this point the LEDs will dim gradually for one of two reasons: • The voltage at VBUCK decreases and the buck converter runs out of headroom and causes LED current to decrease as VBUCK decreases. • Minimum on-time is reached which fixes the duty-cycle and therefore reduces the voltage at VBUCK. FIGURE 6. Simplified Buck Regulation Circuit Constant off-time control architecture operates by simply defining the off-time and allowing the on-time, and therefore the switching frequency, to vary as either VIN or VO changes. The output voltage is equal to the LED string voltage (VLED), and should not change significantly for a given application. The input voltage or VBUCK in this analysis will vary as the input line varies. The length of the on-time is determined by the sensed inductor current through a resistor to a voltage reference at a comparator. During the on-time denoted by tON, the SW FET is on causing the inductor current to increase (see Figure 7). During the on-time, current flows from VBUCK through the LEDs, L2, the LM3448's internal SW FET and finally through R3 to ground. At some point in time the inductor current reaches a maximum (IL2-PK) determined by the voltage at the ISNS pin. This sensed voltage across R3 is compared against the dim decoder voltage on FLTR2 at which point the SW FET is turned off by the regulator. During the off-period denoted by tOFF, the current through L2 continues to flow through the LEDs via D10. Capacitor C12 eliminates most of the ripple current seen in the inductor. Resistor R4, capacitor 9 www.ti.com LM3448 The Angle Detect circuit and its filter produce a DC level which corresponds to the duty cycle (relative on-time) of the TRIAC dimmer. As a result, the LM3448 will work equally well with 50Hz or 60Hz line voltages. LM3448 C11 and transistor Q3 provide a linear current ramp that in conjunction with the COFF comparator threshold sets the constant off-time for a given output voltage. VCC BIAS SUPPLY The LM3448 requires a supply voltage at the VCC pin in the range of 8V to 12V. The device has VCC under-voltage lockout (UVLO) with rising and falling thresholds of 7.4V and 6.4V respectively. Methods for supplying the VCC voltage are discussed in the “Design Considerations” section of this datasheet. THERMAL SHUTDOWN Thermal shutdown limits total power dissipation by turning off the internal SW FET when the IC junction temperature exceeds 165°C. After thermal shutdown occurs, the SW FET will not turn on until the junction temperature drops to approximately 145°C. 30125825 FIGURE 7. Inductor Current Waveform in CCM www.ti.com 10 LM3448 The peak voltage of a two stage valley-fill capacitor is: Design Considerations VALLEY-FILL POWER FACTOR CORRECTION For the non-isolated buck converter, a valley-fill power factor correction (PFC) circuit shown in Figure 8 provides a simple means of improving the converter’s power factor performance. As the AC line decreases from its peak value every cycle, there will be a point where the voltage magnitude of the AC line is equal to the voltage that each capacitor is charged. At this point diode D3 becomes reversed biased, and the capacitors are placed in parallel to each other (see Figure 10) and VBUCK equals the capacitor voltage. 30125818 FIGURE 8. Two Stage Valley Fill Circuit The valley-fill circuit allows the buck regulator to draw power throughout a larger portion of the AC line. This allows the capacitance needed at VBUCK to be lower than if there were no valley-fill circuit and adds passive power factor correction (PFC) to the application. Besides better power factor correction, a valley-fill circuit allows the buck converter to operate while separate circuitry translates the dimming information. This allows for dimming that isn’t subject to 120Hz flicker that can possibly be perceived by the human eye. VBUCK supplies the power which drives the LED string. Diode D3 allows VBUCK to remain high while V+ cycles on and off. VBUCK has a relatively small hold capacitor C10 which reduces the voltage ripple when the valley-fill capacitors are being charged. However, the network of diodes and capacitors shown between D3 and C10 make up a "valley-fill" circuit. The valley-fill circuit can be configured with two or three stages. The most common configuration is two stages which is illustrated in Figure 8. When the “input line is high”, power is derived directly through D3. The term “input line is high” can be explained as follows. The valley-fill circuit charges capacitors C7 and C9 in series when the input line is high (see Figure 9). 30125821 FIGURE 10. Two stage Valley-Fill Circuit when AC Line is Low The valley-fill circuit can be optimized for power factor, voltage hold-up and overall application size and cost. The LM3448 will operate with a single stage or a three stage valley-fill circuit as well. Resistor R8 functions as a current limiting resistor during start-up and during the transition from series to parallel connection. Resistors R6 and R7 are 1MΩ bleeder resistors and may or may not be necessary for each application. FLTR2 LINE-INJECTION The technique of line-injection is another very effective means of improving power factor performance. When using this method, the valley-fill circuit can be eliminated which results in a much simpler driver design. The trade off will be an increase of 120Hz ripple on the LED current. Different FLTR2 circuits are shown in Figure 11. Figure 11(a) shows how to set up FLTR2 when a passive PFC circuit (e.g. valley-fill) is already being used and no line-injection is utilized. If passive PFC is not being implemented, then the “direct line-injection” of Figure 11(b) or “AC line-injection” of Figure 11(c) can be used. Direct line-injection involves injecting a small portion (750mV to 1.00V) of rectified AC line voltage (i.e. V+) into the FLTR2 pin. The result is that current shaping of the input current will yield power factor values greater than 0.94. AC coupled line-injection goes one step further by adding a capacitor C14 between R15 and C11. This improves LED line regulation but does so by trading out a small portion of the power factor improvement from the direct-injection circuit. For example with AC coupled line-injection, LED current regulation of up to +/- 3% is possible for an input voltage range of 105VAC to 135VAC when operating at a nominal 120VAC. 30125819 FIGURE 9. Two stage Valley-Fill Circuit when AC Line is High 11 www.ti.com LM3448 30125886 FIGURE 11. (a) No line-injection, (b) Direct line-injection, (c) AC-coupled line injection By using the line voltage injection technique, the FLTR2 pin has the voltage wave shape shown in Figure 12 on it with no TRIAC dimmer in-line. Peak voltage at the FLTR2 pin should be kept below 1.25V otherwise current limit will be tripped. Capacitor C11 is chosen small enough so as not to distort the AC signal but just add a little filtering. Although the on-time is probably never truly constant, it can be observed in Figure 13 how (by injecting the rectified voltage) the on-time is adjusted. DIRECT LINE-INJECTION FOR FLYBACK TOPOLOGY For flyback converters using the LM3448, direct-line injection can result in power factors greater than 0.95. Using this technique, the LM3448 circuit is essentially turned into a constant power flyback converter operating in discontinuous conduction mode (DCM). The LM3448 normally works as a constant off-time regulator, but by injecting a 1.0VPK rectified AC voltage into the FLTR2 pin, the on-time can be made to be constant. With a DCM flyback converter the primary side current, i, needs to increase as the rectified input voltage, V+, increases as shown in the following equations, or, 30125895 FIGURE 12. FLTR2 Waveform with No Dimmer Therefore a constant on-time (since inductor L is constant) can be obtained. 30125896 FIGURE 13. Typical Operation of Direct Line-Injection into FLTR2 Pin www.ti.com 12 30125885 FIGURE 14. Basic holding current circuit signed to identify the type of phase dimmer in-line with the LM3448, add holding current for different dimming conditions, or to discharge parasitic capacitances. The objective is to only add enough holding current as needed regardless if the dimmer is of a forward or reverse phase type. This allows the lighting manufacturer to optimize efficiency and gain Energy Star approval if desired. OPTIMIZING THE HOLDING CURRENT For optimal system performance and efficiency, only enough holding current should be applied at the right time in the cycle to keep the TRIAC operating properly. This will ensure no variation or ‘flicker’ is seen in the LED light output while improving the circuit efficiency. Circuits that do this are outlined individually as blocks in Figure 15. These circuits are de- 30125878 FIGURE 15. TRIAC holding current circuits 13 www.ti.com LM3448 With a single LM3448 circuit on a common TRIAC dimmer, a holding current resistor between 3kΩ and 5kΩ will be required. As the number of LM3448 circuits added to a single dimmer increases, R4’s resistance can also be increased. A few TRIAC dimmers will require a resistor as low as 1kΩ or smaller for a single LM3448 circuit. Therefore the trade-off will be dimming performance versus efficiency. As the holding resistor R4 is increased, the overall system efficiency will also increase. TRIAC DIMMER HOLDING CURRENT In order to emulate an incandescent light bulb (essentially a resistor) with any LED driver, the existing TRIAC will require a small amount of holding current throughout the AC line cycle. As shown in Figure 14, a simple circuit consisting of R3, D1, Q1 and R4 can accomplish this. With R4 placed on the source of Q1, additional holding current can be pulled from the TRIAC. Most TRIAC dimmers only require a few milliamps of current to hold them on. A few “less expensive” TRIACs sold on the market will require a bit more current. The value of resistor R4 will depend on the type of TRIAC being used and how many light fixtures are running off the TRIAC. LM3448 This circuit adds holding current when a forward phase TRIAC edge is detected. The TRIAC edge detect R-C circuit creates a positive pulse on the base of Q3 each cycle when a forward phase dimmer is present and dimming. The positive pulse turns on Q3 which results in additional holding current being pulled through R9. Reverse Phase Holding Current Circuit This circuit adds holding current when a reverse phase TRIAC edge is detected. The TRIAC edge detect R-C circuit creates a negative pulse on the emitter of Q2 each cycle when a reverse phase dimmer is present and dimming. This turns on Q8 and connects R23 to the Q1 pass MOSFET, adding holding current and sharpening the turn-off of the reverse phase dimmer. Linear Hold Insertion Circuit This circuit adds holding current during low TRIAC conduction angles. A variable voltage between 0 and 5 volts is generated at the Q6 gate by averaging the square wave output signal on the DIM pin. The duty cycle of this square wave varies with the TRIAC firing angle. As the LEDs are dimmed, the voltage at the Q6 gate will rise pulling a “holding current” equal to the Q6 source voltage divided by resistor R19. Valley-Fill Holding Current Circuit As described in the section on valley-fill PFC operation, when the valley-fill capacitors are in parallel there is a brief period of time where the output load is being supplied by these two capacitors. Therefore there is minimal or no line current being drawn from the AC line and the minimum holding current requirement is not met. The TRIAC may turn off at this time which causes phase dimming decode issues. A circuit can be added that detects when the valley-fill capacitors are in parallel. The result is that the gate of Q4 is pulled low, allowing additional hold current to be sourced through resistor R10. TRIAC Edge Detect Circuit During initial turn on (forward phase) or turn off (reverse phase) of a phase dimmer, a little extra holding current is sometimes required to latch the phase dimmer on or discharge any parasitic capacitances on the AC line. In order to determine which dimmer is being used, a TRIAC edge detect circuit is needed. When the TRIAC fires, a sharp edge is created that can be captured by a properly sized R-C circuit. The combination of C3 and R6 creates a positive pulse on R7 for a forward phase dimmer or a negative pulse on R7 for a reverse phase dimmer. The pulse polarity determines whether the forward or reverse phase holding current circuit will be used. The value of R7 can be adjusted to vary the sensitivity of the edge detect circuit. Forward Phase Holding Current Circuit START-UP AND BIAS SUPPLY Figure 16 shows how to generate the necessary VCC bias supply at start-up. Since the AC line peak voltage is always higher than the rating of the regulator, all designs require an N-channel MOSFET (passFET). The passFET (Q1) is connected with its drain attached to the rectified AC. The gate of Q1 is connected to a zener diode (D1) which is then biased from the rectified AC line through series resistance (R3). The source of Q1 is held at a VGS below the zener voltage and current flows through Q1 to charge up whatever capacitance is present. If the capacitance is large enough, the source voltage will remain relatively constant over the line cycle and this becomes the input bias supply at VCC. This bias circuit also enables instant turn-on. However once the circuit is operational, it can be desirable to bootstrap VCC to an auxiliary winding of the inductor or transformer as shown in Figure 17. The two bias paths are each connected to VCC through a diode to ensure the higher of the two is providing VCC current. This bootstrapping greatly improves efficiency while still maintaining quick start-up response. 30125885 FIGURE 16. VCC start-up circuit www.ti.com 14 LM3448 30125887 FIGURE 17. VCC auxiliary winding bias circuit voltage reference for the current source with inherent VCC ripple rejection. LED loads can exhibit voltage drift due to self-heating or external thermal conditions. A change in the LED stack voltage will result in the LED current to drift as well. Figure 18(c) addresses this issue by having the COFF current source referenced to the LED stack voltage using Q1 and ROFF and thereby compensating for LED voltage drift. Another benefit is that the number of series LEDs in the LED string can be changed while still maintaining the same output drive current. COFF CURRENT SOURCE CIRCUITS There are a few different current source circuits that can be used for establishing the LM3448 constant-off time control as shown in Figure 18. Figure 18(a) shows the simplest current source circuit. Capacitor COFF will be charged with a constant current from VCC through resistor ROFF. If there is large noise or ripple on the VCC pin, then the previously described circuit will fluctuate and the off-time will not be constant. The circuit of Figure 18(b) addresses this by using a zener diode D1 across ROFF which establishes a stable 15 www.ti.com LM3448 30125888 FIGURE 18. COFF Current Source Circuits www.ti.com 16 LM3448 Design Guide 30125801 FIGURE 19. Typical Non-Isolated Buck Converter with Valley-Fill PFC The following design guide is an example of how to design the LM3448 as a non-isolated buck converter with valley-fill PFC as shown in Figure 19. For simplicity, choose efficiency between 75% and 85%. CALCULATING OFF-TIME The “Off-Time” of the LM3448 is set by the user and remains fairly constant as long as the voltage of the LED stack remains constant. Calculating the off-time is the first step in determining the switching frequency (fSW) of the converter, which is integral in determining some external component values. PNP transistor Q3, resistor R4, and the LED string voltage define a charging current into capacitor C11. A constant current into a capacitor creates a linear charging characteristic. DETERMINING DUTY-CYCLE (D) Duty cycle (D) approximately equals: With efficiency considered: 17 www.ti.com LM3448 Resistor R4, capacitor C11 and the current through resistor R4 (iCOLL), which is approximately equal to VLED/R4, are all fixed. Therefore, dv is fixed and linear, and dt (i.e. tOFF) can now be calculated. Worst case scenario for minimum on time is when VBUCK is at its maximum voltage (AC high line) and the LED string voltage (VLED) is at its minimum value. The maximum voltage seen by the Buck Converter is: Common equations for determining duty cycle and switching frequency in any buck converter: INDUCTOR SELECTION The controlled off-time architecture of the LM3448 regulates the average current through the inductor (L2), and therefore the LED string current (see Figure 20). The input voltage to the buck converter (VBUCK) changes with line variations and over the course of each half-cycle of the input line voltage. The voltage across the LED string is relatively constant, and therefore the current through R4 is constant. This current sets the off-time of the converter and therefore the output voltsecond product (VLED x off-time) remains constant. A constant volt-second product makes it possible to keep the ripple through the inductor constant as the voltage at VBUCK varies. Therefore: With efficiency of the buck converter in mind: Substitute equations and rearrange: Off-time and switching frequency can now be calculated using the equations above. SETTING THE SWITCHING FREQUENCY Selecting the switching frequency for nominal operating conditions is based on tradeoffs between efficiency (better at low frequency) and solution size/cost (smaller at high frequency). The input voltage to the buck converter (VBUCK) changes with both line variations and over the course of each half-cycle of the input line voltage. The voltage across the LED string will, however, remain constant and therefore the off-time remains constant. The on-time (tON) and therefore the switching frequency, will vary as the VBUCK voltage changes with line voltage. A good design practice is to choose a desired nominal switching frequency knowing that the switching frequency will decrease as the line voltage drops and increase as the line voltage increases. The off-time of the LM3448 can be programmed for switching frequencies ranging from 30 kHz to over 1MHz. A trade-off between efficiency and solution size must be considered when designing the LM3448 application. The maximum switching frequency attainable is limited only by the minimum on-time requirement (200 ns). www.ti.com 30125840 FIGURE 20. Simplified LM3448 Buck Converter The equation for an ideal inductor is: 18 Since the voltage across the SW FET (VDS) is relatively small as is the voltage across sense resistor R3, we can simplify this as approximately, During the off-time, the voltage seen by the inductor is approximately, 30125825 FIGURE 21. Inductor Current Waveform in CCM The value of VL(OFF-TIME) will be relatively constant, because the LED stack voltage will remain constant. If we rewrite the equation for an inductor inserting what we know about the circuit during the off-time, we get, Knowing the desired average LED current (IAVE) and the nominal inductor current ripple (ΔiL), the peak current for an application running in CCM is defined as follows: Or, the maximum (i.e. un-dimmed) LED current would then be, Re-arranging this gives, This is important to calculate because this peak current multiplied by the sense resistor R3 will determine when the internal comparator is tripped. The internal comparator turns the SW FET off once the peak sensed voltage reaches 750 mV. From this we can see that the ripple current (Δi) is proportional to off-time (tOFF) multiplied by a voltage which is dominated by VLED divided by a constant inductance (L2). These equations can be rearranged to calculate the desired value for inductor L2. CURRENT LIMIT Under normal circumstances, the trip voltage on the PWM comparator would be less than or equal to 750 mV depending on the amount of dimming. However if there is a short circuit or an excessive load on the output, higher than normal switch currents will cause a voltage above 1.27V on the ISNS pin which will trip the I-LIM comparator. The I-LIM comparator will reset the RS latch, turning off the internal SW FET. It will also inhibit the Start Pulse Generator and the COFF comparator by holding the COFF pin low. A delay circuit will prevent the start of another cycle for 180µs. where, and finally, VALLEY FILL CAPACITORS The maximum voltage seen by the valley-fill capacitors is, Refer to “Design Example” section of the datasheet to better understand the design process. SETTING THE LED CURRENT Figure 21 shows the inductor current waveform (IL2) when operating in continuous conduction mode (CCM). The This assumes that the capacitors chosen have identical capacitance values and split the line voltage equally. Often a 20% difference in capacitance could be observed between 19 www.ti.com LM3448 LM3448 constant off-time control loop regulates the peak inductor current (IL2-PK). Since the average inductor current equals the average LED current (IAVE), LED current is controlled by regulating the peak inductor current. Given a fixed inductor value, L, this equation states that the change in the inductor current over time is proportional to the voltage applied across the inductor. During the on-time, the voltage applied across the inductor is, LM3448 like capacitors. Therefore a voltage rating margin of 25% to 50% should be considered. The valley-fill capacitors should be sized to supply energy to the buck converter (VBUCK) when the input line is less than its peak divided by the number of stages used in the valley-fill. The capacitance value should be calculated when the TRIAC is not firing (i.e. when full LED current is being drawn by the LED string). The maximum power is delivered to the LED string at this time and therefore the most capacitance will be needed. converter will be before the maximum number of series LEDs allowed can be determined. Two variables will have to be determined in order to accomplish this. 1. AC line operating voltage. This is usually 90VAC to 135VAC for North America. Although the LM3448 can operate at much lower and higher input voltages a range is needed to illustrate the design process. 2. Number of stages being implemented in the valley-fill circuit. In this example a two-stage valley-fill circuit will be used. Figure 23 shows three TRIAC dimmed waveforms. One can easily see that the peak voltage (VPEAK) from 0° to 90° will always be, Once the TRIAC is firing at an angle greater than 90° the peak voltage will lower and be equal to, The voltage at VBUCK with a valley-fill stage of two will look similar to the waveforms of Figure 24. The purpose of the valley-fill circuit is to allow the buck converter to pull power directly off of the AC line when the line voltage is greater than its peak voltage divided by two (for a two stage valley-fill circuit). During this time, the capacitors within the valley-fill circuit (C7 and C9) are charged up to the peak of the AC line voltage. Once the line drops below its peak divided by two, the two capacitors are placed in parallel and deliver power to the buck converter. One can now see that if the peak of the AC line voltage is lowered due to variations in the line voltage, or if the TRIAC is firing at an angle above 90°, the DC offset (VDC) will lower. VDC is the lowest value that voltage VBUCK will encounter. 30125852 FIGURE 22. Two Stage Valley-Fill VBUCK Voltage with no TRIAC Dimming From Figure 22 and the equation for current in a capacitor, the amount of capacitance needed at VBUCK can be calculated using the following method. At 60Hz and a valley-fill circuit of two stages, the hold-up time (tX) required at VBUCK is calculated as follows. The total angle of an AC half cycle is 180° and the total time of a half AC line cycle is 8.33ms. When the angle of the AC waveform is at 30° and 150°, the voltage of the AC line is exactly ½ of its peak. With a two stage valley-fill circuit, this is the point where the LED string switches from power being derived from AC line to power being derived from the hold-up capacitors (C7 and C9). At 60° out of 180° of the cycle or 1/3 of the cycle, the power is derived from the hold-up capacitors (1/3 x 8.33 ms = 2.78 ms). This is equal to the hold-up time (dt) from the above equation, and dv is the amount of voltage the circuit is allowed to droop. From the next section (“Determining Maximum Number of Series Connected LEDs Allowed”) we know the minimum VBUCK voltage will be about 45V for a 90VAC to 135VAC line. At a 90VAC low line operating condition input, ½ of the peak voltage is 64V. Therefore with some margin the voltage at VBUCK cannot droop more than about 15V (dv). (i) is equal to (POUT/ VBUCK), where POUT is equal to (VLED x ILED). Total capacitance (C7 in parallel with C9) can now be calculated. See “ Design Example" section for further calculations of the valley-fill capacitors. Example: Line voltage = 90VAC to 135VAC Valley-fill stages = 2 Depending on what type and value of capacitors are used, some derating should be used for voltage droop when the capacitors are delivering power to the buck converter. When the TRIAC is firing at 135° the current through the LED string will be small. Therefore the droop should be small at this point and a 5% voltage droop should be a sufficient derating. With this derating, the lowest voltage the buck converter will see is about 42.5V in this example. To determine how many LEDs can be driven, take the minimum voltage the buck converter will see (42.5V) and divide it by the worst case forward voltage drop of a single LED. Example: 42.5V/3.7V = 11.5 LEDs (11 LEDs with margin) MAXIMUM NUMBER OF SERIES CONNECTED LEDS A buck converter topology requires that the input voltage (VBUCK) of the output circuit must be greater than the voltage of the LED stack (VLED) for proper regulation. One must determine what the minimum voltage observed by the buck www.ti.com 20 LM3448 30125855 FIGURE 23. VBUCK Waveforms with Various TRIAC Firing Angles 30125856 FIGURE 24. Two Stage Valley-Fill VBUCK Waveforms with Various TRIAC Firing Angles seen at VBUCK. For a common 110VAC ± 20% line, the reverse voltage could be as high as 190V. OUTPUT CAPACITOR A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while keeping the same average current through both the inductor and the LED array. With a buck topology the output inductance (L2) can now be lowered, making the magnetics smaller and less expensive. With a well designed converter, you can assume that all of the ripple will be seen by the capacitor and not the LEDs. One must ensure that the capacitor you choose can handle the RMS current of the inductor. Refer to manufacture’s datasheets to ensure compliance. Usually an X5R or X7R capacitor between 1µF and 10µF of the proper voltage rating will be sufficient. The current rating must be at least, or, RE-CIRCULATING DIODE The LM3448 Buck converter requires a re-circulating diode D10 to carry the inductor current during the off-time of the internal SW FET. The most efficient choice for D10 is a diode with a low forward drop and near-zero reverse recovery time that can withstand a reverse voltage of the maximum voltage Another consideration when choosing a diode is to make sure that the diode’s reverse recovery time is much greater than the leading edge blanking time for proper operation. 21 www.ti.com LM3448 tON(MIN) > 200ns, Design Calculation Example The following design example illustrates the process of actually calculating external component values for a LM3448 nonisolated buck converter with valley-fill PFC according to the following specifications. SPECIFICATIONS: 1. Input voltage range (90VAC – 135VAC) 2. Nominal input voltage = 115VAC 3. Number of LEDs in series = 7 4. Forward voltage drop of a single LED = 3.6V 5. LED stack voltage = (7 x 3.6V) = 25.2V CHOSEN VALUES: 1. Target nominal switching frequency, fSW = 250kHz 2. ILED(AVE) = 400mA 3. POUT = (25.2V) x (400mA) = 10.1W 4. Ripple current Δi (usually 15% - 30% of ILED(AVE)) = (0.30 x 400mA) = 120mA 5. Valley fill stages = 2 6. Assumed minimum efficiency = 80% 5. 6. Calculate C11 and R4: Choose current through R4 (between 50µA and 100µA): 70µA Calculate R4, 7. 8. 9. Choose a standard value of 365kΩ Calculate C11, 10. Choose standard value of 120pF. 11. Calculate inductor value at tOFF = 3µs, CALCULATIONS: 1. Calculate minimum voltage VBUCK equals: 2. Calculate maximum voltage VBUCK voltage, 3. Calculate tOFF at VBUCK nominal line voltage, 12. Choose C10 = 1.0µF, 200V. 13. Calculate valley-fill capacitor values, VAC low line = 90VAC, VBUCK minimum equals 45V (no TRIAC dimming at maximum LED current). Set droop for 20V maximum at full load and low line. Since "i" equals POUT/VBUCK = 224mA, "dV" equals 20V, "dt" equals 2.78ms, and then CTOTAL equals 31µF. Therefore choose C7 = C9 = 15µF. 4. Calculate tON(MIN) at high line to ensure that www.ti.com 22 LM3448 Applications Information DESIGN #1: 7W, 120VAC Non-isolated Buck LED Driver with Valley-Fill PFC SPECIFICATIONS: • AC Input Voltage: 120VAC nominal (85VAC – 135VAC) • Output Voltage: 21.1VDC • LED Output Current: 342mA This TRIAC dimmer compatible design incorporates the following features: • • • • Passive valley-fill PFC for improved power factor performance, Comprehensive TRIAC holding current coverage, Standard VCC start-up and bias circuit, Constant-off time control with LED voltage drift compensation. 23 www.ti.com LM3448 30125877 www.ti.com 24 Part ID Description Manufacturer Part Number U1 IC LED Driver National Semiconductor LM3448MA BR1 Bridge Rectifier Vr = 400V, Io = 0.8A, Vf = 1V Diodes Inc. HD04-T C2 Ceramic, 0.01uF, X7R, 25V, 10% MuRata GRM188R71E103KA01D C3 Ceramic, 1000pF 500V X7R 1206 Kemet C1206C102KCRACTU C12 .01uF KEMIT C1808C103KDRACTU C6, C10 CAP 33uF 100V ELECT NHG RADIAL Panasonic-ECG ECA-2AHG330 C7 22uF, Ceramic, X5R, 25V, 10% MuRata GRM32ER61E226KE15L C8 DNP - - C9 4.7uF C11 DNP - C3216X7R1E475K - C13 Ceramic, 1.0uF 100V X7R 1206 Murata GRM31CR72A105KA01 C14 Ceramic, X7R, 16V, 10% MuRata GRM188R71C474KA88D C15 Ceramic, 0.1uF, X7R, 16V, 10% MuRata GRM188R71C104KA01D C16 Ceramic, 0.22uF, X7R, 16V, 10% Murata GRM188R71E224KA88D C17 Ceramic, 330pF 100V C0G 0603 Murata GCM1885C2A331JA16D D1 DIODE ZENER 225MW 15V SOT23 ON Semiconductor BZX84C15LT1G D2, D3, D5, D6, D7 DIODE FAST REC 200V 1A Rohm Semiconductor RF071M2STR D4 DIODE SWITCH SS DUAL 70V SOT323 Fairchild BAV99WT1G D8 DIODE SUPER FAST 200V 1A SMB Diodes Inc MURS120-13-F F1 FUSE 1A 125V FAST Cooper/Bussman 6125FA1A L2 10mH, FERRITE CHIP POWER 160 OHM Steward HI1206T161R-10 MSS1260-105 L3 1mH, Shielded Drum Core, Coilcraft Inc. Q1 MOSFET N-CHAN 250V 4.4A DPAK Fairchild FDD6N25 Q2, Q3 TRANS NPN 350MW 40V SMD SOT23 Diodes Inc MMBT4401-7-F Q4 MOSFET P-CH 50V 130MA SOT-323 Diodes Inc BSS84W-7-F Q5 TRANS HIVOLT PNP AMP SOT-23 Fairchild MMBTA92 Q6 MOSFET N-CHANNEL 100V SOT323 Diodes Inc BSS123W-7-F Q8 TRANS PNP LP 100MA 30V SOT23 ON Semiconductor BC858CLT1G R2 4.75M, 0805, 1%, 0.125W Vishay-Dale CRCW08054M75FKEA R3 1%, 0.25W Vishay-Dale CRCW1206332kFKEA R4 DNP - - R5, R16 RES 49.9K OHM, 0.1W, 1% 0603 Vishay-Dale CRCW060349k9FKEA R6 RES 100K OHM, 0.25W1%, 1206 Vishay-Dale CRCW1206100kFKEA R7 RES 7.50K OHM, 0.1W, 1% 0603 Vishay-Dale CRCW06037k50FKEA R8 RES 10.0K OHM, 0.1W, 1% 0603 Vishay-Dale CRCW060310k0FKEA R9 RES 100 OHM, 0.25W1%, 1206 Vishay-Dale CRCW1206100RFKEA R10 RES 124 OHM, 0.25W1%, 1206 Vishay-Dale CRCW1206124RFKEA R11 RES 200K OHM, 0.125W, 1%, 0805 Vishay-Dale CRCW0805200kFKEA R12, R13 RES 1.0M OHM, 0.125W, 1%, 0805 Vishay-Dale CRCW08051M00FKEA R14 RES 576K OHM, 1/10W 1% 0603 Vishay-Dale CRCW0603576kFKEA R15 RES 280K OHM, 1/10W 1% 0603 Vishay-Dale CRCW0603280kFKEA R17 DNP - - R18 RES 301 OHM, 0.25W1%, 1206 Vishay-Dale CRCW1206301RFKEA R19 RES 49.9 OHM, 0.125W, 1%, 0805 Vishay-Dale CRCW080549R9FKEA R21 RES 12.1 OHM, 0.25W1%, 1206 Vishay-Dale CRCW120612R1FKEA R22 RES 1.8 OHM 1/3W 5% 1210 Vishay-Dale CRCW12101R80JNEA R23 RES 499 OHM, 0.25W1%, 1206 Vishay-Dale CRCW1206499RFKEA RT1 CURRENT LIM INRUSH 60OHM 20% Canterm MF72-060D5 25 www.ti.com LM3448 DESIGN #1 BILL OF MATERIALS LM3448 DESIGN #2: 6.5W, 120VAC Non-isolated “A19 Edison” Retrofit with AC-Coupled Line Injection SPECIFICATIONS: • • • AC Input Voltage: 120VAC nominal (85VAC – 135VAC) Output Voltage: 35.7VDC LED Output Current: 181mA This TRIAC dimmer compatible design incorporates the following features: • • • AC coupled line-injection for improved power factor performance and LED current regulation, Standard VCC start-up and bias circuit, VCC derived COFF current source. NOTE: Refer to LM3448 Application Note, AN-2127, for additional information and BOM regarding this design. 30125874 www.ti.com 26 LM3448 DESIGN #3: 6W, 120VAC Isolated Flyback LED Driver with Direct Line Injection SPECIFICATIONS: • • • AC Input Voltage: 120VAC nominal (85VAC – 135VAC) Flyback Output Voltage: 27.1VDC LED Output Current: 228mA This TRIAC dimmer compatible design incorporates the following features: • • • • • Direct line-injection for improved power factor performance, Standard VCC start-up with auxiliary winding bias circuit for improved system efficiency, Zener diode derived COFF current source for improved VCC ripple rejection, Additional TRIAC holding current circuit for improved dimmer performance at low conduction angles, Output overvoltage protection (OVP). NOTE: Refer to LM3448 Application Note, AN-2090, for additional information and BOM regarding this design. 30125875 27 www.ti.com LM3448 DESIGN #4: 6W, 230VAC Isolated Flyback LED Driver with Direct Line Injection SPECIFICATIONS: • • • AC Input Voltage: 230VAC nominal (180VAC – 265VAC) Flyback Output Voltage: 27.0VDC LED Output Current: 226mA This TRIAC dimmer compatible design incorporates the following features: • • • • • Direct line-injection for improved power factor performance Standard VCC start-up with auxiliary winding bias circuit for improved system efficiency, VCC derived COFF current source, Additional TRIAC holding current circuit for improved dimmer performance at low conduction angles, Output overvoltage protection (OVP). NOTE: Refer to LM3448 Application Note, AN-2091, for additional information and BOM regarding this design. 30125876 www.ti.com 28 LM3448 Physical Dimensions inches (millimeters) unless otherwise noted Narrow SOIC-16 Pin Package For Ordering, Refer to Ordering Information Table NS Package Number M16A 29 www.ti.com LM3448 Phase Dimmable Offline LED Driver with Integrated FET Notes TI/NATIONAL INTERIM IMPORTANT NOTICE Texas Instruments has purchased National Semiconductor. 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