LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs General Description Features Integrated with a 1A power switch, the LM3405 is a currentmode control switching buck regulator designed to provide a simple, high efficiency solution for driving high power LEDs. With a 0.205V reference voltage feedback control to minimize power dissipation, an external resistor sets the current as needed for driving various types of LEDs. Switching frequency is internally set to 1.6MHz, allowing small surface mount inductors and capacitors to be used. The LM3405 utilizes current-mode control and internal compensation offering ease of use and predictable, high performance regulation over a wide range of operating conditions. Additional features include user accessible EN/DIM pin for enabling and PWM dimming of LEDs, thermal shutdown, cycle-by-cycle current limit and over-current protection. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ VIN operating range of 3V to 15V Thin SOT23-6 package 1.6MHz switching frequency 300mΩ NMOS switch 40nA shutdown current at VIN = 5V EN/DIM input for enabling and PWM dimming of LEDs Internally compensated current-mode control Cycle-by-cycle current limit Input voltage UVLO Over-current protection Thermal shutdown Applications ■ ■ ■ ■ LED Driver Constant Current Source Industrial Lighting LED Flashlights Typical Application Circuit Efficiency vs LED Current (VIN = 5V) 20178901 20178971 © 2007 National Semiconductor Corporation 201789 www.national.com LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs February 2007 LM3405 Connection Diagrams 20178905 20178960 6-Lead TSOT NS Package Number MK06A Pin 1 Identification Ordering Information Part Number LM3405XMK LM3405XMKX Package Type TSOT-6 NS Package Drawing MK06A Package Marking Supplied As SPNB 1000 Units on Tape and Reel SPNB 3000 Units on Tape and Reel *NOPB versions are available Pin Descriptions Pin(s) Name 1 BOOST 2 GND 3 FB 4 EN/DIM Application Information Voltage at this pin drives the internal NMOS power switch. A bootstrap capacitor is connected between the BOOST and SW pins. Signal and Power ground pin. Place the LED current-setting resistor as close as possible to this pin for accurate current regulation. Feedback pin. Connect an external resistor from FB to GND to set the LED Current. Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square wave of varying duty cycle at different frequencies controls the brightness of LEDs. Do not allow this pin to float or be greater than VIN + 0.3V. 5 VIN Input supply voltage. Connect a bypass capacitor locally from this pin to GND. 6 SW Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor. www.national.com 2 If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN SW Voltage Boost Voltage Boost to SW Voltage FB Voltage EN/DIM Voltage Junction Temperature -0.5V to 20V -0.5V to 20V -0.5V to 26V -0.5V to 6.0V -0.5V to 3.0V -0.5V to (VIN + 0.3V) 150°C Operating Ratings 2kV -65°C to +150°C 220°C (Note 1) VIN 3V to 15V -0.5V to (VIN + 0.3V) 2.5V to 5.5V -40°C to +125°C EN/DIM voltage Boost to SW Voltage Junction Temperature Range Thermal Resistance θJA (Note 3) 118°C/W Electrical Characteristics Unless otherwise specified, VIN = 12V. Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm, and are provided for reference purposes only. Symbol VFB Parameter Conditions Feedback Voltage Min Typ Max Units 0.188 0.205 0.220 V ΔVFB/(ΔVINxVFB) Feedback Voltage Line Regulation VIN = 3V to 15V IFB UVLO Feedback Input Bias Current Sink/Source Under-voltage Lockout VIN Rising Under-voltage Lockout VIN Falling 0.01 1.9 UVLO Hysteresis %/V 10 250 nA 2.74 2.95 V 2.3 V 0.44 Switching Frequency DMAX Maximum Duty Cycle VFB = 0V RDS(ON) Switch ON Resistance VBOOST - VSW = 3V Switch Current Limit VBOOST - VSW = 3V, VIN = 3V Quiescent Current Switching, VFB = 0.195V Quiescent Current (Shutdown) VEN/DIM = 0V Enable Threshold Voltage VEN/DIM Rising Shutdown Threshold Voltage VEN/DIM Falling EN/DIM Pin Current Sink/Source 0.01 µA Switch Leakage VIN = 15V 0.1 µA ICL IQ VEN/DIM_TH IEN/DIM ISW 1.2 85 1.2 1.6 V fSW 1.9 MHz 300 600 mΩ 2.0 2.8 A 1.8 2.8 mA 94 % 0.3 µA 1.8 V 0.4 V Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under which the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: Human body model, 1.5kΩ in series with 100pF. Note 3: Thermal shutdown will occur if the junction temperature (TJ) exceeds 165°C. The maximum allowable power dissipation (PD) at any ambient temperature (TA) is PD = (TJ(MAX) – TA)/θJA . This number applies to packages soldered directly onto a 3" x 3" PC board with 2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still air, θJA = 204°C/W. 3 www.national.com LM3405 ESD Susceptibility (Note 2) Storage Temperature Soldering Information Infrared/Convection Reflow (15sec) Absolute Maximum Ratings (Note 1) LM3405 Typical Performance Characteristics Unless otherwise specified, VIN = 12V, VBOOST - VSW = 5V and TA = 25°C. Efficiency vs LED Current Efficiency vs Input Voltage (IF = 1A) 20178973 20178931 Efficiency vs Input Voltage (IF = 0.7A) Efficiency vs Input Voltage (IF = 0.35A) 20178932 20178933 VFB vs Temperature Oscillator Frequency vs Temperature 20178936 20178927 www.national.com 4 RDS(ON) vs Temperature (VBOOST - VSW = 3V) 20178972 20178930 Quiescent Current vs Temperature Startup Response to EN/DIM Signal (VIN = 15V, IF = 0.2A) 20178934 20178968 5 www.national.com LM3405 Current Limit vs Temperature LM3405 Block Diagram 20178906 FIGURE 1. Simplified Block Diagram Application Information THEORY OF OPERATION The LM3405 is a PWM, current-mode control switching buck regulator designed to provide a simple, high efficiency solution for driving LEDs with a preset switching frequency of 1.6MHz. This high frequency allows the LM3405 to operate with small surface mount capacitors and inductors, resulting in LED drivers that need only a minimum amount of board space. The LM3405 is internally compensated, simple to use, and requires few external components. The following description of operation of the LM3405 will refer to the Simplified Block Diagram (Figure 1) and to the waveforms in Figure 2. The LM3405 supplies a regulated output current by switching the internal NMOS power switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the internal NMOS power switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and the inductor current (IL) increases with a linear slope. IL is measured by the current sense amplifier, which generates an output proportional to the switch current. The sense signal is summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the feedback voltage and VREF. When the PWM comparator output goes high, the internal power switch turns off until the next switching cycle begins. During the switch off-time, inductor current discharges through the catch diode D1, which forces the SW pin to swing below ground by the forward voltage (VD1) of the catch diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output current (IF) through the LED, by forcing FB pin voltage to be equal to VREF (0.205V). www.national.com 20178907 FIGURE 2. SW Pin Voltage and Inductor Current Waveforms of LM3405 BOOST FUNCTION Capacitor C3 and diode D2 in Figure 1 are used to generate a voltage VBOOST. The voltage across C3, VBOOST - VSW, is the gate drive voltage to the internal NMOS power switch. To properly drive the internal NMOS switch during its on-time, VBOOST needs to be at least 2.5V greater than VSW. Large value of VBOOST - VSW is recommended to achieve better efficiency by minimizing both the internal switch ON resistance (RDS(ON)), and the switch rise and fall times. However, VBOOST - VSW should not exceed the maximum operating limit of 5.5V. 6 placing a zener diode D3 in series with D2 as shown in Figure 4. When using a series zener diode from the input, the gate drive voltage is VIN - VD3 - VD2 + VD1. VBOOST - VSW = VIN - VD2 + VD1 FIGURE 4. VBOOST derived from VIN through a Series Zener 20178999 When the NMOS switch turns on (TON), the switch pin rises to: An alternate method is to place the zener diode D3 in a shunt configuration as shown in Figure 5. A small 350mW to 500mW, 5.1V zener in a SOT-23 or SOD package can be used for this purpose. A small ceramic capacitor such as a 6.3V, 0.1µF capacitor (C5) should be placed in parallel with the zener diode. When the internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The 0.1µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time. Resistor R2 should be chosen to provide enough RMS current to the zener diode and to the BOOST pin. A recommended choice for the zener current (IZENER) is 1mA. The current IBOOST into the BOOST pin supplies the gate current of the NMOS power switch. It reaches a maximum of around 3.6mA at the highest gate drive voltage of 5.5V over the LM3405 operating range. For the worst case IBOOST, increase the current by 50%. In that case, the maximum boost current will be: VSW = VIN – (RDS(ON) x IL) Since the voltage across C3 remains unchanged, VBOOST is forced to rise thus reverse biasing D2. The voltage at VBOOST is then: VBOOST = 2VIN – (RDS(ON) x IL) – VD2 + VD1 Depending on the quality of the diodes D1 and D2, the gate drive voltage in this method can be slightly less or larger than the input voltage VIN. For best performance, ensure that the variation of the input supply does not cause the gate drive voltage to fall outside the recommended range: 2.5V < VIN - VD2 + VD1 < 5.5V The second method for deriving the boost voltage is to connect D2 to the output as shown in Figure 3. The gate drive voltage in this configuration is: VBOOST - VSW = VOUT – VD2 + VD1 IBOOST-MAX = 1.5 x 3.6mA = 5.4mA Since the gate drive voltage needs to be in the range of 2.5V to 5.5V, the output voltage VOUT should be limited to a certain range. For the calculation of VOUT, see OUTPUT VOLTAGE section. R2 will then be given by: R2 = (VIN - VZENER) / (IBOOST_MAX + IZENER) For example, let VIN = 12V, VZENER = 5V, IZENER = 1mA, then: R2 = (12V - 5V) / (5.4mA + 1mA) = 1.09kΩ 20178993 FIGURE 3. VBOOST derived from VOUT 20178994 The third method can be used in the applications where both VIN and VOUT are greater than 5.5V. In these cases, C3 cannot be charged directly from these voltages; instead C3 can be charged from VIN or VOUT minus a zener voltage (VD3) by FIGURE 5. VBOOST derived from VIN through a Shunt Zener 7 www.national.com LM3405 When the LM3405 starts up, internal circuitry from VIN supplies a 20mA current to the BOOST pin, flowing out of the BOOST pin into C3. This current charges C3 to a voltage sufficient to turn the switch on. The BOOST pin will continue to source current to C3 until the voltage at the feedback pin is greater than 123mV. There are various methods to derive VBOOST: 1. From the input voltage (VIN) 2. From the output voltage (VOUT) 3. From a shunt or series zener diode 4. From an external distributed voltage rail (VEXT) The first method is shown in the Simplified Block Diagram of Figure 1. Capacitor C3 is charged via diode D2 by VIN. During a normal switching cycle, when the internal NMOS power switch is off (TOFF) (refer to Figure 2), VBOOST equals VIN minus the forward voltage of D2 (VD2), during which the current in the inductor (L1) forward biases the catch diode D1 (VD1). Therefore the gate drive voltage stored across C3 is: LM3405 The fourth method can be used in an application which has an external low voltage rail, VEXT. C3 can be charged through D2 from VEXT, independent of VIN and VOUT voltage levels. Again for best performance, ensure that the gate drive voltage, VEXT - VD2 + VD1, falls in the range of 2.5V to 5.5V. adding this Ra-Ca network on the LED current is shown in Figure 8. For a fast rising input voltage (200µs for example), there is no need to delay the EN/DIM signal since soft-start can smoothly bring up the LED current as shown in Figure 9. SETTING THE LED CURRENT LM3405 is a constant current buck regulator. The LEDs are connected between VOUT and FB pin as shown in the Typical Application Circuit. The FB pin is at 0.205V in regulation and therefore the LED current IF is set by VFB and the resistor R1 from FB to ground by the following equation: IF = VFB / R1 IF should not exceed the 1A current capability of LM3405 and therefore R1 minimum must be approximately 0.2Ω. IF should also be kept above 200mA for stable operation, and therefore R1 maximum must be approximately 1Ω. If average LED currents less than 200mA are desired, the EN/DIM pin can be used for PWM dimming. See LED PWM DIMMING section. OUTPUT VOLTAGE The output voltage is primarily determined by the number of LEDs (n) connected from VOUT to FB pin and therefore VOUT can be written as : 20178976 VOUT = ((n x VF) + VFB) FIGURE 6. Startup Response to VIN with 5ms rise time where VF is the forward voltage of one LED at the set LED current level (see LED manufacturer datasheet for forward characteristics curve). ENABLE MODE / SHUTDOWN MODE The LM3405 has both enable and shutdown modes that are controlled by the EN/DIM pin. Connecting a voltage source greater than 1.8V to the EN/DIM pin enables the operation of LM3405, while reducing this voltage below 0.4V places the part in a low quiescent current (0.3µA typical) shutdown mode. There is no internal pull-up on EN/DIM pin, therefore an external signal is required to initiate switching. Do not allow this pin to float or rise to 0.3V above VIN. It should be noted that when the EN/DIM pin voltage rises above 1.8V while the input voltage is greater than UVLO, there is a finite delay before switching starts. During this delay the LM3405 will go through a power on reset state after which the internal softstart process commences. The soft-start process limits the inrush current and brings up the LED current (IF) in a smooth and controlled fashion. The total combined duration of the power on reset delay, soft-start delay and the delay to fully establish the LED current is in the order of 100µs (refer to Figure 11). The simplest way to enable the operation of LM3405 is to connect the EN/DIM pin to VIN which allows self start-up of LM3405 whenever the input voltage is applied. However, when an input voltage of slow rise time is used to power the application and if both the input voltage and the output voltage are not fully established before the soft-start time elapses, the control circuit will command maximum duty cycle operation of the internal power switch to bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.205V, the duty cycle will have to reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for this reduction of duty cycle and this will result in a spike in LED current for a short duration as shown in Figure 6. In applications where this LED current overshoot is undesirable, EN/ DIM pin voltage can be delayed with respect to VIN such that VIN is fully established before the EN/DIM pin voltage reaches the enable threshold. This delay can be implemented by a simple Ra-Ca network as shown in Figure 7. The effect of www.national.com 20178998 FIGURE 7. EN/DIM delayed with respect to VIN 20178977 FIGURE 8. Startup Response to VIN with EN/DIM delayed 8 LM3405 20178967 20178975 FIGURE 11. Startup Response to EN/DIM with IF = 1A FIGURE 9. Startup Response to VIN with 200µs rise time LED PWM DIMMING The LED brightness can be controlled by applying a periodic pulse signal to the EN/DIM pin and varying its frequency and/ or duty cycle. This so-called PWM dimming method controls the average light output by pulsing the LED current between the set value and zero. A logic high level at the EN/DIM pin turns on the LED current whereas a logic low level turns off the LED current. Figure 10 shows a typical LED current waveform in PWM dimming mode. As explained in the previous section, there is approximately a 100µs delay from the EN/ DIM signal going high to fully establishing the LED current as shown in Figure 11. This 100µs delay sets a maximum frequency limit for the driving signal that can be applied to the EN/DIM pin for PWM dimming. Figure 12 shows the average LED current versus duty cycle of PWM dimming signal for various frequencies. The applicable frequency range to drive LM3405 for PWM dimming is from 100Hz to 5kHz. The dimming ratio reduces drastically when the applied PWM dimming frequency is greater than 5kHz. 20178983 FIGURE 12. Average LED Current versus Duty Cycle of PWM Dimming Signal at EN/DIM Pin UNDER-VOLTAGE LOCKOUT Under-voltage lockout (UVLO) prevents the LM3405 from operating until the input voltage exceeds 2.74V (typical). The UVLO threshold has approximately 440mV of hysteresis, so the part will operate until VIN drops below 2.3V (typical). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic. CURRENT LIMIT The LM3405 uses cycle-by-cycle current limit to protect the internal power switch. During each switching cycle, a current limit comparator detects if the power switch current exceeds 2.0A (typical), and turns off the switch until the next switching cycle begins. 20178966 OVER-CURRENT PROTECTION The LM3405 has a built in over-current comparator that compares the FB pin voltage to a threshold voltage that is 60% higher than the internal reference VREF. Once the FB pin voltage exceeds this threshold level (typically 328mV), the internal NMOS power switch is turned off, which allows the feedback voltage to decrease towards regulation. This FIGURE 10. PWM Dimming of LEDs using the EN/DIM Pin 9 www.national.com LM3405 threshold provides an upper limit for the LED current. LED current overshoot is limited to 328mV/R1 by this comparator during transients. sponse. Ceramic capacitors allow higher inductor ripple without significantly increasing LED current ripple. See the output capacitor and feed-forward capacitor sections for more details on LED current ripple. Now that the ripple current or ripple ratio is determined, the inductance is calculated by: THERMAL SHUTDOWN Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction temperature exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on until the junction temperature drops below approximately 150°C. Design Guide where fSW is the switching frequency and IF is the LED current. When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating. Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating correctly. Because of the operating frequency of LM3405, ferrite based inductors are preferred to minimize core losses. This presents little restriction since the variety of ferrite based inductors is huge. Lastly, inductors with lower series resistance (DCR) will provide better operating efficiency. For recommended inductor selection, refer to Circuit Examples and Recommended Inductance Range in Table 1. Note that it is a good practice to use small inductance value at light load (for example, IF = 0.2A) to increase inductor current ramp signal, such that noise immunity is improved. INDUCTOR (L1) The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN): The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to calculate a more accurate duty cycle. Calculate D by using the following formula: TABLE 1. Recommended Inductance Range IF Inductance Range and Inductor Current Ripple 4.7µH-10µH VSW can be approximated by: 1.0A VSW = IF x RDS(ON) ΔiL / IF* The diode forward drop (VD1) can range from 0.3V to 0.7V depending on the quality of the diode. The lower VD1 is, the higher the operating efficiency of the converter. The inductor value determines the output ripple current (ΔiL, as defined in Figure 2). Lower inductor values decrease the size of the inductor, but increases the output ripple current. An increase in the inductor value will decrease the output ripple current. The ratio of ripple current to LED current is optimized when it is set between 0.3 and 0.4 at 1A LED current. This ratio r is defined as: 4.7µH 6.8µH 10µH 51% 35% 24% 6.8µH-15µH 0.6A Inductance 6.8µH 10µH 15µH ΔiL / IF* 58% 40% 26% 4.7µH**-22µH 0.2A Inductance 10µH 15µH 22µH ΔiL / IF* 119% 79% 54% *Maximum over full range of VIN and VOUT. **Small inductance improves stability without causing a significant increase in LED current ripple. INPUT CAPACITOR (C1) An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL (Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any recommended deratings and also verify if there is any significant change in capacitance at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current rating (IRMS-IN) must be greater than: One must also ensure that the minimum current limit (1.2A) is not exceeded, so the peak current in the inductor must be calculated. The peak current (ILPK) in the inductor is calculated as: ILPK = IF + ΔiL/2 When the designed maximum output current is reduced, the ratio r can be increased. At a current of 0.2A, r can be made as high as 0.7. The ripple ratio can be increased at lighter loads because the net ripple is actually quite low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for the maximum ripple ratio at any current below 2A is: It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always calculate the RMS at the point where the duty cycle D, is closest to 0.5. The ESL of an input capacitor is usually determined by the effective cross sectional area of the current path. A large leaded r = 0.387 x IOUT-0.3667 Note that this is just a guideline. The LM3405 operates at a high frequency allowing the use of ceramic output capacitors without compromising transient rewww.national.com Inductance 10 OUTPUT CAPACITOR (C2) The output capacitor is selected based upon the desired reduction in LED current ripple. A 1µF ceramic capacitor results in very low LED current ripple for most applications. Due to the high switching frequency, the 1µF capacitor alone (without feed-forward capacitor C4) can filter more than 90% of the inductor current ripple for most applications where the sum of LED dynamic resistance and R1 is larger than 1Ω. Since the internal compensation is tailored for small output capacitance with very low ESR, it is strongly recommended to use a ceramic capacitor with capacitance less than 3.3µF. Given the availability and quality of MLCCs and the expected output voltage of designs using the LM3405, there is really no need to review other capacitor technologies. A benefit of ceramic capacitors is their ability to bypass high frequency noise. A certain amount of switching edge noise will couple through the parasitic capacitances in the inductor to the output. A ceramic capacitor will bypass this noise. In cases where large capacitance is required, use Electrolytic or Tantalum capacitors with large ESR, and verify the loop performance on bench. Like the input capacitor, recommended multilayer ceramic capacitors are X7R or X5R. Again, verify actual capacitance at the desired operating voltage and temperature. Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor is: 20178969 FIGURE 13. PWM Dimming without Feed-Forward Capacitor 20178970 FIGURE 14. PWM Dimming with a 1µF Feed-Forward Capacitor One may select a 1206 size ceramic capacitor for C2, since its current rating is typically higher than 1A, more than enough for the requirement. CATCH DIODE (D1) The catch diode (D1) conducts during the switch off-time. A Schottky diode is required for its fast switching time and low forward voltage drop. The catch diode should be chosen such that its current rating is greater than: FEED-FORWARD CAPACITOR (C4) The feed-forward capacitor (designated as C4) connected in parallel with the LED string is required to provide multiple benefits to the LED driver design. It greatly improves the large signal transient response and suppresses LED current overshoot that may otherwise occur during PWM dimming; it also helps to shape the rise and fall times of the LED current pulse during PWM dimming thus reducing EMI emission; it reduces LED current ripple by bypassing some of inductor ripple from flowing through the LED. For most applications, a 1µF ceramic capacitor is sufficient. In fact, the combination of a 1µF feed-forward ceramic capacitor and a 1µF output ceramic capacitor leads to less than 1% current ripple flowing through the LED. Lower and higher C4 values can be used, but bench validation is required to ensure the performance meets the application requirement. Figure 13 shows a typical LED current waveform during PWM dimming without feed-forward capacitor. At the beginning of ID1 = IF x (1-D) The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin. To improve efficiency, choose a Schottky diode with a low forward voltage drop. BOOST DIODE (D2) A standard diode such as the 1N4148 type is recommended. For VBOOST circuits derived from voltages less than 3.3V, a small-signal Schottky diode is recommended for better efficiency. A good choice is the BAT54 small signal diode. 11 www.national.com LM3405 each PWM cycle, overshoot can be seen in the LED current. Adding a 1µF feed-forward capacitor can totally remove the overshoot as shown in Figure 14. capacitor will have high ESL and a 0805 ceramic chip capacitor will have very low ESL. At the operating frequency of the LM3405, certain capacitors may have an ESL so large that the resulting inductive impedance (2πfL) will be higher than that required to provide stable operation. It is strongly recommended to use ceramic capacitors due to their low ESR and low ESL. A 10µF multilayer ceramic capacitor (MLCC) is a good choice for most applications. In cases where large capacitance is required, use surface mount capacitors such as Tantalum capacitors and place at least a 1µF ceramic capacitor close to the VIN pin. For MLCCs it is recommended to use X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating conditions. LM3405 BOOST CAPACITOR (C3) A 0.01µF ceramic capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide the best performance. PCB Layout Considerations When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The most important consideration when completing the layout is the close coupling of the GND connections of the input capacitor C1 and the catch diode D1. These ground ends should be close to one another and be connected to the GND plane with at least two through-holes. Place these components as close to the IC as possible. The next consideration is the location of the GND connection of the output capacitor C2, which should be near the GND connections of C1 and D1. There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching node island. The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup that causes inaccurate regulation. The LED current setting resistor R1 should be placed as close as possible to the IC, with the GND of R1 placed as close as possible to the GND of the IC. The VOUT trace to LED anode should be routed away from the inductor and any other traces that are switching. High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible. Radiated noise can be decreased by choosing a shielded inductor. The remaining components should also be placed as close as possible to the IC. Please see Application Note AN-1229 for further considerations and the LM3405 demo board as an example of a four-layer layout. POWER LOSS ESTIMATION The main power loss in LM3405 includes three basic types of loss in the internal power switch: conduction loss, switching loss, and gate charge loss. In addition, there is loss associated with the power required for the internal circuitry of IC. The conduction loss is calculated as: If the inductor ripple current is fairly small (for example, less than 40%) , the conduction loss can be simplified to: PCOND = IF2 x RDS(ON) x D The switching loss occurs during the switch on and off transition periods, where voltage and current overlap resulting in power loss. The simplest means to determine this loss is to empirically measure the rise and fall times (10% to 90%) of the voltage at the switch pin. Switching power loss is calculated as follows: PSW = 0.5 x VIN x IF x fSW x ( TRISE + TFALL ) The gate charge loss is associated with the gate charge QG required to drive the switch: PG = fSW x VIN x QG The power loss required for operation of the internal circuitry: PQ = IQ x VIN IQ is the quiescent operating current, and is typically around 1.8mA for the LM3405. The total power loss in the IC is: PINTERNAL = PCOND + PSW + PG + PQ An example of power losses for a typical application is shown in Table 2: TABLE 2. Power Loss Tabulation Conditions Power loss VIN 12V VOUT 4.1V IOUT 1.0A VD1 0.45V RDS(ON) 300mΩ PCOND 111mW PSW 288mW fSW 1.6MHz TRISE 18ns TFALL 12ns IQ 1.8mA PQ 22mW QG 1.4nC PG 27mW D is calculated to be 0.37 Σ ( PCOND + PSW + PQ + PG ) = PINTERNAL PINTERNAL = 448mW www.national.com 12 LM3405 LM3405 Circuit Examples 20178942 FIGURE 15. VBOOST derived from VIN ( VIN = 5V, IF = 1A ) Bill of Materials for Figure 15 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405 National Semiconductor C1, Input Cap 10µF, 6.3V, X5R C3216X5R0J106M TDK C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.37V at 1A, VR = 10V MBRM110LT1G ON Semiconductor D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor L1 4.7µH, 1.6A SLF6028T-4R7M1R6 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1A, White LED LXHL-PW09 Lumileds 13 www.national.com LM3405 20178943 FIGURE 16. VBOOST derived from VOUT ( VIN = 12V, IF = 1A ) Bill of Materials for Figure 16 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405 National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor L1 4.7µH, 1.6A SLF6028T-4R7M1R6 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1A, White LED LXHL-PW09 Lumileds www.national.com 14 LM3405 20178944 FIGURE 17. VBOOST derived from VIN through a Shunt Zener Diode (D3) ( VIN = 15V, IF = 1A ) Bill of Materials for Figure 17 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405 National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C5, Shunt Cap 0.1µF, 16V, X7R GRM219R71C104KA01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Zener Diode 4.7V, 350mW, SOT-23 BZX84C4V7 Fairchild L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay R2 1.91kΩ, 1% CRCW08051K91FKEA Vishay LED1 1A, White LED LXHL-PW09 Lumileds 15 www.national.com LM3405 20178949 FIGURE 18. VBOOST derived from VIN through a Series Zener Diode (D3) ( VIN = 15V, IF = 1A ) Bill of Materials for Figure 18 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405 National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 10V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Zener Diode 11V, 350mW, SOT-23 BZX84C11 Fairchild L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1A, White LED LXHL-PW09 Lumileds www.national.com 16 LM3405 20178950 FIGURE 19. VBOOST derived from VOUT through a Series Zener Diode (D3) ( VIN = 15V, IF = 1A ) Bill of Materials for Figure 19 Part ID Part Value Part Number Manufacturer U1 1A LED Driver LM3405 National Semiconductor C1, Input Cap 10µF, 25V, X5R ECJ-3YB1E106K Panasonic C2, Output Cap 1µF, 16V, X7R GRM319R71A105KC01D Murata C3, Boost Cap 0.01µF, 16V, X7R 0805YC103KAT2A AVX C4, Feedforward Cap 1µF, 16V, X7R GRM319R71A105KC01D Murata D1, Catch Diode Schottky, 0.5V at 1A, VR = 30V SS13 Vishay D2, Boost Diode Schottky, 0.36V at 15mA CMDSH-3 Central Semiconductor D3, Zener Diode 3.9V, 350mW, SOT-23 BZX84C3V9 Fairchild L1 6.8µH, 1.5A SLF6028T-6R8M1R5 TDK R1 0.2Ω, 0.5W, 1% WSL2010R2000FEA Vishay LED1 1A, White LED LXHL-PW09 Lumileds LED2 1A, White LED LXHL-PW09 Lumileds 17 www.national.com LM3405 Physical Dimensions inches (millimeters) unless otherwise noted 6-Lead TSOT Package NS Package Number MK06A www.national.com 18 LM3405 Notes 19 www.national.com LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs Notes THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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