LM25115 Secondary Side Post Regulator Controller General Description Features The LM25115 controller contains all of the features necessary to implement multiple output power converters utilizing the Secondary Side Post Regulation (SSPR) technique. The SSPR technique develops a highly efficient and well regulated auxiliary output from the secondary side switching waveform of an isolated power converter. Regulation of the auxiliary output voltage is achieved by leading edge pulse width modulation (PWM) of the main channel duty cycle. Leading edge modulation is compatible with either current mode or voltage mode control of the main output. The LM25115 drives external high side and low side NMOS power switches configured as a synchronous buck regulator. A current sense amplifier provides overload protection and operates over a wide common mode input range. Additional features include a low dropout (LDO) bias regulator, error amplifier, precision reference, adaptive dead time control of the gate signals and thermal shutdown. n n n n n n n n n n n n n Self-synchronization to main channel output Free-run mode for buck regulation of DC input Leading edge pulse width modulation Voltage-mode control with current injection and input line feed-forward Operates from AC or DC input up to 42V Wide 4.5V to 30V bias supply range Wide 0.75V to 13.5V output range. Top and bottom gate drivers sink 2.5A peak Adaptive gate driver dead-time control Wide bandwidth error amplifier (4MHz) Programmable soft-start Thermal shutdown protection TSSOP-16 or thermally enhanced LLP-16 packages Typical Application Circuit 20172601 FIGURE 1. Simplified Multiple Output Power Converter Utilizing SSPR Technique © 2005 National Semiconductor Corporation DS201726 www.national.com LM25115 Secondary Side Post Regulator Controller November 2005 LM25115 Connection Diagram 20172602 16-Lead TSSOP, LLP See NS Package Numbers MTC16 and SDA16A Ordering Information Package Type NSC Package Drawing Supplied As LM25115MT Ordering Number TSSOP-16 MTC16 92 Units Per Anti-Static Tube LM25115MTX TSSOP-16 MTC16 2500 shipped as Tape & Reel LM25115SD LLP-16 SDA16A Available Soon LM25115SDX LLP-16 SDA16A Available Soon Pin Descriptions Pin Name 1 CS Current Sense amplifier positive input A low inductance current sense resistor is connected between CS and VOUT. Current limiting occurs when the differential voltage between CS and VOUT exceeds 45mV (typical). 2 VOUT Current sense amplifier negative input Connected directly to the output voltage. The current sense amplifier operates over a voltage range from 0V to 13.5V at the VOUT pin. 3 AGND Analog ground Connect directly to the power ground pin (PGND). 4 CO Current limit output For normal current limit operation, connect the CO pin to the COMP pin. Leave this pin open to disable the current limit function. 5 COMP Compensation. Error amplifier output COMP pin pull-up is provided by an internal 300uA current source. 6 FB Feedback. Error amplifier inverting input Connected to the regulated output through the feedback resistor divider and compensation components. The non-inverting input of the error amplifier is internally connected to the SS pin. 7 SS Soft-start control An external capacitor and the equivalent impedance of an internal resistor divider connected to the bandgap voltage reference set the soft-start time. The steady state operating voltage of the SS pin equal to 0.75V (typical). 8 RAMP PWM Ramp signal An external capacitor connected to this pin sets the ramp slope for the voltage mode PWM. The RAMP capacitor is charged with a current that is proportional to current into the SYNC pin. The capacitor is discharged at the end of every cycle by an internal MOSFET. www.national.com Description Application Information 2 Pin Name 9 SYNC 10 PGND 11 LO 12 VCC 13 LM25115 Pin Descriptions (Continued) Description Synchronization input Application Information A low impedance current input pin. The current into this pin sets the RAMP capacitor charge current and the frequency of an internal oscillator that provides a clock for the free-run (DC input) mode . Power Ground Connect directly to the analog ground pin (AGND). Low side gate driver output Connect to the gate of the low side synchronous MOSFET through a short, low inductance path. Output of bias regulator Nominal 7V output from the internal LDO bias regulator. Locally decouple to PGND using a low ESR/ESL capacitor located as close to controller as possible. HS High side MOSFET source connection Connect to negative terminal of the bootstrap capacitor and the source terminal of the high side MOSFET. 14 HO High side gate driver output Connect to the gate of high side MOSFET through a short, low inductance path. 15 HB High side gate driver bootstrap rail Connect to the cathode of the bootstrap diode and the positive terminal of the bootstrap capacitor. The bootstrap capacitor supplies current to charge the high side MOSFET gate and should be placed as close to controller as possible. 16 VBIAS Supply Bias Input Input to the LDO bias regulator and current sense amplifier that powers internal blocks. Input range of VBIAS is 4.5V to 30V. - Exposed Pad Exposed Pad, underside of LLP package Internally bonded to the die substrate. Connect to system (LLP ground for low thermal impedance. Package Only) 3 www.national.com LM25115 Block Diagram 20172603 www.national.com 4 ESD Rating HBM (Note 2) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VBIAS to GND –0.3V to 9V HS to GND –1V to 45V VOUT, CS to GND – 0.3V to 15V All other inputs to GND −0.3V to 7.0V Storage Temperature Range VBIAS supply voltage 5V to 30V VCC supply voltage 5V to 7.5V HS voltage 0V to 42V HB voltage VCC + HS Operating Junction Temperature –55˚C to +150˚C Junction Temperature 2 kV Operating Ratings –0.3V to 32V VCC to GND LM25115 Absolute Maximum Ratings (Note 1) –40˚C to +125˚C +150˚C Typical Operating Conditions Parameter Min Typ Max Units Supply Voltage, VBIAS 4.5 30 V Supply Voltage, VCC 4.5 7 V Supply voltage bypass, CVBIAS 0.1 1 0.1 1 Reference bypass capacitor, CVCC HB-HS bootstrap capacitor 10 µF 0.047 SYNC Current Range (VCC = 4.5V) µF 50 RAMP Saw Tooth Amplitude VOUT regulation voltage (VBIAS min = 3V + VOUT) Electrical Characteristics µF 150 µA 1 1.75 V 0.75 13.5 V Unless otherwise specified, TJ = –40˚C to +125˚C, VBIAS = 12V, No Load on LO or HO. Symbol Parameter Conditions Min Typ Max Units 4 mA 7.15 V VBIAS SUPPLY Ibias VBIAS Supply Current FSYNC = 200kHz VCC LOW DROPOUT BIAS REGULATOR VccReg VCC Regulation VCC open circuit. Outputs not switching VCC Current Limit (Note 4) 6.65 7 40 VCC Under-voltage Lockout Voltage Positive going VCC VCC Under-voltage Hysteresis 4 0.2 mA 4.5 V 0.25 0.3 V 60 77 kΩ SOFT-START SS Source Impedance 43 SS Discharge Impedance Ω 100 ERROR AMPLIFIER and FEEDBACK REFERENCE VREF GBW Vio FB Reference Voltage Measured at FB pin FB Input Bias Current FB = 2V 0.737 0.75 0.763 V 0.2 0.5 µA COMP Source Current 300 µA Open Loop Voltage Gain 60 dB Gain Bandwidth Product 4 Input Offset Voltage -7 COMP Offset Threshold for VHO = high RAMP = CS = VOUT = 0V RAMP Offset Threshold for VHO = high COMP = 1.5V, CS = VOUT = 0V 0 MHz 7 mV 2 V 1.1 V CURRENT SENSE AMPLIFIER Current Sense Amplifier Gain 16 V/V Output DC Offset 1.27 V Amplifier Bandwidth 500 kHz 5 www.national.com LM25115 Electrical Characteristics Unless otherwise specified, TJ = –40˚C to +125˚C, VBIAS = 12V, No Load on LO or HO. (Continued) Symbol Parameter Conditions Min Typ Max Units CURRENT LIMIT VCLneg ILIMIT Amp Transconductance 16 mA / V Overall Transconductance 237 mA / V Positive Current Limit VCL = VCS - VVOUT VOUT = 6V and CO/COMP = 1.5V 37 45 53 mV Positive Current Limit Foldback VCL = VCS - VVOUT VOUT = 0V and CO/COMP = 1.5V 31 38 45 mV Negative Current Limit VOUT = 6V VCL = VCS - VVOUT to cause LO to shutoff -17 mV 2.5 kΩ RAMP GENERATOR SYNC Input Impedance SYNC Threshold End of cycle detection threshold Free Run Mode Peak Threshold RAMP peak voltage with dc current applied to SYNC. Current Mirror Gain Ratio of RAMP charge current to SYNC input current. 15 2.7 Discharge Impedance µA 2.3 V 3.3 A/A Ω 100 LOW SIDE GATE DRIVER VOLL LO Low-state Output Voltage ILO = 100mA 0.2 0.5 V VOHL LO High-state Output Voltage ILO = -100mA, VOHL = VCC -VLO 0.4 0.8 V LO Rise Time CLOAD = 1000pF 15 ns LO Fall Time CLOAD = 1000pF 12 ns IOHL Peak LO Source Current VLO = 0V 2 A IOLL Peak LO Sink Current VLO = 12V 2.5 A HIGH SIDE GATE DRIVER VOLH HO Low-state Output Voltage IHO = 100mA 0.2 0.5 V VOHH HO High-state Output Voltage IHO = -100mA, VOHH = VHB –VHO 0.4 0.8 V HO Rise Time CLOAD = 1000pF 15 ns HO High Side Fall Time CLOAD = 1000pF 12 ns IOHH Peak HO Source Current VHO = 0V 2 A IOLH Peak HO Sink Current VHO = 12V 2.5 A LO Fall to HO Rise Delay CLOAD = 0 70 ns HO Fall to LO Rise Delay CLOAD = 0 50 ns SYNC Fall to HO Fall Delay CLOAD = 0 120 ns SYNC Rise to LO Fall Delay CLOAD = 0 50 ns 165 ˚C 25 ˚C SWITCHING CHARACTERISITCS THERMAL SHUTDOWN TSD Thermal Shutdown Temp. 150 Thermal Shutdown Hysteresis THERMAL RESISTANCE θJA Junction to Ambient MTC Package 125 ˚C/W θJA Junction to Ambient SDA Package 32 ˚C/W Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under which operation of the device is guaranteed. Operating Ratings do not imply guaranteed performance limits. For guaranteed performance limits and associated test conditions, see the Electrical Characteristics tables. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin. Note 3: Min and Max limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlation using Statistical Quality Control (SQC) methods. Limits are used to calculate National’s Average Outgoing Quality Level (AOQL). Note 4: Device thermal limitations may limit usable range. www.national.com 6 LM25115 Typical Performance Characteristics VCC Regulator Start-Up Characteristics, VCC vs VBIAS VCC Load Regulation to Current Limit 20172605 20172604 Current Value (CV) vs Current Limit (VCL) Current Sense Amplifier Gain and Phase vs Frequency 20172606 20172607 Current Error Amplifier Transconductance Overall Current Amplifier Transconductance 20172608 20172609 7 www.national.com LM25115 Typical Performance Characteristics (Continued) Common Mode Output Voltage vs Negative Current Limit (Room Temp) Common Mode Output Voltage vs Positive Current Limit 20172610 www.national.com 20172611 8 The LM25115 controller contains all of the features necessary to implement multiple output power converters utilizing the Secondary Side Post Regulation (SSPR) technique. The SSPR technique develops a highly efficient and well regulated auxiliary output from the secondary side switching waveform of an isolated power converter. Regulation of the auxiliary output voltage is achieved by leading edge pulse width modulation (PWM) of the main channel duty cycle. Leading edge modulation is compatible with either current mode or voltage mode control of the main output. The LM25115 drives external high side and low side NMOS power switches configured as a synchronous buck regulator. A current sense amplifier provides overload protection and operates over a wide common mode input range from 0V to 13.5V. Additional features include a low dropout (LDO) bias regulator, error amplifier, precision reference, adaptive dead time control of the gate driver signals and thermal shutdown. A programmable oscillator provides a PWM clock signal when the LM25115 is powered by a dc input (free-run mode) instead of the phase signal of the main channel converter (SSPR mode). Low Drop-out Bias Regulator (VCC) The LM25115 contains an internal LDO regulator that operates over an input supply range from 4.5V to 30V. The output of the regulator at the VCC pin is nominally regulated at 7V and is internally current limited to 40mA. VCC is the main supply to the internal logic, PWM controller, and gate driver circuits. When power is applied to the VBIAS pin, the regulator is enabled and sources current into an external capacitor connected to the VCC pin. The recommended output capacitor range for the VCC regulator is 0.1uF to 100uF. When the voltage at the VCC pin reaches the VCC undervoltage lockout threshold of 4.25V, the controller is enabled. The controller is disabled if VCC falls below 4.0V (250mV hysteresis). In applications where an appropriate regulated dc bias supply is available, the LM25115 controller can be powered directly through the VCC pin instead of the VBIAS pin. In this configuration, it is recommended that the VCC and the VBIAS pins be connected together such that the external bias voltage is applied to both pins. The allowable VCC range when biased from an external supply is 4.5V to 7V. 20172612 FIGURE 2. Line Feed-forward Diagram The RAMP and SYNC functions illustrated in Figure 2 provide line voltage feed-forward to improve the regulation of the auxiliary output when the input voltage of the main converter changes. Varying the input voltage to the main converter produces proportional variations in amplitude of the phase signal. The main channel PWM controller adjusts the pulse width of the phase signal to maintain constant volt*seconds and a regulated main output as shown in Figure 3. The variation of the phase signal amplitude and duration are reflected in the slope and duty cycle of the RAMP signal of the LM25115 (ISYNC α phase signal amplitude). As a result, the duty cycle of the LM25115 is automatically adjusted to regulate the auxiliary output voltage with virtually no change in the PWM threshold voltage. Transient line regulation is improved because the PWM duty cycle of the auxiliary converter is immediately corrected, independent of the delays of the voltage regulation loop. Synchronization (SYNC) and Feed-forward (RAMP) The pulsing “phase signal” from the main converter synchronizes the PWM ramp and gate drive outputs of the LM25115. The phase signal is the square wave output from the trans- 9 www.national.com LM25115 former secondary winding before rectification (Figure 1). A resistor connected from the phase signal to the low impedance SYNC pin produces a square wave current (ISYNC) as shown in Figure 2. A current comparator at the SYNC input monitors ISYNC relative to an internal 15µA reference. When ISYNC exceeds 15µA, the internal clock signal (CLK) is reset and the capacitor connected to the RAMP begins to charge. The current source that charges the RAMP capacitor is equal to 3 times the ISYNC current. The falling edge of the phase signal sets the CLK signal and discharges the RAMP capacitor until the next rising edge of the phase signal. The RAMP capacitor is discharged to ground by a low impedance (100Ω) n-channel MOSFET. The input impedance at SYNC pin is 2.5kΩ which is normally much less than the external SYNC pin resistance. Detailed Operating Description LM25115 Synchronization (SYNC) and Feed-forward (RAMP) (Continued) 20172613 FIGURE 3. Line Feed-forward Waveforms The recommended SYNC input current range is 50µA to 150µA. The SYNC pin resistor (RSYNC) should be selected to set the SYNC current (ISYNC) to 150µA with the maximum phase signal amplitude, VPHASE(max). This will guarantee that ISYNC stays within the recommended range over a 3:1 change in phase signal amplitude. The SYNC pin resistor is therefore: RSYNC = (VPHASE(max) / 150µA) - 2.5kΩ Once ISYNC has been established by selecting RSYNC, the RAMP signal amplitude may be programmed by selecting the proper RAMP pin capacitor value. The recommended peak amplitude of the RAMP waveform is 1V to 1.75V. The CRAMP capacitor is chosen to provide the desired RAMP amplitude with the nominal phase signal voltage and pulse width. CRAMP = (3 x ISYNC x TON ) / VRAMP Where CRAMP = RAMP pin capacitance ISYNC = SYNC pin current current Error Amplifier and Soft-Start (FB, CO, & COMP, SS) An internal wide bandwidth error amplifier is provided within the LM25115 for voltage feedback to the PWM controller. The amplifier’s inverting input is connected to the FB pin. The output of the auxiliary converter is regulated by connecting a voltage setting resistor divider between the output and the FB pin. Loop compensation networks are connected between the FB pin and the error amplifier output (COMP). The amplifier’s non-inverting input is internally connected to the SS pin. The SS pin is biased at 0.75V by a resistor divider connected to the internal 1.27V bandgap reference. When the VCC voltage is below the UVLO threshold, the SS pin is discharged to ground. When VCC rises and exceeds the positive going UVLO threshold (4.25V), the SS pin is released and allowed to rise. If an external capacitor is connected to the SS pin, it will be charged by the internal resistor divider to gradually increase the non-inverting input of the error amplifier to 0.75V. The equivalent impedance of the SS resistor divider is nominally 60kΩ which determines the charging time constant of the SS capacitor. During startup, the output of the LM25115 converter will follow the exponential equation: VOUT(t) = VOUT(final) x (1 - exp(-t/RSS x CSS)) Where Rss = internal resistance of SS pin (60kΩ) Css = external Soft-Start capacitor VOUT(final) = regulator output set point The initial ∆v / ∆t of the output voltage is VOUT(final) / Rss x Css and VOUT will be within 1% of the final regulation level after 4.6 time constants or when t = 4.6 x Rss x Css. Pull-up current for the error amplifier output is provided by an internal 300µA current source. The PWM threshold signal at the COMP pin can be controlled by either the open drain error amplifier or the open drain current amplifier connected through the CO pin to COMP. Since the internal error amplifier is configured as an open drain output it can be disabled by connecting FB to ground. The current sense amplifier and current limiting function will be described in a later section. TON = corresponding phase signal pulse width VRAMP = desired RAMP amplitude (1V to 1.75V) For example, Main channel output = 3.3V. Phase signal maximum amplitude = 12V. Phase signal frequency = 250kHz • Set ISYNC = 150µA with phase signal at maximum amplitude (12V): ISYNC = 150µA = VPHASE(max) / (RSYNC + 2.5 kΩ) = 12V / (RSYNC + 2.5 kΩ) RSYNC = 12V/150µA - 2.5kΩ = 77.5kΩ • TON = Main channel duty cycle / Phase frequency = (3.3V/12V) / 250kHz = 1.1µs • Assume desired VRAMP = 1.5V • CRAMP = (3 x ISYNC x TON ) / VRAMP = (3 x 150µA x 1.1µs) / 1.5V • CRAMP = 330pF www.national.com 10 Unlike conventional voltage mode controllers, the LM25115 implements leading edge pulse width modulation. A current source equal to 3 times the ISYNC current is used to charge the capacitor connected to the RAMP pin as shown in Figure 4. The ramp signal and the output of the error amplifier (COMP) are combined through a resistor network to produce a voltage ramp with variable dc offset (CRMIX in Figure 4). 20172614 FIGURE 4. Synchronization and Leading Edge Modulation Leading edge modulation of the auxiliary PWM controller is required if the main converter is implemented with peak current mode control. If trailing edge modulation were used, the additional load on the transformer secondary from the auxiliary channel would be drawn only during the first portion of the phase signal pulse. Referring to Figure 5, the turn off the high side MOSFET of the auxiliary regulator would create a non-monotonic negative step in the transformer cur- rent. This negative current step would produce instability in a peak current mode controller. With leading edge modulation, the additional load presented by the auxiliary regulator on the transformer secondary will be present during the latter portion of the phase signal. This positive step in the phase signal current can be accommodated by a peak current mode controller without instability. 20172620 FIGURE 5. Leading versus Trailing Edge Modulation 11 www.national.com LM25115 The high side MOSFET which drives the HS pin is held in the off state at the beginning of the phase signal. When the voltage of CRMIX exceeds the internal threshold voltage CV, the PWM comparator turns on the high side MOSFET. The HS pin rises and the MOSFET delivers current from the main converter phase signal to the output of the auxiliary regulator. The PWM cycle ends when the phase signal falls and power is no longer supplied to the drain of the high side MOSFET. Leading Edge Pulse Width Modulation LM25115 The design of the voltage feedback path through the error amp begins with the selection of R1 and R2 in Figure 7 to set the regulated output voltage. The steady state output voltage after soft-start is determined by the following equation: VOUT(final) = 0.75V x (1+R1/R2) Voltage Mode Control with Current Injection The LM25115 controller uniquely combines elements and benefits of current mode control in a voltage mode PWM controller. The current sense amplifier shown in Figure 6 monitors the inductor current as it flows through a sense resistor connected between CS and VOUT. The voltage gain of the sense amplifier is nominally equal to 16. The current sense output signal is shifted by 1.27V to produce the internal CV reference signal. The CV signal is applied to the negative input of the PWM comparator and compared to CRMIX as illustrated in Figure 4. Thus the PWM threshold of the voltage mode controller (CV) varies with the instantaneous inductor current. Insure that the Vbias voltage is at least 3V above the regulated output voltage (VOUT). Injecting a signal proportional to the instantaneous inductor current into a voltage mode controller improves the control loop stability and bandwidth. This current injection eliminates the lead R-C lead network in the feedback path that is normally required with voltage mode control (see Figure 7). Eliminating the lead network not only simplifies the compensation, but also reduces sensitivity to output noise that could pass through the lead network to the error amplifier. The parallel impedance of the R1, R2 resistor divider should be approximately 2kΩ (between 0.5kΩ and 5kΩ). Lower resistance values may not be properly driven by the error amplifier output and higher feedback resistances can introduce noise sensitivity. The next step in the design process is selection of R3, which sets the ac gain of the error amplifier. The ac gain is given by the following equation and should be set to a value less than 30. GAIN(ac) = R3/(R1|| R2) < 30 The capacitor C1 is connected in series with R3 to increase the dc gain of the voltage regulation loop and improve output voltage accuracy. The corner frequency set by R3 x C1 should be less than 1/10th of the cross-over frequency of the overall converter such that capacitor C1 does not add phase lag at the crossover frequency. Capacitor C2 is added to reduce the noise in the voltage control loop. The value of C2 should be less than 500pF and C2 may not be necessary with very careful PC board layout. 20172615 FIGURE 6. Current Sensing and Limiting www.national.com 12 LM25115 Voltage Mode Control with Current Injection (Continued) 20172616 FIGURE 7. Voltage Sensing and Feedback Current Limiting (CS, CO and VOUT) Negative Current Limit When inductor current flows from the regulator output through the low side MOSFET, the input to the current sense comparator becomes negative. The intent of the negative current comparator is to protect the low-side MOSFET from excessive currents. Negative current can lead to large negative voltage spikes on the output at turn off which can damage circuitry powered by the output. The negative current comparator threshold is sufficiently negative to allow inductor current to reverse at no load or light load conditions. It is not intended to support discontinuous conduction mode with diode emulation by the low-side MOSFET. The negative current comparator illustrated in Figure 6 monitors the CV signal and compares this signal to a fixed 1V threshold. This corresponds to a negative VCL voltage between CS and VOUT of -17mV. The negative current limit comparator turns off the low-side MOSFET for the remainder of the cycle when the VCL input falls below this threshold. Current limiting is implemented through the current sense amplifier as illustrated in Figure 6. The current sense amplifier monitors the inductor current that flows through a sense resistor connected between CS and VOUT. The voltage gain of the current sense amplifier is nominally equal to 16. The output of current sense amplifier is level shifted by 1.27V to produce the internal CV reference signal. The CV signal drives a current limit amplifier with nominal transconductance of 16mA/V. The current limit amplifier has an open drain (sink only) output stage and its output pin, CO is typically connected to the COMP pin. During normal operation, the voltage error amplifier controls the COMP pin voltage which adjusts the PWM duty cycle by varying the internal CRMIX level (Figure 4). However, when the current sense input voltage VCL exceeds 45mV, the current limit amplifier pulls down on COMP through the CO pin. Pulling COMP low reduces the CRMIX signal below the CV signal level. When CRMIX does not exceed the CV signal, the PWM comparator inhibits output pulses until the CRMIX signal increases to a normal operating level. A current limit fold-back feature is provided by the LM25115 to reduce the peak output current delivered to a shorted load. When the common mode input voltage to the current sense amplifier (CS and VOUT pins) falls below 2V, the current limit threshold is reduced from the normal level. At common mode voltages > 2V, the current limit threshold is nominally 45mV. When VOUT is reduced to 0V the current limit threshold drops to 36mV to reduce stress on the inductor and power MOSFETs. Gate Drivers Outputs (HO & LO) The LM25115 provides two gate driver outputs, the floating high-side gate driver HO and the synchronous rectifier lowside driver LO. The low-side driver is powered directly by the VCC regulator. The high-side gate driver is powered from a bootstrap capacitor connected between HB and HS. An external diode connected between VCC and HB charges the bootstrap capacitor when the HS is low. When the high-side MOSFET is turned on, HB rises with HS to a peak voltage equal to VCC + VHS - VD where VD is the forward drop of the external bootstrap diode. Both output drivers have adaptive dead-time control to avoid shoot through currents. The adaptive dead-time control circuit monitors the state of each driver to ensure that the opposing MOSFET is turned off before the other is turned on. The HB and VCC capacitors should be placed close to the pins of the LM25115 to mini13 www.national.com LM25115 Gate Drivers Outputs (HO & LO) before beginning a new PWM cycle. The 300ns reset time of the RAMP pin sets the minimum off time of the PWM controller in the free-run mode. The internal clock frequency in the free-run mode is set by the synchronization current, ramp capacitor, free-run peak threshold, and 300ns deadtime. FCLK ) 1 / ((CRAMP x 2.25V) / (ISYNC x 3) + 300ns) (Continued) mize voltage transients due to parasitic inductances and the high peak output currents of the drivers. The recommended range of the HB capacitor is 0.047µF to 0.22µF. Both drivers are controlled by the PWM logic signal from the PWM latch. When the phase signal is low, the outputs are held in the reset state with the low-side MOSFET on and the high-side MOSFET off. When the phase signal switches to the high state, the PWM latch reset signal is de-asserted. The high-side MOSFET remains off until the PWM latch is set by the PWM comparator (CRMIX > CV as shown in Figure 4). When the PWM latch is set, the LO driver turns off the low-side MOSFET and the HO driver turns on the highside MOSFET. The high-side pulse is terminated when the phase signal falls and the SYNC input comparator resets the PWM latch. Note that the VCC supply can be used as the dc bias to produce ISYNC. Note that the input voltage feedforward is no longer functional in this operating mode, so the loop gain will vary as a function of Vin. The LM25115 controls the buck power stage with leading edge pulse width modulaton to hold off the high-side driver until the necessary volt*seconds is established for regulation. Other features described for the secondary side post regulator apply in the free run mode operation. They include voltage mode control with current injection, positive and negative current limit, programmable soft-start, adaptive delays for outputs, and thermal protection. Free-Run Mode Thermal Protection The LM25115 can be operated as a conventional synchronous buck controller with a dc input supply instead of the square wave phase signal. In the dc or free-run mode, the LM25115 PWM controller synchronizes to an internal clock signal instead of the phase signal pulses. The clock frequency in the free-run mode is programmed by the SYNC pin resistor and RAMP pin capacitor. Connecting a resistor between a dc bias supply and the SYNC pin produces a current ISYNC which controls the charging current of the RAMP pin capacitor . The RAMP capacitor is charged until its voltage reaches the free-run mode peak threshold of 2.25V. The RAMP capacitor is then discharged for 300ns www.national.com Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event the maximum junction temperature limit is exceeded. When activated, typically at 165 degrees Celsius, the controller is forced into a low power standby state with the output drivers and the bias regulator disabled. The device will restart when the junction temperature falls below the thermal shutdown hysteresis, which is typically 25 degrees. The thermal protection feature is provided to prevent catastrophic failures from accidental device overheating. 14 Application Circuit LM25115 Secondary Side Post Regulator (Inputs from LM5025 Forward Active Clamp Converter, 36V to 78V) 20172617 LM25115 15 www.national.com LM25115 Physical Dimensions inches (millimeters) unless otherwise noted TSSOP-16 Outline Drawing NS Package Number MTC16 LLP-16 Outline Drawing NS Package Number SDA16A www.national.com 16 LM25115 Secondary Side Post Regulator Controller Notes National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. 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