AD AD7711

a
FEATURES
Charge Balancing ADC
24 Bits No Missing Codes
ⴞ0.0015% Nonlinearity
Two-Channel Programmable Gain Front End
Gains from 1 to 128
One Differential Input
One Single-Ended Input
Low-Pass Filter with Programmable Filter Cutoffs
Ability to Read/Write Calibration Coefficients
RTD Excitation Current Sources
Bidirectional Microcontroller Serial Interface
Internal/External Reference Option
Single or Dual Supply Operation
Low Power (25 mW typ) with Power-Down Mode
(7 mW typ)
APPLICATIONS
RTD Transducers
Process Control
Smart Transmitters
Portable Industrial Instruments
GENERAL DESCRIPTION
The AD7711 is a complete analog front end for low frequency
measurement applications. The device accepts low level signals
directly from a transducer and outputs a serial digital word. It
employs a sigma-delta conversion technique to realize up to
24 bits of no missing codes performance. The input signal is
applied to a proprietary programmable gain front end based
around an analog modulator. The modulator output is processed by an on-chip digital filter. The first notch of this digital
filter can be programmed via the on-chip control register allowing adjustment of the filter cutoff and settling time.
The part features one differential analog input and one single
ended analog input as well as a differential reference input.
Normally, one of the input channels will be used as the main
channel with the second channel used as an auxiliary input to
periodically measure a second voltage. It can be operated from a
single supply (by tying the VSS pin to AGND) provided that the
input signals on the analog inputs are more positive than
–30 mV. By taking the VSS pin negative, the part can convert
signals down to –VREF on its inputs. The part provides two
current sources that can be used to provide excitation in threewire and four-wire RTD configurations. The AD7711 thus
performs all signal conditioning and conversion for a single or
dual channel system.
The AD7711 is ideal for use in smart, microcontroller based
systems. Gain settings, signal polarity, input channel selection
LC2MOS Signal Conditioning ADC
with RTD Excitation Currents
AD7711*
FUNCTIONAL BLOCK DIAGRAM
AVDD
REF
REF
IN (–) IN (+)
DVDD
VBIAS
REF OUT
AVDD
2.5V REFERENCE
4.5mA
CHARGE-BALANCING A/D
CONVERTER
AIN1(+)
AUTO-ZEROED
M
U
X
AIN1(–)
PGA
MODULATOR
DIGITAL
FILTER
SYNC
A = 1 – 128
AIN2
200mA
CLOCK
GENERATION
AVDD
RTD1
MCLK
IN
MCLK
OUT
SERIAL INTERFACE
200mA
CONTROL
REGISTER
RTD2
OUTPUT
REGISTER
AD7711
AGND DGND
VSS
RFS TFS MODE SDATA SCLK DRDY A0
and RTD current control can be configured in software using
the bidirectional serial port. The AD7711 contains selfcalibration, system calibration and background calibration
options and also allows the user to read and write the on-chip
calibration registers.
CMOS construction ensures low power dissipation, and a software programmable power-down mode reduces the standby
power consumption to only 7 mW typical. The part is available
in a 24-lead, 0.3 inch wide, plastic and hermetic dual-in-line
package (DIP) as well as a 24-lead small outline (SOIC)
package.
PRODUCT HIGHLIGHTS
1. The programmable gain front end allows the AD7711 to
accept input signals directly from an RTD transducer,
removing a considerable amount of signal conditioning.
On-chip current sources provide excitation for three-wire and
four-wire RTD configurations.
2. No Missing Codes ensure true, usable, 23-bit dynamic range
coupled with excellent ± 0.0015% accuracy. The effects of
temperature drift are eliminated by on-chip self-calibration,
which removes zero-scale and full-scale errors.
3. The AD7711 is ideal for microcontroller or DSP processor
applications with an on-chip control register which allows
control over filter cutoff, input gain, channel selection, signal
polarity, RTD current control and calibration modes.
REV. F
4. The AD7711 allows the user to read and to write the on-chip
calibration registers. This means that the microcontroller has
much greater control over the calibration procedure.
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1998
*Protected by U.S. Patent No. 5,134,401.
(AV = +5␣ V ⴞ 5%; DV = +5␣ V ⴞ 5%; V = 0␣ V or –5 V ⴞ 5%; REF IN(+) =
AD7711–SPECIFICATIONS
+2.5␣ V; REF␣ IN(–) = AGND; MCLK IN = 10␣ MHz unless otherwise stated. All specifications T to T unless otherwise noted.)
DD
DD
SS
MIN
Parameter
STATIC PERFORMANCE
No Missing Codes
Output Noise
Integral Nonlinearity @ +25°C
TMIN to T MAX
Positive Full-Scale Error2, 3
Full-Scale Drift5
Unipolar Offset Error2
Unipolar Offset Drift5
Bipolar Zero Error2
Bipolar Zero Drift5
Gain Drift
Bipolar Negative Full-Scale Error2 @ +25°C
TMIN to T MAX
Bipolar Negative Full-Scale Drift5
ANALOG INPUTS/REFERENCE INPUTS
Normal-Mode 50 Hz Rejection6
Normal-Mode 60 Hz Rejection6
DC Input Leakage Current @ +25°C6
TMIN to T MAX
Sampling Capacitance6
AIN1/REF IN
Common-Mode Rejection (CMR)
Common-Mode 50 Hz Rejection6
Common-Mode 60 Hz Rejection6
Common-Mode Voltage Range7
Analog Inputs8
Input Voltage Range9
Input Sampling Rate, fS
AIN2 Offset Error
AIN2 Offset Drift
Reference Inputs
REF IN(+) – REF IN(–) Voltage11
Input Sampling Rate, fS
REFERENCE OUTPUT
Output Voltage
Initial Tolerance @ +25°C
Drift
Output Noise
Line Regulation (AVDD)
Load Regulation
External Current
MAX
A, S Versions1
Units
Conditions/Comments
24
22
18
15
12
See Tables I & II
± 0.0015
± 0.003
See Note 4
1
0.3
See Note 4
0.5
0.25
See Note 4
0.5
0.25
2
± 0.003
± 0.006
1
0.3
Bits min
Bits min
Bits min
Bits min
Bits min
µV/°C typ
µV/°C typ
Guaranteed by Design. For Filter Notches ≤ 60 Hz
For Filter Notch = 100 Hz
For Filter Notch = 250 Hz
For Filter Notch = 500 Hz
For Filter Notch = 1 kHz
Depends on Filter Cutoffs and Selected Gain
Filter Notches ≤ 60 Hz
Typically ± 0.0003%
Excluding Reference
Excluding Reference. For Gains of 1, 2
Excluding Reference. For Gains of 4, 8, 16, 32, 64, 128
µV/°C typ
µV/°C typ
For Gains of 1, 2
For Gains of 4, 8, 16, 32, 64, 128
µV/°C typ
µV/°C typ
ppm/°C typ
% FSR max
% FSR max
µV/°C typ
µV/°C typ
For Gains of 1, 2
For Gains of 4, 8, 16, 32, 64, 128
% FSR max
% FSR max
Excluding Reference
Typically ± 0.0006%
Excluding Reference. For Gains of 1, 2
Excluding Reference. For Gains of 4, 8, 16, 32, 64, 128
100
100
10
1
20
dB min
dB min
pA max
nA max
pF max
For Filter Notches of 10, 25, 50 Hz, ± 0.02 × fNOTCH
For Filter Notches of 10, 30, 60 Hz, ± 0.02 × fNOTCH
100
150
150
VSS to AVDD
dB min
dB min
dB min
V min to V max
At DC
For Filter Notches of 10, 25, 50 Hz, ± 0.02 × fNOTCH
For Filter Notches of 10, 30, 60 Hz, ± 0.02 × fNOTCH
0 to +VREF10
± VREF
See Table III
2.5
1.5
max
max
mV max
µV/°C typ
Removed by System Calibrations but not by Self-Calibration
+2.5 to +5
V min to V max
For Specified Performance. Part Is Functional with
Lower V REF Voltages
For Normal Operation. Depends on Gain Selected
Unipolar Input Range (B/U Bit of Control Register = 1)
Bipolar Input Range (B/U Bit of Control Register = 0)
fCLK IN/256
2.5
±1
20
30
1
1.5
1
V nom
% max
ppm/°C typ
µV typ
mV/V max
mV/mA max
mA max
pk-pk Noise. 0.1 Hz to 10 Hz Bandwidth
Maximum Load Current 1 mA
NOTES
1
Temperature range is as follows: A Version = – 40°C to +85°C; S Version = –55°C to +125°C. See also Note 16.
2
Applies after calibration at the temperature of interest.
3
Positive full-scale error applies to both unipolar and bipolar input ranges.
4
These errors will be of the order of the output noise of the part as shown in Table I after system calibration. These errors will be 20 µV typical after self-calibration or
background calibration.
5
Recalibration at any temperature or use of the background calibration mode will remove these drift errors.
6
These numbers are guaranteed by design and/or characterization.
7
This common-mode voltage range is allowed, provided that the input voltage on AIN(+) and AIN(–) does not exceed AV DD + 30 mV and V SS – 30 mV.
8
The analog inputs present a very high impedance dynamic load which varies with clock frequency and input sample rate. The maximum recommended source
resistance depends on the selected gain (see Tables IV and V).
9
The analog input voltage range on the AIN1(+) input is given here with respect to the voltage on the AIN1(–) input. The input voltage range on the AIN2 input is
with respect to AGND. The absolute voltage on the analog inputs should not go more positive than A VDD + 30 mV or go more negative than VSS – 30 mV.
10
VREF = REF IN(+) – REF IN(–).
11
The reference input voltage range may be restricted by the input voltage range requirement on the V BIAS input.
–2–
REV. F
AD7711
Parameter
A, S Versions1
Units
AVDD – 0.85 × VREF
or AVDD – 3.5
V max
Conditions/Comments
12
VBIAS INPUT
Input Voltage Range
VBIAS Rejection
or AVDD – 2.1
VSS + 0.85 × VREF
or VSS + 3
V max
or VSS + 2.1
65 to 85
V min
dB typ
± 10
µA max
0.8
2.0
V max
V min
0.8
3.5
V max
V min
V min
LOGIC INPUTS
Input Current
All Inputs except MCLK IN
VINL, Input Low Voltage
VINH, Input High Voltage
MCLK IN Only
VINL, Input Low Voltage
VINH, Input High Voltage
LOGIC OUTPUTS
VOL, Output Low Voltage
VOH, Output High Voltage
Floating State Leakage Current
Floating State Output Capacitance13
TRANSDUCER BURNOUT
Current
Initial Tolerance @ +25°C
Drift
0.4
4.0
± 10
9
V max
V min
µA max
pF typ
4.5
± 10
0.1
µA nom
% typ
%/°C typ
RTD EXCITATION CURRENTS (RTD1, RTD2)
Output Current
Initial Tolerance @ +25°C
Drift
Initial Matching @ +25°C
Drift Matching
Line Regulation (AVDD)
Load Regulation
Output Compliance
200
± 20
20
±1
3
200
200
AVDD – 2
µA nom
% max
ppm/°C typ
% max
ppm/°C typ
nA/V max
nA/V max
V max
(1.05 × VREF)/GAIN
–(1.05 × VREF)/GAIN
–(1.05 × VREF)/GAIN
0.8 × VREF/GAIN
(2.1 × VREF)/GAIN
V max
V max
V max
V min
V max
SYSTEM CALIBRATION
Positive Full-Scale Calibration Limit14
Negative Full-Scale Calibration Limit14
Offset Calibration Limit15
Input Span15
See VBIAS Input Section
Whichever Is Smaller; +5 V/–5 V or +10 V/0 V
Nominal AVDD/VSS
Whichever Is Smaller; +5 V/0 V Nominal AVDD/VSS
See VBIAS Input Section
Whichever Is Greater; +5 V/–5 V or +10 V/0 V
Nominal AVDD/VSS
Whichever Is Greater; +5 V/0 V Nominal AVDD/VSS
Increasing with Gain
ISINK = 1.6 mA
ISOURCE = 100 µA
Matching Between RTD1 and RTD2 Currents
Matching Between RTD1 and RTD2 Current Drift
AV DD = +5 V
GAIN Is the Selected PGA Gain (Between 1 and 128)
GAIN Is the Selected PGA Gain (Between 1 and 128)
GAIN Is the Selected PGA Gain (Between 1 and 128)
GAIN Is the Selected PGA Gain (Between 1 and 128)
GAIN Is the Selected PGA Gain (Between 1 and 128)
NOTES
12
The AD7711 is tested with the following V BIAS voltages. With AVDD = +5 V and V SS = 0 V, VBIAS = +2.5 V; with AV DD = +10 V and VSS = 0 V, VBIAS = +5 V and
with AVDD = +5 V and V SS = –5 V, VBIAS = 0 V.
13
Guaranteed by design, not production tested.
14
After calibration, if the analog input exceeds positive full scale, the converter will output all 1s. If the analog input is less than negative full scale, then the device will
output all 0s.
15
These calibration and span limits apply provided the absolute voltage on the analog inputs does not exceed AV DD + 30 mV or go more negative than V SS – 30 mV.
The offset calibration limit applies to both the unipolar zero point and the bipolar zero point.
REV. F
–3–
AD7711–SPECIFICATIONS
Parameter
POWER REQUIREMENTS
Power Supply Voltages
AVDD Voltage16
DVDD Voltage17
AVDD – VSS Voltage
Power Supply Currents
AVDD Current
DVDD Current
VSS Current
Power Supply Rejection18
Positive Supply (AVDD and DVDD )
Negative Supply (VSS)
Power Dissipation
Normal Mode
Standby (Power-Down) Dissipation
A, S Versions1
Units
Conditions/Comments
+5 to +10
+5
+10.5
V nom
V nom
V max
± 5% for Specified Performance
± 5% for Specified Performance
For Specified Performance
4
4.5
1.5
mA max
mA max
mA max
See Note 19
90
dB typ
dB typ
45
52.5
15
mW max
mW max
mW max
VSS = –5 V
Rejection w.r.t. AGND; Assumes VBIAS Is Fixed
AV DD = DVDD = +5 V, VSS = 0 V; Typically 25 mW
AV DD = DVDD = +5␣ V, VSS = –5 V; Typically 30 mW
AV DD = DVDD = +5␣ V, VSS = 0 V or –5 V; Typically 7 mW
NOTES
16
The AD7711 is specified with a 10 MHz clock for AV DD voltages of +5 V ± 5%. It is specified with an 8 MHz clock for AV DD voltages greater than 5.25 V and less
than 10.5 V.
17
The ±5% tolerance on the DV DD input is allowed provided that DVDD does not exceed AV DD by more than 0.3 V.
18
Measured at dc and applies in the selected passband. PSRR at 50 Hz will exceed 120 dB with filter notches of 10 Hz, 25 Hz or 50 Hz. PSRR at 60 Hz will exceed
120 dB with filter notches of 10 Hz, 30 Hz or 60 Hz.
19
PSRR depends on gain: Gain of 1 = 70 dB typ; Gain of 2: 75 dB typ; Gain of 4 = 80 dB typ; Gains of 8 to 128 = 85 dB typ. These numbers can be improved (to
95 dB typ) by deriving the VBIAS voltage (via Zener diode or reference) from the AV DD supply.
Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS*
(TA = +25°C, unless otherwise noted)
AVDD to DVDD . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V
AVDD to VSS . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V
AVDD to AGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V
AVDD to DGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +12 V
DVDD to AGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +6 V
DVDD to DGND . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +6 V
VSS to AGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –6 V
VSS to DGND . . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V to –6 V
Analog Input Voltage to AGND
. . . . . . . . . . . . . . . . . . . . . . . . . VSS – 0.3 V to AVDD + 0.3 V
Reference Input Voltage to AGND
. . . . . . . . . . . . . . . . . . . . . . . . . VSS – 0.3 V to AVDD + 0.3 V
REF OUT to AGND . . . . . . . . . . . . . . . . . . . . –0.3 V to AVDD
Digital Input Voltage to DGND . . . . . –0.3 V to AVDD + 0.3 V
Digital Output Voltage to DGND . . . –0.3 V to DVDD + 0.3 V
Operating Temperature Range
Commercial (A Version) . . . . . . . . . . . . . . . . –40°C to +85°C
Extended (S Version) . . . . . . . . . . . . . . . . . –55°C to +125°C
Storage Temperature Range . . . . . . . . . . . . . –65°C to +150°C
Lead Temperature (Soldering, 10 secs) . . . . . . . . . . . . +300°C
Power Dissipation (Any Package) to +75°C . . . . . . . . 450 mW
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those listed in the operational
sections of the specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
ORDERING GUIDE
Model
Temperature Range
Package Option*
AD7711AN
AD7711AR
AD7711AQ
AD7711SQ
EVAL-AD7711EB
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–55°C to +125°C
Evaluation Board
N-24
R-24
Q-24
Q-24
*N = Plastic DIP, Q = Cerdip; R = SOIC.
CAUTION
ESD (electrostatic discharge) sensitive device. The digital control inputs are diode protected;
however, permanent damage may occur on unconnected devices subject to high energy electrostatic fields. Unused devices must be stored in conductive foam or shunts. The protective foam
should be discharged to the destination socket before devices are inserted.
–4–
WARNING!
ESD SENSITIVE DEVICE
REV. F
AD7711
3
1, 2 (DVDD = +5␣ V ⴞ 5%; AV DD = +5␣ V or +10 V ⴞ 5%; VSS = 0 V or –5 V ⴞ 10%; AGND = DGND =
TIMING CHARACTERISTICS
0 V; fCLK IN = 10␣ MHz; Input Logic 0 = 0 V, Logic 1 = DVDD, unless otherwise noted.)
Parameter
Limit at TMIN, T MAX
(A, S Versions)
Units
Conditions/Comments
fCLK IN4, 5
400
kHz min
Master Clock Frequency: Crystal Oscillator or Externally
Supplied for Specified Performance
10
0.4 × tCLK IN
0.4 × tCLK IN
50
50
1000
MHz max
ns min
ns min
ns max
ns max
ns min
0
0
2 × tCLK IN
0
4 × tCLK IN + 20
4 × tCLK IN + 20
tCLK IN/2
tCLK IN/2 + 30
tCLK IN/2
3 × tCLK IN/2
50
0
4 × tCLK IN + 20
4 × tCLK IN
0
10
ns min
ns min
ns min
ns min
ns max
ns max
ns min
ns max
ns nom
ns nom
ns min
ns min
ns max
ns min
ns min
ns min
tCLK IN LO
tCLK IN HI
tr 6
tf 6
t1
Self-Clocking Mode
t2
t3
t4
t5
t6
t77
t87
t9
t10
t14
t15
t16
t17
t18
t19
REV. F
Master Clock Input Low Time; tCLK IN = 1/fCLK IN
Master Clock Input High Time
Digital Output Rise Time. Typically 20 ns
Digital Output Fall Time. Typically 20 ns
SYNC Pulsewidth
DRDY to RFS Setup Time
DRDY to RFS Hold Time
A0 to RFS Setup Time
A0 to RFS Hold Time
RFS Low to SCLK Falling Edge
Data Access Time (RFS Low to Data Valid)
SCLK Falling Edge to Data Valid Delay
SCLK High Pulsewidth
SCLK Low Pulsewidth
A0 to TFS Setup Time
A0 to TFS Hold Time
TFS to SCLK Falling Edge Delay Time
TFS to SCLK Falling Edge Hold Time
Data Valid to SCLK Setup Time
Data Valid to SCLK Hold Time
–5–
2
AD7711
Parameter
External Clocking Mode
fSCLK
t20
t21
t22
t23
t247
t257
t26
t27
t28
t298
t30
t318
t32
t33
t34
t35
t36
Limit at TMIN, T MAX
(A, S Versions)
Units
Conditions/Comments
fCLK IN/5
0
0
2 × tCLK IN
0
4 × tCLK IN
10
2 × tCLK IN + 20
2 × tCLK IN
2 × tCLK IN
tCLK IN + 10
10
tCLK IN + 10
10
5 × tCLK IN/2 + 50
0
0
4 × tCLK IN
2 × tCLK IN – SCLK High
30
MHz max
ns min
ns min
ns min
ns min
ns max
ns min
ns max
ns min
ns min
ns max
ns min
ns max
ns min
ns max
ns min
ns min
ns min
ns min
ns min
Serial Clock Input Frequency
DRDY to RFS Setup Time
DRDY to RFS Hold Time
A0 to RFS Setup Time
A0 to RFS Hold Time
Data Access Time (RFS Low to Data Valid)
SCLK Falling Edge to Data Valid Delay
SCLK High Pulsewidth
SCLK Low Pulsewidth
SCLK Falling Edge to DRDY High
SCLK to Data Valid Hold Time
RFS/TFS to SCLK Falling Edge Hold Time
RFS to Data Valid Hold Time
A0 to TFS Setup Time
A0 to TFS Hold Time
SCLK Falling Edge to TFS Hold Time
Data Valid to SCLK Setup Time
Data Valid to SCLK Hold Time
NOTES
1
Guaranteed by design, not production tested. All input signals are specified with tr = tf = 5 ns (10% to 90% of 5 V) and timed from a voltage level of 1.6 V.
2
See Figures 10 to 13.
3
The AD7711 is specified with a 10 MHz clock for AV DD voltages of +5 V ± 5%. It is specified with an 8 MHz clock for AV DD voltages greater than 5.25 V and less
than 10.5 V.
4
CLK IN duty cycle range is 45% to 55%. CLK IN must be supplied whenever the AD7711 is not in STANDBY mode. If no clock is present in this case, the device
can draw higher current than specified and possibly become uncalibrated.
5
The AD7711 is production tested with f CLK IN at 10 MHz (8 MHz for AV DD > +5.25 V). It is guaranteed by characterization to operate at 400 kHz.
6
Specified using 10% and 90% points on waveform of interest.
7
These numbers are measured with the load circuit of Figure 1 and defined as the time required for the output to cross 0.8 V or 2.4 V.
8
These numbers are derived from the measured time taken by the data output to change 0.5 V when loaded with the circuit of Figure 1. The measured number is then
extrapolated back to remove effects of charging or discharging the 100 pF capacitor. This means that the times quoted in the timing characteristics are the true bus
relinquish times of the part and, as such, are independent of external bus loading capacitances.
Specifications subject to change without notice.
PIN CONFIGURATION
DIP AND SOIC
1.6mA
TO OUTPUT
PIN
+2.1V
100pF
200mA
Figure 1. Load Circuit for Access Time and Bus Relinquish
Time
–6–
SCLK
1
24
DGND
MCLK IN
2
23
DVDD
MCLK OUT
3
22
SDATA
A0
4
21
DRDY
SYNC
5
20
RFS
MODE
6
AD7711
AIN1(+)
TOP VIEW 19 TFS
7 (Not to Scale) 18 AGND
AIN1(–)
8
17
AIN2
RTD1
9
16
REF OUT
RTD2 10
15
REF IN(+)
VSS 11
14
REF IN(–)
AVDD 12
13
VBIAS
REV. F
AD7711
PIN FUNCTION DESCRIPTION
Pin Mnemonic
1
SCLK
Function
Serial Clock. Logic Input/Output depending on the status of the MODE pin. When MODE is high, the
device is in its self-clocking mode and the SCLK pin provides a serial clock output. This SCLK becomes
active when RFS or TFS goes low and it goes high impedance when either RFS or TFS returns high or when
the device has completed transmission of an output word. When MODE is low, the device is in its external
clocking mode and the SCLK pin acts as an input. This input serial clock can be a continuous clock with all
data transmitted in a continuous train of pulses. Alternatively, it can be a noncontinuous clock with the
information being transmitted to the AD7711 in smaller batches of data.
2
MCLK IN
Master Clock signal for the device. This can be provided in the form of a crystal or external clock. A crystal can
be tied across the MCLK IN and MCLK OUT pins. Alternatively, the MCLK IN pin can be driven with a
CMOS-compatible clock and MCLK OUT left unconnected. The clock input frequency is nominally 10 MHz.
3
MCLK OUT
When the master clock for the device is a crystal, the crystal is connected between MCLK IN and MCLK OUT.
4
A0
Address Input. With this input low, reading and writing to the device is to the control register. With this input
high, access is to either the data register or the calibration registers.
5
SYNC
Logic Input which allows for synchronization of the digital filters when using a number of AD7711s. It resets
the nodes of the digital filter.
6
MODE
Logic Input. When this pin is high, the device is in its self-clocking mode; with this pin low, the device is in its
external clocking mode.
7
AIN1(+)
Analog Input Channel 1. Positive input of the programmable gain differential analog input. The AIN1(+) input
is connected to an output current source which can be used to check that an external transducer has burned out
or gone open circuit. This output current source can be turned on/off via the control register.
8
AIN1(–)
Analog Input Channel 1. Negative input of the programmable gain differential analog input.
9
RTD1
Constant Current Output. A nominal 200 µA constant current is provided at this pin, and this can be used
as the excitation current for RTDs. This current can be turned on or off via the control register.
10
RTD2
Constant Current Output. A nominal 200 µA constant current is provided at this pin, and this can be used
as the excitation current for RTDs. This current can be turned on or off via the control register. This
second current can be used to eliminate lead resistance errors in three-wire RTD configurations.
11
VSS
Analog Negative Supply, 0 V to –5 V. Tied to AGND for single supply operation. The input voltage on AIN1
or AIN2 should not go > 30 mV negative w.r.t. VSS for correct operation of the device.
12
AVDD
Analog Positive Supply Voltage, +5 V to +10 V.
13
VBIAS
Input Bias Voltage. This input voltage should be set such that VBIAS + 0.85 × VREF < AVDD and VBIAS – 0.85
× VREF > VSS where VREF is REF IN(+) – REF IN(–). Ideally, this should be tied halfway between AVDD
and VSS. Thus with AVDD = +5 V and VSS = 0 V, it can be tied to REF OUT; with AVDD = +5 V and VSS =
–5 V, it can be tied to AGND, while with AVDD = +10 V, it can be tied to +5 V.
14
REF IN(–)
Reference Input. The REF IN(–) can lie anywhere between AVDD and VSS provided REF IN(+) is greater
than REF IN(–).
15
REF IN(+)
Reference Input. The reference input is differential providing that REF IN(+) is greater than REF IN(–).
REF IN(+) can lie anywhere between AVDD and VSS.
16
REF OUT
Reference Output. The internal +2.5 V reference is provided at this pin. This is a single-ended output
which is referred to AGND. It is a buffered output which is capable of providing 1 mA to an external load.
17
AIN2
Analog Input Channel 2. Single-ended programmable gain analog input.
18
AGND
Ground reference point for analog circuitry.
19
TFS
Transmit Frame Synchronization. Active low logic input used to write serial data to the device with serial
data expected after the falling edge of this pulse. In the self-clocking mode, the serial clock becomes active
after TFS goes low. During a write operation to the AD7711, the SDATA line should not return to high
impedance until after TFS returns high.
REV. F
–7–
2
AD7711
Pin Mnemonic
Function
20
RFS
Receive Frame Synchronization. Active low logic input used to access serial data from the device. In the
self-clocking mode, the SCLK and SDATA lines both become active after RFS goes low. In the external
clocking mode, the SDATA line becomes active after RFS goes low.
21
DRDY
Logic output. A falling edge indicates that a new output word is available for transmission. The DRDY pin
will return high upon completion of transmission of a full output word. DRDY is also used to indicate
when the AD7711 has completed its on-chip calibration sequence.
22
SDATA
Serial Data. Input/Output with serial data being written to either the control register or the calibration
registers and serial data being accessed from the control register, calibration registers or the data register.
During an output data read operation, serial data becomes active after RFS goes low (provided DRDY is
low). During a write operation, valid serial data is expected on the rising edges of SCLK when TFS is low.
The output data coding is natural binary for unipolar inputs and offset binary for bipolar inputs.
23
DVDD
Digital Supply Voltage, +5 V. DVDD should not exceed AV DD by more than 0.3 V in normal operation.
24
DGND
Ground reference point for digital circuitry.
TERMINOLOGY
POSITIVE FULL-SCALE OVERRANGE
INTEGRAL NONLINEARITY
Positive full-scale overrange is the amount of overhead available
to handle input voltages on AIN1(+) input greater than
AIN1(–) + VREF/GAIN or on the AIN2 input greater than +
VREF /GAIN (for example, noise peaks or excess voltages due to
system gain errors in system calibration routines) without introducing errors due to overloading the analog modulator or to
overflowing the digital filter.
This is the maximum deviation of any code from a straight line
passing through the endpoints of the transfer function. The endpoints of the transfer function are zero-scale (not to be confused
with bipolar zero), a point 0.5 LSB below the first code transition (000 . . . 000 to 000 . . . 001) and full scale, a point 0.5 LSB
above the last code transition (111 . . . 110 to 111 . . . 111). The
error is expressed as a percentage of full scale.
NEGATIVE FULL-SCALE OVERRANGE
POSITIVE FULL-SCALE ERROR
Positive full-scale error is the deviation of the last code transition (111 . . . 110 to 111 . . . 111) from the ideal input full-scale
voltage. For AIN1(+), the ideal full-scale input voltage is
(AIN1(–) + VREF/GAIN – 3/2 LSBs); for AIN2, the ideal fullscale input voltage is VREF/GAIN – 3/2 LSBs. It applies to both
unipolar and bipolar analog input ranges.
This is the amount of overhead available to handle voltages on
AIN1(+) below AIN1(–) – VREF/GAIN or on AIN2 below
–VREF/GAIN without overloading the analog modulator or overflowing the digital filter. Note that the analog input will accept
negative voltage peaks on AIN1(+) even in the unipolar mode
provided that AIN1(+) is greater than AIN1(–) and greater than
VSS – 30␣ mV.
UNIPOLAR OFFSET ERROR
OFFSET CALIBRATION RANGE
Unipolar offset error is the deviation of the first code transition
from the ideal voltage. For AIN1(+), the ideal input voltage is
(AIN1(–) + 0.5 LSB); for AIN2, the ideal input is 0.5 LSB
when operating in the unipolar mode.
In the system calibration modes, the AD7711 calibrates its
offset with respect to the analog input. The offset calibration
range specification defines the range of voltages that the
AD7711 can accept and still calibrate offset accurately.
BIPOLAR ZERO ERROR
FULL-SCALE CALIBRATION RANGE
This is the deviation of the midscale transition (0111 . . . 111
to 1000 . . . 000) from the ideal input voltage. For AIN1(+), the
ideal input voltage is (AIN1(–) – 0.5 LSB); for AIN2, the ideal
input is – 0.5 LSB when operating in the bipolar mode.
This is the range of voltages that the AD7711 can accept in the
system calibration mode and still calibrate full-scale correctly.
INPUT SPAN
In system calibration schemes, two voltages applied in sequence
to the AD7711’s analog input define the analog input range.
The input span specification defines the minimum and maximum input voltages from zero to full-scale that the AD7711
can accept and still calibrate gain accurately.
BIPOLAR NEGATIVE FULL-SCALE ERROR
This is the deviation of the first code transition from the ideal
input voltage. For (AIN1(+), the ideal input voltage is (AIN1(–)
– VREF/GAIN + 0.5 LSB); for AIN2 the ideal input is – VREF/
GAIN + 0.5 LSB when operating in the bipolar mode.
–8–
REV. F
AD7711
CONTROL REGISTER (24 BITS)
A write to the device with the A0 input low writes data to the control register. A read to the device with the A0 input low accesses the
contents of the control register. The control register is 24-bits wide and when writing to the register 24 bits of data must be written
otherwise the data will not be loaded to the control register. In other words, it is not possible to write just the first 12-bits of data into
the control register. If more than 24 clock pulses are provided before TFS returns high, then all clock pulses after the 24th clock
pulse are ignored. Similarly, a read operation from the control register should access 24 bits of data.
MSB
2
MD2
MD1
MD0
G2
G1
G0
CH
PD
WL
RO
BO
B/U
FS11
FS10
FS9
FS8
FS7
FS6
FS5
FS4
FS3
FS2
FS1
FS0
LSB
Operating Mode
MD2
MD1
MD0
Operating Mode
0
0
0
Normal Mode. This is the normal mode of operation of the device whereby a read to the device with A0
high accesses data from the data register. This is the default condition of these bits after the internal
power on reset.
0
0
1
Activate Self-Calibration. This activates self-calibration on the channel selected by CH. This is a one-step
calibration sequence, and when complete, the part returns to normal mode (with MD2, MD1, MD0 of
the control register returning to 0, 0, 0). The DRDY output indicates when this self-calibration is complete.
For this calibration type, the zero-scale calibration is done internally on shorted (zeroed) inputs and the
full-scale calibration is done internally on VREF.
0
1
0
Activate System Calibration. This activates system calibration on the channel selected by CH. This is a
two-step calibration sequence, with the zero-scale calibration done first on the selected input channel and
DRDY indicating when this zero-scale calibration is complete. The part returns to normal mode at the
end of this first step in the two-step sequence.
0
1
1
Activate System Calibration. This is the second step of the system calibration sequence with full-scale
calibration being performed on the selected input channel. Once again, DRDY indicates when the fullscale calibration is complete. When this calibration is complete, the part returns to normal mode.
1
0
0
Activate System Offset Calibration. This activates system offset calibration on the channel selected by
CH. This is a one-step calibration sequence and, when complete, the part returns to normal mode with
DRDY indicating when this system offset calibration is complete. For this calibration type, the zero-scale
calibration is done on the selected input channel and the full-scale calibration is done internally on VREF.
1
0
1
Activate Background Calibration. This activates background calibration on the channel selected by CH. If
the background calibration mode is on, then the AD7711 provides continuous self-calibration of the
reference and shorted (zeroed) inputs. This calibration takes place as part of the conversion sequence,
extending the conversion time and reducing the word rate by a factor of six. Its major advantage is that
the user does not have to worry about recalibrating the device when there is a change in the ambient
temperature. In this mode, the shorted (zeroed) inputs and VREF, as well as the analog input voltage, are
continuously monitored and the calibration registers of the device are automatically updated.
1
1
0
Read/Write Zero-Scale Calibration Coefficients. A read to the device with A0 high accesses the contents
of the zero-scale calibration coefficients of the channel selected by CH. A write to the device with A0 high
writes data to the zero-scale calibration coefficients of the channel selected by CH. The word length for
reading and writing these coefficients is 24 bits, regardless of the status of the WL bit of the control
register. Therefore, when writing to the calibration register 24 bits of data must be written, otherwise the
new data will not be transferred to the calibration register.
1
1
1
Read/Write Full-Scale Calibration Coefficients. A read to the device with A0 high accesses the contents of
the full-scale calibration coefficients of the channel selected by CH. A write to the device with A0 high
writes data to the full-scale calibration coefficients of the channel selected by CH. The word length for
reading and writing these coefficients is 24 bits, regardless of the status of the WL bit of the control
register. Therefore, when writing to the calibration register 24 bits of data must be written, otherwise the
new data will not be transferred to the calibration register.
REV. F
–9–
AD7711
PGA Gain
G2
Gl
G0
0
0
0
0
1
1
1
1
0
1
0
1
0
1
0
1
0
0
1
1
0
0
1
1
Gain
1
2
4
8
16
32
64
128
(Default Condition After the Internal Power-On Reset)
Channel Selection
CH
0
1
Channel
AIN1
AIN2
(Default Condition After the Internal Power-On Reset)
Power-Down
PD
0
1
Normal Operation
Power-Down
(Default Condition After the Internal Power-On Reset)
Word Length
WL
Output Word Length
0
1
16-bit
24-bit
(Default Condition After Internal Power-On Reset)
RTD Excitation Current
IO
0
1
Off
On
(Default Condition After Internal Power-On Reset)
Burnout Current
BO
0
1
Off
On
(Default Condition After Internal Power-On Reset)
Bipolar/Unipolar Selection (Both Inputs)
B/U
0
1
Bipolar
Unipolar
(Default Condition After Internal Power-On Reset)
Filter Selection (FS11–FS0)
The on-chip digital filter provides a Sinc3 (or (Sinx/x)3) filter response. The 12 bits of data programmed into these bits determine
the filter cutoff frequency, the position of the first notch of the filter and the data rate for the part. In association with the gain selection, it also determines the output noise (and hence the effective resolution) of the device.
The first notch of the filter occurs at a frequency determined by the relationship: filter first notch frequency = (fCLK IN /512)/code
where code is the decimal equivalent of the code in bits FS0 to FS11 and is in the range 19 to 2,000. With the nominal fCLK IN of
10 MHz, this results in a first notch frequency range from 9.76 Hz to 1.028 kHz. To ensure correct operation of the AD7711, the
value of the code loaded to these bits must be within this range. Failure to do this will result in unspecified operation of the device.
Changing the filter notch frequency, as well as the selected gain, impacts resolution. Tables I and II and Figure 2 show the effect of
the filter notch frequency and gain on the effective resolution of the AD7711. The output data rate (or effective conversion time) for
the device is equal to the frequency selected for the first notch of the filter. For example, if the first notch of the filter is selected at
50 Hz, then a new word is available at a 50 Hz rate or every 20 ms. If the first notch is at 1 kHz, a new word is available every 1 ms.
The settling time of the filter to a full-scale step input change is worst case 4 × 1/(output data rate). This settling time is to 100% of
the final value. For example, with the first filter notch at 50 Hz, the settling time of the filter to a full-scale step input change is
80 ms max. If the first notch is at 1 kHz, the settling time of the filter to a full-scale input step is 4 ms max. This settling time can be
reduced to 3 × 1/(output data rate) by synchronizing the step input change to a reset of the digital filter. In other words, if the step
input takes place with SYNC low, the settling time will be 3 × 1/(output data rate). If a change of channels takes place, the settling
time is 3 × 1/(output data rate) regardless of the SYNC input.
The –3 dB frequency is determined by the programmed first notch frequency according to the relationship: filter –3 dB frequency
= 0.262 × first notch frequency.
–10–
REV. F
AD7711
Tables I and II show the output rms noise for some typical notch and –3 dB frequencies. The numbers given are for the bipolar
input ranges with a VREF of +2.5 V. These numbers are typical and are generated with an analog input voltage of 0 V. The output
noise from the part comes from two sources. The first is the electrical noise in the semiconductor devices used in the implementation
of the modulator (device noise). The second occurs when the analog input signal is converted into the digital domain adding quantization noise. The device noise is at a low level and is largely independent of frequency. The quantization noise starts at an even
lower level but rises rapidly with increasing frequency to become the dominant noise source. Consequently, lower filter notch settings (below 60 Hz approximately) tend to be device noise dominated while higher notch settings are dominated by quantization
noise. Changing the filter notch and cutoff frequency in the quantization noise dominated region results in a more dramatic improvement in noise performance than it does in the device noise dominated region as shown in Table I. Furthermore, quantization
noise is added after the PGA, so effective resolution is independent of gain for the higher filter notch frequencies. Meanwhile, device
noise is added in the PGA and, therefore, effective resolution suffers a little at high gains for lower notch frequencies.
At the lower filter notch settings (below 60 Hz), the no missing codes performance of the device is at the 24-bit level. At the higher
settings, more codes will be missed until at 1 kHz notch setting, no missing codes performance is only guaranteed to the 12-bit level.
However, since the effective resolution of the part is 10.5 bits for this filter notch setting, this no missing codes performance should
be more than adequate for all applications.
The effective resolution of the device is defined as the ratio of the output rms noise to the input full scale. This does not remain
constant with increasing gain or with increasing bandwidth. Table II shows the same table as Table I except that the output is now
expressed in terms of effective resolution (the magnitude of the rms noise with respect to 2 × VREF/GAIN, i.e., the input full scale). It
is possible to do post filtering on the device to improve the output data rate for a given –3 dB frequency and also to further reduce
the output noise (see Digital Filtering section).
Table I. Output Noise vs. Gain and First Notch Frequency
First Notch of
Filter and O/P –3␣ dB
Data Rate 1
Frequency
Gain of
1
Gain of
2
Typical Output RMS Noise (␮V)
Gain of
Gain of
Gain of
4
8
16
Gain of
32
Gain of
64
Gain of
128
10␣ Hz2
25␣ Hz2
30␣ Hz2
50␣ Hz2
60␣ Hz2
100␣ Hz3
250␣ Hz3
500␣ Hz3
1␣ kHz3
1.0
1.8
2.5
4.33
5.28
13
130
0.6 × 103
3.1 × 103
0.78
1.1
1.31
2.06
2.36
6.4
75
0.26 × 10 3
1.6 × 103
0.48
0.63
0.84
1.2
1.33
3.7
25
140
0.7 × 10 3
0.25
0.41
0.43
0.46
0.62
0.9
4
25
120
0.25
0.38
0.4
0.46
0.6
0.65
2.7
15
70
0.25
0.38
0.4
0.46
0.56
0.65
1.7
8
40
2.62␣ Hz
6.55␣ Hz
7.86␣ Hz
13.1 Hz
15.72 Hz
26.2 Hz
65.5 Hz
131 Hz
262 Hz
0.33
0.50
0.57
0.64
0.87
1.8
12
70
0.29 × 10 3
0.25
0.44
0.46
0.54
0.63
1.1
7.5
35
180
NOTES
1
The default condition (after the internal power-on reset) for the first notch of filter is 60 Hz.
2
For these filter notch frequencies, the output rms noise is primarily dominated by device noise and as a result is independent of the value of the reference voltage.
Therefore, increasing the reference voltage will give an increase in the effective resolution of the device (i.e., the ratio of the rms noise to the input full scale is increased since the output rms noise remains constant as the input full scale increases).
3
For these filter notch frequencies, the output rms noise is dominated by quantization noise and as a result is proportional to the value of the reference voltage.
Table II. Effective Resolution vs. Gain and First Notch Frequency
First Notch of
Filter and O/P –3␣ dB
Data Rate
Frequency
Gain of
1
10␣ Hz
25␣ Hz
30␣ Hz
50␣ Hz
60␣ Hz
100␣ Hz
250␣ Hz
500␣ Hz
1␣ kHz
22.5
21.5
21
20
20
18.5
15
13
10.5
2.62␣ Hz
6.55␣ Hz
7.86␣ Hz
13.1␣ Hz
15.72␣ Hz
26.2␣ Hz
65.5␣ Hz
131␣ Hz
262␣ Hz
Gain of
2
Effective Resolution 1 (Bits)
Gain of
Gain of
Gain of
4
8
16
Gain of
32
Gain of
64
Gain of
128
21.5
21
21
20
20
18.5
15
13
10.5
21.5
21
20.5
20
20
18.5
15.5
13
11
19.5
18.5
18.5
18.5
18
17.5
15.5
12.5
10.5
18.5
17.5
17.5
17.5
17
17
15
12.5
10
17.5
16.5
16.5
16.5
16
16
14.5
12.5
10
21
20
20
20
19.5
18.5
15.5
13
11
20.5
19.5
19.5
19
19
18
15.5
13
11
NOTE
1
Effective resolution is defined as the magnitude of the output rms noise with respect to the input full scale (i.e., 2 × VREF /GAIN). The above table applies for
a VREF of +2.5 V and resolution numbers are rounded to the nearest 0.5 LSB.
REV. F
–11–
2
AD7711
Figure 2 gives similar information to that outlined in Table I. In this plot, the output rms noise is shown for the full range of available
cutoffs frequencies rather than for some typical cutoff frequencies as in Tables I and II. The numbers given in these plots are typical
values at 25°C.
1000
10000
GAIN OF 1
GAIN OF 2
GAIN OF 16
GAIN OF 4
GAIN OF 32
100
GAIN OF 8
OUTPUT NOISE – mV
OUTPUT NOISE – mV
1000
100
10
GAIN OF 64
GAIN OF 128
10
1
1
0.1
10
100
1000
NOTCH FREQUENCY – Hz
0.1
10
10000
100
1000
NOTCH FREQUENCY – Hz
10000
Figure 2b. Plot of Output Noise vs. Gain and Notch
Frequency (Gains of 16 to 128)
Figure 2a. Plot of Output Noise vs. Gain and Notch
Frequency (Gains of 1 to 8)
CIRCUIT DESCRIPTION
The AD7711 is a sigma-delta A/D converter with on-chip digital
filtering, intended for the measurement of wide dynamic range,
low frequency signals such as those in RTD applications, industrial control or process control applications. It contains a sigmadelta (or charge-balancing) ADC, a calibration microcontroller
with on-chip static RAM, a clock oscillator, a digital filter and a
bidirectional serial communications port.
The part contains two analog input channels, a programmable
gain differential analog input and a programmable gain single
ended input. The gain range is from 1 to 128 allowing the part
to accept unipolar signals of between 0 mV to +20 mV and 0 V
to +2.5 V or bipolar signals in the range from ±20 mV to ± 2.5 V
when the reference input voltage equals +2.5 V. The input
signal to the selected analog input channel is continuously
sampled at a rate determined by the frequency of the master
clock, MCLK IN, and the selected gain (see Table III). A
charge balancing A/D converter (Sigma-Delta Modulator) converts the sampled signal into a digital pulse train whose duty
cycle contains the digital information. The programmable gain
function on the analog input is also incorporated in this sigmadelta modulator with the input sampling frequency being modified to give the higher gains. A sinc3 digital low-pass filter
processes the output of the sigma-delta modulator and updates
the output register at a rate determined by the first notch frequency of this filter. The output data can be read from the serial
port randomly or periodically at any rate up to the output register update rate. The first notch of this digital filter (and hence
its –3 dB frequency) can be programmed via an on-chip control
register. The programmable range for this first notch frequency
is from 9.76 Hz to 1.028 kHz, giving a programmable range for
the –3 dB frequency of 2.58 Hz to 269 Hz.
The basic connection diagram for the part is shown in Figure 3.
This shows the AD7711 in the external clocking mode with
both the AVDD and DVDD pins of the AD7711 being driven
from the analog +5 V supply. Some applications will have
separate supplies for both AVDD and DVDD, and in some of
these cases, the analog supply will exceed the +5 V digital supply (see Power Supplies and Grounding section).
ANALOG
+5V SUPPLY
10mF
0.1mF
0.1mF
AVDD
DIFFERENTIAL
ANALOG INPUT
SINGLE-ENDED
ANALOG INPUT
DRDY
AIN2
AD7711
RTD1
RTD2
ANALOG GROUND
DIGITAL GROUND
DVDD
AIN1(+)
AIN1(–)
AGND
VSS
DGND
REF OUT
REF IN(+)
VBIAS
REF IN(–)
DATA READY
TFS
TRANSMIT (WRITE)
RFS
RECEIVE (READ)
SDATA
SCLK
A0
SERIAL DATA
SERIAL CLOCK
ADDRESS INPUT
MODE
SYNC
+5V
MCLK OUT
MCLK IN
Figure 3. Basic Connection Diagram
The AD7711 provides a number of calibration options which
can be programmed via the on-chip control register. A calibration cycle may be initiated at any time by writing to this control
register. The part can perform self-calibration using the on-chip
calibration microcontroller and SRAM to store calibration parameters. Other system components may also be included in the
calibration loop to remove offset and gain errors in the input
channel using the system calibration mode. Another option is a
background calibration mode where the part continuously performs self-calibration and updates the calibration coefficients.
Once the part is in this mode, the user does not have to worry
about issuing periodic calibration commands to the device or
asking the device to recalibrate when there is a change in the
ambient temperature or power supply voltage.
–12–
REV. F
AD7711
The AD7711 gives the user access to the on-chip calibration
registers allowing the microprocessor to read the device’s calibration coefficients and also to write its own calibration coefficients to the part from prestored values in E2PROM. This gives
the microprocessor much greater control over the AD7711’s
calibration procedure. It also means that the user can verify that
the device has performed its calibration correctly by comparing the
coefficients after calibration with prestored values in E2PROM.
Sigma-delta ADCs are generally described by the order of the
analog low-pass filter. A simple example of a first order sigmadelta ADC is shown in Figure 5. This contains only a first order
low-pass filter or integrator. It also illustrates the derivation of
the alternative name for these devices: Charge-Balancing ADCs.
The AD7711 can be operated in single supply systems provided
that the analog input voltage does not go more negative than
–30 mV. For larger bipolar signals, a VSS of –5 V is required by
the part. For battery operation, the AD7711 also offers a software-programmable standby mode that reduces idle power
consumption to typically 7 mW.
e
DIFFERENTIAL
AMPLIFIER
INTEGRATOR
COMPARATOR
VIN
+FS
DAC
–FS
Figure 5. Basic Charge-Balancing ADC
THEORY OF OPERATION
It consists of a differential amplifier (whose output is the difference between the analog input and the output of a 1-bit DAC),
an integrator and a comparator. The term charge-balancing,
comes from the fact that this system is a negative feedback loop
that tries to keep the net charge on the integrator capacitor at
zero, by balancing charge injected by the input voltage with
charge injected by the 1-bit DAC. When the analog input is
zero, the only contribution to the integrator output comes from
the 1-bit DAC. For the net charge on the integrator capacitor to
be zero, the DAC output must spend half its time at +FS and
half its time at –FS. Assuming ideal components, the duty cycle
of the comparator will be 50%.
The general block diagram of a sigma-delta ADC is shown in
Figure 4. It contains the following elements:
1. A sample-hold amplifier.
2. A differential amplifier or subtracter.
3. An analog low-pass filter.
4. A 1-bit A/D converter (comparator).
5. A 1-bit DAC.
6. A digital low-pass filter.
S/H AMP
+
–
COMPARATOR
ANALOG
LOW-PASS
FILTER
DIGITAL
FILTER
DAC
DIGITAL
DATA
When a positive analog input is applied, the output of the 1-bit
DAC must spend a larger proportion of the time at +FS, so the
duty cycle of the comparator increases. When a negative input
voltage is applied, the duty cycle decreases.
Figure 4. General Sigma-Delta ADC
In operation, the analog signal sample is fed to the subtracter,
along with the output of the 1-bit DAC. The filtered difference
signal is fed to the comparator, whose output samples the difference signal at a frequency many times that of the analog signal
sampling frequency (oversampling).
Oversampling is fundamental to the operation of sigma-delta
ADCs. Using the quantization noise formula for an ADC:
SNR = (6.02 × number of bits + 1.76) dB,
a 1-bit ADC or comparator yields an SNR of 7.78 dB.
The AD7711 samples the input signal at a frequency of 39 kHz or
greater (see Table III). As a result, the quantization noise is
spread over a much wider frequency than that of the band of
interest. The noise in the band of interest is reduced still further
by analog filtering in the modulator loop, which shapes the
quantization noise spectrum to move most of the noise energy to
frequencies outside the bandwidth of interest. The noise performance is thus improved from this 1-bit level to the performance
outlined in Tables I and II and in Figure 2.
The output of the comparator provides the digital input for the
1-bit DAC, so that the system functions as a negative feedback
loop that tries to minimize the difference signal. The digital data
that represents the analog input voltage is contained in the duty
cycle of the pulse train appearing at the output of the comparator. It can be retrieved as a parallel binary data word using a
digital filter.
REV. F
The AD7711 uses a second order sigma-delta modulator and a
digital filter that provides a rolling average of the sampled output. After power-up, or if there is a step change in the input
voltage, there is a settling time that must elapse before valid
data is obtained.
Input Sample Rate
The modulator sample frequency for the device remains at
fCLK IN/512 (19.5 kHz @ fCLK IN = 10 MHz) regardless of the
selected gain. However, gains greater than ×1 are achieved by a
combination of multiple input samples per modulator cycle and
a scaling of the ratio of reference capacitor to input capacitor.
As a result of the multiple sampling, the input sample rate of
the device varies with the selected gain (see Table III). The
effective input impedance is 1/C × fS where C is the input sampling capacitance and fS is the input sample rate.
Table III. Input Sampling Frequency vs. Gain
Gain
Input Sampling Frequency (fS)
1
2
4
8
16
32
64
128
fCLK IN/256 (39 kHz @ fCLK IN = 10 MHz)
2 × fCLK IN/256 (78 kHz @ fCLK IN = 10 MHz)
4 × fCLK IN/256 (156 kHz @ fCLK IN = 10 MHz)
8 × fCLK IN/256 (312 kHz @ fCLK IN = 10 MHz)
8 × fCLK IN/256 (312 kHz @ fCLK IN = 10 MHz)
8 × fCLK IN/256 (312 kHz @ fCLK IN = 10 MHz)
8 × fCLK IN/256 (312 kHz @ fCLK IN = 10 MHz)
8 × fCLK IN/256 (312 kHz @ fCLK IN = 10 MHz)
–13–
2
AD7711
DIGITAL FILTERING
Post Filtering
The AD7711’s digital filter behaves like a similar analog filter,
with a few minor differences.
First, since digital filtering occurs after the A-to-D conversion
process, it can remove noise injected during the conversion
process. Analog filtering cannot do this.
On the other hand, analog filtering can remove noise superimposed on the analog signal before it reaches the ADC. Digital
filtering cannot do this and noise peaks riding on signals near
full scale have the potential to saturate the analog modulator
and digital filter, even though the average value of the signal is
within limits. To alleviate this problem, the AD7711 has
overrange headroom built into the sigma-delta modulator and
digital filter which allows overrange excursions of 5% above the
analog input range. If noise signals are larger than this, consideration should be given to analog input filtering, or to reducing
the input channel voltage so that its full scale is half that of the
analog input channel full scale. This will provide an overrange
capability greater than 100% at the expense of reducing the
dynamic range by 1 bit (50%).
Filter Characteristics
The cutoff frequency of the digital filter is determined by the
value loaded to bits FS0 to FS11 in the control register. At the
maximum clock frequency of 10 MHz, the minimum cutoff
frequency of the filter is 2.58 Hz while the maximum programmable cutoff frequency is 269 Hz.
0
–40
–60
GAIN – dB
–80
–100
–120
–140
–160
–180
–200
–220
10
20
30
40
FREQUENCY – Hz
50
Post filtering can also be used to reduce the output noise from
the device for bandwidths below 2.62 Hz. At a gain of 128, the
output rms noise is 250 nV. This is essentially device noise or
white noise, and since the input is chopped, the noise has a flat
frequency response. By reducing the bandwidth below 2.62 Hz,
the noise in the resultant passband can be reduced. A reduction
in bandwidth by a factor of two results in a √2 reduction in the
output rms noise. This additional filtering will result in a longer
settling time.
The digital filter does not provide any rejection at integer multiples of the modulator sample frequency (n × 19.5 kHz, where
n = 1, 2, 3 . . . ). This means that there are frequency bands,
± f3 dB wide (f3 dB is cutoff frequency selected by FS0 to FS11)
where noise passes unattenuated to the output. However, due to
the AD7711’s high oversampling ratio, these bands occupy only
a small fraction of the spectrum and most broadband noise is
filtered. In any case, because of the high oversampling ratio a
simple, RC, single pole filter is generally sufficient to attenuate
the signals in these bands on the analog input and thus provide
adequate antialiasing filtering.
–20
0
For example, if the required bandwidth is 7.86 Hz but the required update rate is 100 Hz, the data can be taken from the
AD7711 at the 100 Hz rate giving a –3 dB bandwidth of
26.2 Hz. Post filtering can be applied to this to reduce the bandwidth and output noise, to the 7.86 Hz bandwidth level, while
maintaining an output rate of 100 Hz.
Antialias Considerations
Figure 6 shows the filter frequency response for a cutoff frequency of 2.62 Hz which corresponds to a first filter notch frequency of 10 Hz. This is a (sinx/x)3 response (also called sinc3)
that provides >100 dB of 50 Hz and 60 Hz rejection. Programming a different cutoff frequency via FS0–FS11 does not alter
the profile of the filter response; it changes the frequency of the
notches as outlined in the Control Register section.
–240
The on-chip modulator provides samples at a 19.5 kHz output
rate. The on-chip digital filter decimates these samples to provide data at an output rate which corresponds to the programmed first notch frequency of the filter. Since the output
data rate exceeds the Nyquist criterion, the output rate for a
given bandwidth will satisfy most application requirements.
However, there may be some applications which require a
higher data rate for a given bandwidth and noise performance.
Applications which need this higher data rate will require some
post filtering following the digital filter of the AD7711.
60
Figure 6. Frequency Response of AD7711 Filter
Since the AD7711 contains this on-chip, low-pass filtering,
there is a settling time associated with step function inputs, and
data on the output will be invalid after a step change until the
settling time has elapsed. The settling time depends upon the
notch frequency chosen for the filter. The output data rate
equates to this filter notch frequency and the settling time of the
filter to a full-scale step input is four times the output data period. In applications using both input channels, the settling time
of the filter must be allowed to elapse before data from the
second channel is accessed.
If passive components are placed in front of the AD7711, care
must be taken to ensure that the source impedance is low enough
so as not to introduce gain errors in the system. The dc input
impedance for the AD7711 is over 1 GΩ. The input appears as
a dynamic load which varies with the clock frequency and with
the selected gain (see Figure 7). The input sample rate, as
shown in Table III, determines the time allowed for the analog
input capacitor, CIN, to be charged. External impedances result
in a longer charge time for this capacitor and this may result
in gain errors being introduced on the analog inputs. Table IV
shows the allowable external resistance/capacitance values such
that no gain error to the 16-bit level is introduced while Table V
shows the allowable external resistance/capacitance values such
that no gain error to the 20-bit level is introduced. Both inputs
of the differential input channel (AIN1) look into similar input
circuitry.
–14–
REV. F
AD7711
allowed flow into the transducer and a measurement of the
input voltage on the AIN1 input is taken, it can indicate that the
transducer has burned out or gone open circuit. For normal
operation, this burnout current is turned off by writing a 0 to
the BO bit in the control register.
AD7711
RINT
7kV TYP
AIN
CINT
11.5pF TYP
HIGH
IMPEDANCE
>1GV
RTD Excitation Current
VBIAS
The AD7711 also contains two matched 200 µA constant current sources which are provided at the RTD1 and RTD2 pins of
the device. These currents can be turned on/off via the control
register. Writing a 1 to the RO bit of the control register enables
these excitation currents.
SWITCHING FREQUENCY DEPENDS
ON fCLKIN AND SELECTED GAIN
Figure 7. Analog Input Impedance
Table IV. Typical External Series Resistance Which Will Not
Introduce 16-Bit Gain Error
Gain
0
External Capacitance (pF)
50
100
500
1000
1
2
4
8–128
184 kΩ
88.6 kΩ
41.4 kΩ
17.6 kΩ
45.3 kΩ
22.1 kΩ
10.6 kΩ
4.8 kΩ
27.1 kΩ
13.2 kΩ
6.3 kΩ
2.9 kΩ
7.3 kΩ
3.6 kΩ
1.7 kΩ
790 Ω
4.1 kΩ
2.0 kΩ
970 Ω
440 Ω
For four-wire RTD applications, one of these excitation currents is used to provide the excitation current for the RTD, the
second current source can be left unconnected. For three-wire
RTD configurations, the second on-chip current source can be
used to eliminate errors due to voltage drops across lead resistances. Figures 20 to 22 in the APPLICATIONS section show
some RTD configurations with the AD7711.
5000
1.1 kΩ
560 Ω
270 Ω
The temperature coefficient of the RTD current sources is
typically 20 ppm/°C with a typical matching between the temperature coefficients of both current sources of 3 ppm/°C. For
applications where the absolute value of the temperature coefficient is too large, the following schemes can be used to remove
the drift error.
120 Ω
Table V. Typical External Series Resistance Which Will Not
Introduce 20-Bit Gain Error
Gain
0
External Capacitance (pF)
50
100
500
1000
5000
1
2
4
8–128
145 kΩ
70.5 kΩ
31.8 kΩ
13.4 kΩ
34.5 kΩ
16.9 kΩ
8.0 kΩ
3.6 kΩ
700 Ω
350 Ω
170 Ω
80 Ω
20.4 kΩ
10 kΩ
4.8 kΩ
2.2 kΩ
5.2 kΩ
2.5 kΩ
1.2 kΩ
550 Ω
2.8 kΩ
1.4 kΩ
670 Ω
300 Ω
The numbers in the above tables assume a full-scale change on
the analog input. In any case, the error introduced due to longer
charging times is a gain error which can be removed using the
system calibration capabilities of the AD7711, provided that the
resultant span is within the span limits of the system calibration
techniques for the AD7711.
ANALOG INPUT FUNCTIONS
Analog Input Ranges
Both analog inputs are programmable gain, input channels
which can handle either unipolar or bipolar input signals. The
AIN1 channel is a differential channel having a common-mode
range from VSS to AVDD, provided that the absolute value of the
analog input voltage lies between VSS –30 mV and AVDD
+30 mV. The AIN2 input channel is a single-ended input that is
referred to AGND.
The dc input leakage current is 10 pA maximum at 25°C
(± 1 nA over temperature). This results in a dc offset voltage
developed across the source impedance. However, this dc offset
effect can be compensated for by a combination of the differential input capability of the part and its system calibration mode.
Burnout Current
The AIN1(+) input of the AD7711 contains a 4.5 µA current
source that can be turned on/off via the control register. This
current source can be used in checking that a transducer has not
burned out or gone open circuit before attempting to take measurements on that channel. If the current is turned on and
REV. F
The conversion result from the AD7711 is ratiometric to the
VREF voltage. Therefore, if the VREF voltage varies with the RTD
temperature coefficient, the temperature drift from the current
source will be removed. For four-wire RTD applications, the
reference voltage can be made ratiometric to RTD current
source by using the second current with a low t.c. resistor to
generate the reference voltage for the part. In this case if a
12.5 kΩ resistor is used, the 200 µA current source generates
+2.5 V across the resistor. This +2.5 V can be applied to the
REF IN(+) input of the AD7711 and with the REF IN(–) input
at ground it will supply a VREF of 2.5 V for the part. For threewire RTD configurations, the reference voltage for the part is
generated by placing a low t.c. resistor (12.5 kΩ for 2.5 V reference) in series with one of the constant current sources. The
RTD current sources can be driven to within 2 V of AVDD . The
reference input of the AD7711 is differential so the REF IN(+)
and REF IN(–) of the AD7711 are driven from either side of the
resistor. Both schemes ensure that the reference voltage for the
part tracks the RTD current sources over temperature and,
thereby, removes the temperature drift error.
Bipolar/Unipolar Inputs
The two analog inputs on the AD7711 can accept either unipolar or bipolar input voltage ranges. Bipolar or unipolar options
are chosen by programming the B/U bit of the control register.
This programs both channels for either unipolar or bipolar
operation. Programming the part for either unipolar or bipolar
operation does not change any of the input signal conditioning;
it simply changes the data output coding. The data coding is
binary for unipolar inputs and offset binary for bipolar inputs.
The AIN1 input channel is differential and, as a result, the
voltage to which the unipolar and bipolar signals are referenced
is the voltage on the AIN1(–) input. For example, if AIN1(–) is
+1.25 V and the AD7711 is configured for unipolar operation
with a gain of 1 and a VREF of +2.5 V, the input voltage range
–15–
2
AD7711
on the AIN1(+) input is +1.25 V to +3.75 V. If AIN1(–) is
+1.25 V and the AD7711 is configured for bipolar mode with a
gain of 1 and a VREF of +2.5 V, the analog input range on the
AIN1(+) input is –1.25 V to +3.75 V. For the AIN2 input, the
input signals are referenced to AGND.
REF OUT
AD7711
REFERENCE INPUT/OUTPUT
REF IN(+)
REF IN(–)
Figure 8. REF OUT/REF IN Connection
The AD7711 contains a temperature compensated +2.5 V reference which has an initial tolerance of ± 1%. This reference voltage is provided at the REF OUT pin and it can be used as the
reference voltage for the part by connecting the REF OUT pin
to the REF IN(+) pin. This REF OUT pin is a single-ended
output, referenced to AGND, which is capable of providing up
to 1 mA to an external load. In applications where REF OUT is
connected directly to REF IN(+), REF IN(–) should be tied to
AGND to provide the nominal +2.5 V reference for the
AD7711.
The reference inputs of the AD7711, REF IN(+) and
REF IN(–), provide a differential reference input capability. The
common-mode range for these differential inputs is from VSS to
AVDD . The nominal differential voltage, VREF (REF IN(+) –
REF IN(–)), is +2.5 V for specified operation, but the reference
voltage can go to +5 V with no degradation in performance
provided that the absolute value of REF IN(+) and REF IN(–)
does not exceed its AVDD and VSS limits and the VBIAS input
voltage range limits are obeyed. The part is also functional with
VREF voltages down to 1 V but with degraded performance as
the output noise will, in terms of LSB size, be larger. REF
IN(+) must always be greater than REF IN(–) for correct operation of the AD7711.
Both reference inputs provide a high impedance, dynamic load
similar to the analog inputs. The maximum dc input leakage
current is 10 pA (± 1 nA over temperature) and source resistance may result in gain errors on the part. The reference inputs
look like the analog input (see Figure 7). In this case, RINT is
5 kΩ typ and CINT varies with gain. The input sample rate is
fCLK IN/256 and does not vary with gain. For gains of 1 to 8 CINT
is 20 pF; for a gain of 16 it is 10 pF; for a gain of 32 it is 5 pF;
for a gain of 64 it is 2.5 pF; and for a gain of 128 it is 1.25 pF.
The digital filter of the AD7711 removes noise from the reference input just as it does with the analog input, and the same
limitations apply regarding lack of noise rejection at integer
multiples of the sampling frequency. The output noise performance outlined in Tables I and II assumes a clean reference. If
the reference noise in the bandwidth of interest is excessive, it
can degrade the performance of the AD7711. Using the on-chip
reference as the reference source for the part (i.e., connecting
REF OUT to REF IN) results in somewhat degraded output
noise performance from the AD7711 for portions of the noise
table that are dominated by the device noise. The on-chip
reference noise effect is eliminated in ratiometric applications
where the reference is used to provide the excitation voltage for
the analog front end. The connection shown in Figure 8 is recommended when using the on-chip reference. Recommended
reference voltage sources for the AD7711 include the AD580
and AD680 2.5 V references.
VBIAS Input
The VBIAS input determine at what voltage the internal analog
circuitry is biased. It essentially provides the return path for
analog currents flowing in the modulator and, as such, it should
be driven from a low impedance point to minimize errors.
For maximum internal headroom, the VBIAS voltage should be
set halfway between AVDD and VSS. The difference between
AVDD and (VBIAS + 0.85 × VREF) determines the amount of
headroom the circuit has at the upper end, while the difference
between VSS and (VBIAS – 0.85 × VREF) determines the amount
of headroom the circuit has at the lower end. Care should be
taken in choosing a VBIAS voltage to ensure that it stays within
prescribed limits. For single +5 V operation, the selected VBIAS
voltage must ensure that VBIAS ± 0.85 × VREF does not exceed
AVDD or VSS or that the VBIAS voltage itself is greater than VSS
+ 2.1 V and less than AVDD – 2.1 V. For single +10 V operation
or dual ± 5 V operation, the selected VBIAS voltage must ensure
that VBIAS × 0.85 × VREF does not exceed AVDD or VSS or that
the VBIAS voltage itself is greater than VSS + 3 V or less than
AVDD – 3 V. For example, with AVDD = +4.75 V, VSS = 0 V
and VREF = +2.5 V, the allowable range for the VBIAS voltage is
+2.125 V to +2.625 V. With AVDD = +9.5 V, VSS = 0 V and
VREF = +5 V, the range for VBIAS is +4.25 V to +5.25 V. With
AVDD = +4.75 V, VSS = –4.75 V and VREF = +2.5 V, the VBIAS
range is –2.625 V to +2.625 V.
The VBIAS voltage does have an effect on the AVDD power supply rejection performance of the AD7711. If the VBIAS voltage
tracks the AVDD supply, it improves the power supply rejection
from the AVDD supply line from 80 dB to 95 dB. Using an
external Zener diode, connected between the AVDD line and
VBIAS, as the source for the VBIAS voltage gives the improvement
in AVDD power supply rejection performance.
USING THE AD7711
SYSTEM DESIGN CONSIDERATIONS
The AD7711 operates differently from successive approximation ADCs or integrating ADCs. Since it samples the signal
continuously, like a tracking ADC, there is no need for a start
convert command. The output register is updated at a rate
determined by the first notch of the filter and the output can be
read at any time, either synchronously or asynchronously.
Clocking
The AD7711 requires a master clock input, which may be an
external TTL/CMOS compatible clock signal applied to the
MCLK IN pin with the MCLK OUT pin left unconnected.
Alternatively, a crystal of the correct frequency can be connected between MCLK IN and MCLK OUT, in which case the
clock circuit will function as a crystal controlled oscillator. For
lower clock frequencies, a ceramic resonator may be used
instead of the crystal. For these lower frequency oscillators,
external capacitors may be required on either the ceramic resonator or on the crystal.
–16–
REV. F
AD7711
The input sampling frequency, the modulator sampling frequency, the –3 dB frequency, output update rate and calibration
time are all directly related to the master clock frequency,
fCLK IN. Reducing the master clock frequency by a factor of two
will halve the above frequencies and update rate and will double
the calibration time.
The current drawn from the DVDD power supply is also directly
related to fCLK IN. Reducing fCLK IN by a factor of two will halve
the DVDD current but will not affect the current drawn from the
AVDD power supply.
System Synchronization
If multiple AD7711s are operated from a common master clock,
they can be synchronized to update their output registers simultaneously. A falling edge on the SYNC input resets the filter and
places the AD7711 into a consistent, known state. A common
signal to the AD7711s’ SYNC inputs will synchronize their
operation. This would normally be done after each AD7711 has
performed its own calibration or has had calibration coefficients
loaded to it.
The SYNC input can also be used to reset the digital filter in
systems where the turn-on time of the digital power supply
(DVDD ) is very long. In such cases, the AD7711 will start operating internally before the DVDD line has reached its minimum
operating level, +4.75 V. With a low DVDD voltage, the
AD7711’s internal digital filter logic does not operate correctly.
Thus, the AD7711 may have clocked itself into an incorrect
operating condition by the time that DVDD has reached its correct level. The digital filter will be reset upon issue of a calibration command (whether it is self-calibration, system calibration
or background calibration) to the AD7711. This ensures correct
operation of the AD7711. In systems where the power-on
default conditions of the AD7711 are acceptable, and no calibration is performed after power-on, issuing a SYNC pulse to
the AD7711 will reset the AD7711’s digital filter logic. An R, C
on the SYNC line, with R, C time constant longer than the
DVDD power-on time, will perform the SYNC function.
ACCURACY
Sigma-delta ADCs, like VFCs and other integrating ADCs, do
not contain any source of nonmonotonicity and inherently offer
no missing codes performance. The AD7711 achieves excellent
linearity by the use of high quality, on-chip silicon dioxide
capacitors, which have a very low capacitance/voltage coefficient. The device also achieves low input drift through the use
of chopper stabilized techniques in its input stage. To ensure
excellent performance over time and temperature, the AD7711
uses digital calibration techniques which minimize offset and
gain error.
The AD7711 also provides the facility to write to the on-chip
calibration registers and in this manner the span and offset for
the part can be adjusted by the user. The offset calibration register contains a value which is subtracted from all conversion
results, while the full-scale calibration register contains a value
which is multiplied by all conversion results. The offset calibration coefficient is subtracted from the result prior to the multiplication by the full-scale coefficient. In the first three modes
outlined here, the DRDY line indicates that calibration is complete by going low. If DRDY is low before (or goes low during)
the calibration command, it may take up to one modulator cycle
before DRDY goes high to indicate that calibration is in
progress. Therefore, DRDY should be ignored for up to one
modulator cycle after the last bit of the calibration command is
written to the control register.
Self-Calibration
In the self-calibration mode with a unipolar input range, the
zero-scale point used in determining the calibration coefficients
is with both inputs shorted (i.e., AIN1(+) = AIN1(–) = VBIAS
for AIN1 and AIN2 = VBIAS for AIN2) and the full-scale point is
VREF. The zero-scale coefficient is determined by converting an
internal shorted inputs node. The full-scale coefficient is determined from the span between this shorted inputs conversion
and a conversion on an internal VREF node. The self-calibration
mode is invoked by writing the appropriate values (0, 0, 1) to
the MD2, MD1 and MD0 bits of the control register. In this
calibration mode, the shorted inputs node is switched into the
modulator first and a conversion is performed; the VREF node is
then switched in and another conversion is performed. When
the calibration sequence is complete, the calibration coefficients
updated and the filter resettled to the analog input voltage, the
DRDY output goes low. The self-calibration procedure takes
into account the selected gain on the PGA.
For bipolar input ranges in the self-calibrating mode, the sequence is very similar to that just outlined. In this case, the two
points which the AD7711 calibrates are midscale (bipolar zero)
and positive full scale.
System Calibration
AUTOCALIBRATION
Autocalibration on the AD7711 removes offset and gain errors
from the device. A calibration routine should be initiated on the
device whenever there is a change in the ambient operating
temperature or supply voltage. It should also be initiated if there
is a change in the selected gain, filter notch or bipolar/unipolar
input range. However, if the AD7711 is in its background calibration mode, the above changes are all automatically taken care
of (after the settling time of the filter has been allowed for).
REV. F
The AD7711 offers self-calibration, system calibration and
background calibration facilities. For calibration to occur on the
selected channel, the on-chip microcontroller must record the
modulator output for two different input conditions. These are
“zero-scale” and “full-scale” points. With these readings, the
microcontroller can calculate the gain slope for the input to
output transfer function of the converter. Internally, the part
works with a resolution of 33 bits to determine its conversion
result of either 16 bits or 24 bits.
System calibration allows the AD7711 to compensate for
system gain and offset errors as well as its own internal errors.
System calibration performs the same slope factor calculations
as self-calibration but uses voltage values presented by the system to the AIN inputs for the zero and full-scale points. System
calibration is a two-step process. The zero-scale point must be
presented to the converter first. It must be applied to the converter before the calibration step is initiated and must remain
stable until the step is complete. System calibration is initiated
by writing the appropriate values (0, 1, 0) to the MD2, MD1
and MD0 bits of the control register. The DRDY output from
the device will signal when the step is complete by going low.
–17–
2
AD7711
After the zero-scale point is calibrated, the full-scale point is
applied and the second step of the calibration process is initiated
by again writing the appropriate values (0, 1, 1) to MD2, MD1
and MD0. Again the full-scale voltage must be set up before the
calibration is initiated and it must remain stable throughout the
calibration step. DRDY goes low at the end of this second step
to indicate that the system calibration is complete. In the unipolar mode, the system calibration is performed between the two
endpoints of the transfer function; in the bipolar mode, it is
performed between midscale and positive full scale.
This two-step system calibration mode offers another feature.
After the sequence has been completed, additional offset or gain
calibrations can be performed by themselves to adjust the zero
reference point or the system gain. This is achieved by performing the first step of the system calibration sequence (by writing
0, 1, 0 to MD2, MD1, MD0). This will adjust the zero-scale or
offset point but will not change the slope factor from what was
set during a full system calibration sequence.
System calibration can also be used to remove any errors from
an antialiasing filter on the analog input. A simple R, C antialiasing filter on the front end may introduce a gain error on the
analog input voltage but the system calibration can be used to
remove this error.
System Offset Calibration
System offset calibration is a variation of both the system calibration and self-calibration. In this case, the zero-scale point
for the system is presented to the AIN input of the converter.
System-offset calibration is initiated by writing 1, 0, 0 to MD2,
MD1, MD0. The system zero-scale coefficient is determined by
converting the voltage applied to the AIN input, while the fullscale coefficient is determined from the span between this AIN
conversion and a conversion on VREF. The zero-scale point
should be applied to the AIN input for the duration of the calibration sequence. This is a one-step calibration sequence with
DRDY going low when the sequence is completed. In the unipolar mode, the system offset calibration is performed between
the two end points of the transfer function; in the bipolar mode,
it is performed between midscale and positive full scale.
Background Calibration
The AD7711 also offers a background calibration mode where
the part interleaves its calibration procedure with its normal
conversion sequence. In the background calibration mode, the
same voltages are used as the calibration points as are used in
the self-calibration mode, i.e., shorted inputs and VREF. The
background calibration mode is invoked by writing 1, 0, 1 to
MD2, MD1, MD0 of the control register. When invoked, the
background calibration mode reduces the output data rate of the
AD7711 by a factor of six while the –3 dB bandwidth remains
unchanged. Its advantage is that the part is continually performing calibration and automatically updating its calibration coefficients. As a result, the effects of temperature drift, supply
sensitivity and time drift on zero and full-scale errors are automatically removed. When the background calibration mode is
turned on, the part will remain in this mode until bits MD2,
MD1 and MD0 of the control register are changed. With background calibration mode on, the first result from the AD7711
will be incorrect as the full-scale calibration will not have been
performed. For a step change on the input, the second output
update will have settled to 100% of the final value.
Table VI summarizes the calibration modes and the calibration
points associated with them. It also gives the duration from
when the calibration is invoked to when valid data is available to
the user.
Span and Offset Limits
Whenever a system calibration mode is used, there are limits on
the amount of offset and span that can be accommodated. The
range of input span in both the unipolar and bipolar modes has
a minimum value of 0.8 × VREF/GAIN and a maximum value of
2.1 × VREF/GAIN.
The amount of offset which can be accommodated depends on
whether the unipolar or bipolar mode is being used. This offset
range is limited by the requirement that the positive full-scale
calibration limit is ≤ 1.05 × VREF/GAIN. Therefore, the offset
range plus the span range cannot exceed 1.05 × VREF/GAIN. If
the span is at its minimum (0.8 × VREF/GAIN) the maximum
the offset can be is (0.25 × VREF/GAIN).
In the bipolar mode, the system offset calibration range is again
restricted by the span range. The span range of the converter in
bipolar mode is equidistant around the voltage used for the
zero-scale point thus the offset range plus half the span range
cannot exceed (1.05 × VREF/GAIN). If the span is set to 2 × VREF/
GAIN, the offset span cannot move more than ±(0.05 × VREF/
GAIN) before the endpoints of the transfer function exceed the
input overrange limits ±(1.05 × VREF/GAIN). If the span range
is set to the minimum ± (0.4 × V REF/GAIN) the maximum allowable offset range is ±(0.65 × VREF/GAIN).
Table VI. Calibration Truth Table
Cal Type
MD2, MD1, MD0
Zero-Scale Cal
Full-Scale Cal
Sequence
Duration
Self-Cal
System Cal
System Cal
System Offset Cal
Background Cal
0, 0, 1
0, 1, 0
0, 1, 1
1, 0, 0
1, 0, 1
Shorted Inputs
AIN
–
AIN
Shorted Inputs
VREF
–
AIN
VREF
VREF
One Step
Two Step
Two Step
One Step
One Step
9 × 1/Output Rate
4 × 1/Output Rate
4 × 1/Output Rate
9 × 1/Output Rate
6 × 1/Output Rate
–18–
REV. F
AD7711
POWER-UP AND CALIBRATION
On power-up, the AD7711 performs an internal reset which sets
the contents of the control register to a known state. However,
to ensure correct calibration for the device a calibration routine
should be performed after power-up.
The power dissipation and temperature drift of the AD7711 are
low and no warm up time is required before the initial calibration is performed. However, if an external reference is being
used, this reference must have stabilized before calibration is
initiated.
Drift Considerations
The AD7711 uses chopper stabilization techniques to minimize
input offset drift. Charge injection in the analog switches and dc
leakage currents at the sampling node are the primary sources of
offset voltage drift in the converter. The dc input leakage current is essentially independent of the selected gain. Gain drift
within the converter depends primarily upon the temperature
tracking of the internal capacitors. It is not affected by leakage
currents.
Measurement errors due to offset drift or gain drift can be eliminated at any time by recalibrating the converter or by operating
the part in the background calibration mode. Using the system
calibration mode can also minimize offset and gain errors in the
signal conditioning circuitry. Integral and differential linearity
errors are not significantly affected by temperature changes.
The analog and digital supplies to the AD7711 are independent
and separately pinned out to minimize coupling between the
analog and digital sections of the device. The digital filter will
provide rejection of broadband noise on the power supplies,
except at integer multiples of the modulator sampling frequency.
The digital supply (DVDD ) must not exceed the analog positive
supply (AVDD ) by more than 0.3 V in normal operation. If separate analog and digital supplies are used, the recommended
decoupling scheme is shown in Figure 9. In systems where
AVDD = +5 V and DVDD = +5 V, it is recommended that AVDD
and DVDD are driven from the same +5 V supply, although each
supply should be decoupled separately as shown in Figure 9. It
is preferable that the common supply is the system’s analog +5 V
supply.
It is also important that power is applied to the AD7711 before
signals at REF IN, AIN or the logic input pins in order to avoid
latch-up. If separate supplies are used for the AD7711 and the
system digital circuitry, then the AD7711 should be powered up
first. If it is not possible to guarantee this, then current limiting
resistors should be placed in series with the logic inputs.
10mF
DIGITAL +5V
SUPPLY
0.1mF
0.1mF
AVDD
DVDD
AD7711
POWER SUPPLIES AND GROUNDING
Since the analog inputs and reference input are differential,
most of the voltages in the analog modulator are common-mode
voltages. VBIAS provides the return path for most of the analog
currents flowing in the analog modulator. As a result, the VBIAS
input should be driven from a low impedance to minimize errors
due to charging/discharging impedances on this line. When the
internal reference is used as the reference source for the part,
AGND is the ground return for this reference voltage.
REV. F
ANALOG
SUPPLY
–19–
Figure 9. Recommended Decoupling Scheme
2
AD7711
the output data register. It is reset high when the last bit of data
(either 16th bit or 24th bit) is read from the output register. If
data is not read from the output register, the DRDY line will
remain low. The output register will continue to be updated at
the output update rate but DRDY will not indicate this. A read
from the device in this circumstance will access the most recent
word in the output register. If a new data word becomes available to the output register while data is being read from the
output register, DRDY will not indicate this and the new data
word will be lost to the user. DRDY is not affected by reading
from the control register or the calibration registers.
DIGITAL INTERFACE
The AD7711’s serial communications port provides a flexible
arrangement to allow easy interfacing to industry-standard
microprocessors, microcontrollers and digital signal processors.
A serial read to the AD7711 can access data from the output
register, the control register or from the calibration registers. A
serial write to the AD7711 can write data to the control register
or the calibration registers.
Two different modes of operation are available, optimized for
different types of interface where the AD7711 can act either as
master in the system (it provides the serial clock) or as slave (an
external serial clock can be provided to the AD7711). These
two modes, labelled self-clocking mode and external clocking
mode, are discussed in detail in the following sections.
Data can only be accessed from the output data register when
DRDY is low. If RFS goes low with DRDY high, no data transfer will take place. DRDY does not have any effect on reading
data from the control register or from the calibration registers.
Self-Clocking Mode
The AD7711 is configured for its self-clocking mode by tying
the MODE pin high. In this mode, the AD7711 provides the
serial clock signal used for the transfer of data to and from the
AD7711. This self-clocking mode can be used with processors
that allow an external device to clock their serial port including
most digital signal processors and microcontrollers such as the
68HC11 and 68HC05. It also allows easy interfacing to serialparallel conversion circuits in systems with parallel data communication, allowing interfacing to 74XX299 universal shift
registers without any additional decoding. In the case of shift
registers, the serial clock line should have a pull-down resistor
instead of the pull-up resistor shown in Figures 10 and 11.
Read Operation
Data can be read from either the output register, the control
register or the calibration registers. A0 determines whether the
data read accesses data from the control register or from the
output/calibration registers. This A0 signal must remain valid
for the duration of the serial read operation. With A0 high, data
is accessed from either the output register or from the calibration registers. With A0 low, data is accessed from the control
register.
The function of the DRDY line is dependent only on the output
update rate of the device and the reading of the output data
register. DRDY goes low when a new data word is available in
Figure 10 shows a timing diagram for reading from the AD7711
in the self-clocking mode. The read operation shows a read from
the AD7711’s output data register. A read from the control
register or calibration registers is similar but in these cases the
DRDY line is not related to the read function. Depending on
the output update rate, it can go low at any stage in the control/
calibration register read cycle without affecting the read and its
status should be ignored. A read operation from either the control or calibration registers must always read 24 bits of data
from the respective register.
Figure 10 shows a read operation from the AD7711. For the
timing diagram shown, it is assumed that there is a pull-up
resistor on the SCLK output. With DRDY low, the RFS
input is brought low. RFS going low enables the serial clock of
the AD7711 and also places the MSB of the word on the serial
data line. All subsequent data bits are clocked out on a high to
low transition of the serial clock and are valid prior to the following rising edge of this clock. The final active falling edge of
SCLK clocks out the LSB and this LSB is valid prior to the final
active rising edge of SCLK. Coincident with the next falling
edge of SCLK, DRDY is reset high. DRDY going high turns off
the SCLK and the SDATA outputs. This means that the data
hold time for the LSB is slightly shorter than for all other bits.
DRDY (O)
t3
t2
A0 (I)
t5
t4
RFS (I)
t9
t6
SCLK (O)
t7
SDATA (O)
t8
t10
MSB
LSB
THREE-STATE
Figure 10. Self-Clocking Mode, Output Data Read Operation
–20–
REV. F
AD7711
Write Operation
Read Operation
Data can be written to either the control register or calibration
registers. In either case, the write operation is not affected by
the DRDY line and the write operation does not have any effect
on the status of DRDY. A write operation to the control register
or the calibration register must always write 24 bits to the respective register.
As with the self-clocking mode, data can be read from either the
output register, the control register or the calibration registers.
A0 determines whether the data read accesses data from the
control register or from the output/calibration registers. This A0
signal must remain valid for the duration of the serial read
operation. With A0 high, data is accessed from either the output
register or from the calibration registers. With A0 low, data is
accessed from the control register.
Figure 11 shows a write operation to the AD7711. A0 determines whether a write operation transfers data to the control
register or to the calibration registers. This A0 signal must
remain valid for the duration of the serial write operation. The
falling edge of TFS enables the internally generated SCLK
output. The serial data to be loaded to the AD7711 must be
valid on the rising edge of this SCLK signal. Data is clocked
into the AD7711 on the rising edge of the SCLK signal with the
MSB transferred first. On the last active high time of SCLK, the
LSB is loaded to the AD7711. Subsequent to the next falling
edge of SCLK, the SCLK output is turned off. (The timing
diagram of Figure 11 assumes a pull-up resistor on the SCLK
line.)
The function of the DRDY line is dependent only on the output
update rate of the device and the reading of the output data
register. DRDY goes low when a new data word is available in
the output data register. It is reset high when the last bit of data
(either 16th bit or 24th bit) is read from the output register. If
data is not read from the output register, the DRDY line will
remain low. The output register will continue to be updated at
the output update rate but DRDY will not indicate this. A read
from the device in this circumstance will access the most recent
word in the output register. If a new data word becomes available to the output register while data is being read from the
output register, DRDY will not indicate this and the new data
word will be lost to the user. DRDY is not affected by reading
from the control register or the calibration register.
External Clocking Mode
The AD7711 is configured for its external clocking mode by
tying the MODE pin low. In this mode, SCLK of the AD7711
is configured as an input and an external serial clock must be
provided to this SCLK pin. This external clocking mode is
designed for direct interface to systems which provide a serial
clock output that is synchronized to the serial data output,
including microcontrollers such as the 80C51, 87C51, 68HC11
and 68HC05 and most digital signal processors.
Data can only be accessed from the output data register when
DRDY is low. If RFS goes low while DRDY is high, no data
transfer will take place. DRDY does not have any effect on reading
data from the control register or from the calibration registers.
A0 (I)
t14
t15
TFS (I)
t17
t16
t9
SCLK (O)
t18
SDATA (I)
t19
t10
MSB
LSB
Figure 11. Self-Clocking Mode, Control/Calibration Register Write Operation
REV. F
–21–
2
AD7711
Figures 12a and 12b show timing diagrams for reading from the
AD7711 in the external clocking mode. Figure 12a shows a
situation where all the data is read from the AD7711 in one read
operation. Figure 12b shows a situation where the data is read
from the AD7711 over a number of read operations. Both read
operations show a read from the AD7711’s output data register.
A read from the control register or calibration registers is similar
but in these cases the DRDY line is not related to the read function. Depending on the output update rate, it can go low at any
stage in the control/calibration register read cycle without affecting the read and its status should be ignored. A read operation
from either the control or calibration registers must always read
24 bits of data from the respective register.
Figure 12a shows a read operation from the AD7711 where
RFS remains low for the duration of the data word transmission.
With DRDY low, the RFS input is brought low. The input
SCLK signal should be low between read and write operations.
RFS going low places the MSB of the word to be read on the
serial data line. All subsequent data bits are clocked out on a
high to low transition of the serial clock and are valid prior to
the following rising edge of this clock. The penultimate falling
edge of SCLK clocks out the LSB and the final falling edge
resets the DRDY line high. This rising edge of DRDY turns off
the serial data output.
Figure 12b shows a timing diagram for a read operation where
RFS returns high during the transmission of the word and
returns low again to access the rest of the data word. Timing
parameters and functions are very similar to that outlined for
Figure 12a but Figure 12b has a number of additional times to
show timing relationships when RFS returns high in the middle
of transferring a word.
RFS should return high during a low time of SCLK. On the
rising edge of RFS, the SDATA output is turned off. DRDY
remains low and will remain low until all bits of the data word
are read from the AD7711, regardless of the number of times
RFS changes state during the read operation. Depending on the
time between the falling edge of SCLK and the rising edge of
RFS, the next bit (BIT N+1) may appear on the databus before
RFS goes high. When RFS returns low again, it activates the
SDATA output. When the entire word is transmitted, the
DRDY line will go high turning off the SDATA output as per
Figure 12a.
DRDY (O)
t21
t20
A0 (I)
t22
t23
RFS (I)
t26
t28
SCLK (I)
t24
SDATA (O)
t27
t25
MSB
t29
LSB
THREE-STATE
Figure 12a. External-Clocking Mode, Output Data Read Operation
DRDY (O)
t20
A0 (I)
t22
RFS (I)
t26
t30
SCLK (I)
t24
t24
t27
t31
t25
SDATA (O)
t25
THREE-STATE
MSB
BIT N
BIT N+1
Figure 12b. External-Clocking Mode, Output Data Read Operation ( RFS Returns High During Read Operation)
–22–
REV. F
AD7711
signal. Data is clocked into the AD7711 on the high level of this
SCLK signal with the MSB transferred first. On the last active
high time of SCLK, the LSB is loaded to the AD7711.
Write Operation
Data can be written to either the control register or calibration
registers. In either case, the write operation is not affected by
the DRDY line and the write operation does not have any effect
on the status of DRDY. A write operation to the control register
or the calibration register must always write 24 bits to the
respective register.
Figure 13b shows a timing diagram for a write operation to the
AD7711 with TFS returning high during the write operation
and returning low again to write the rest of the data word. Timing parameters and functions are very similar to that outlined for
Figure 13a, but Figure 13b has a number of additional times to
show timing relationships when TFS returns high in the middle
of transferring a word.
Figure 13a shows a write operation to the AD7711 with TFS
remaining low for the duration of the write operation. A0 determines whether a write operation transfers data to the control
register or to the calibration registers. This A0 signal must
remain valid for the duration of the serial write operation. As
before, the serial clock line should be low between read and
write operations. The serial data to be loaded to the AD7711
must be valid on the high level of the externally applied SCLK
Data to be loaded to the AD7711 must be valid prior to the
rising edge of the SCLK signal. TFS should return high during
the low time of SCLK. After TFS returns low again, the next bit
of the data word to be loaded to the AD7711 is clocked in on
next high level of the SCLK input. On the last active high time
of the SCLK input, the LSB is loaded to the AD7711.
A0 (I)
t32
t33
TFS (I)
t26
t34
SCLK (I)
SDATA (I)
t27
t36
t35
MSB
LSB
Figure 13a. External-Clocking Mode, Control/Calibration Register Write Operation
A0 (I)
t32
TFS (I)
t26
t30
SCLK (I)
t35
t27
t36
SDATA (I)
MSB
BIT N
t35
t36
BIT N+1
Figure 13b. External-Clocking Mode, Control/Calibration Register Write Operation
(TFS Returns High During Write Operation)
REV. F
–23–
2
AD7711
SIMPLIFYING THE EXTERNAL CLOCKING MODE
INTERFACE
START
In many applications, the user may not require the facility of
writing to the on-chip calibration registers. In this case, the
serial interface to the AD7711 in external clocking mode can be
simplified by connecting the TFS line to the A0 input of the
AD7711 (see Figure 14). This means that any write to the device will load data to the control register (since A0 is low while
TFS is low) and any read to the device will access data from the
output data register or from the calibration registers (since A0 is
high while RFS is low). It should be noted that in this arrangement the user does not have the capability of reading from the
control register.
CONFIGURE AND
INITIALIZE mC/mP
SERIAL PORT
BRING
RFS, TFS HIGH
POLL DRDY
RFS
FOUR
INTERFACE
LINES
SDATA
SCLK
AD7711
DRDY
LOW?
TFS
NO
YES
A0
Figure 14. Simplified Interface with TFS Connected to A0
BRING
RFS LOW
Another method of simplifying the interface is to generate the
TFS signal from an inverted RFS signal. However, generating
the signals the opposite way around (RFS from an inverted
TFS) will cause writing errors.
X3
READ
SERIAL BUFFER
MICROCOMPUTER/MICROPROCESSOR INTERFACING
The AD7711’s flexible serial interface allows for easy interface
to most microcomputers and microprocessors. Figure 15 shows
a flowchart diagram for a typical programming sequence for
reading data from the AD7711 to a microcomputer while Figure
16 shows a flowchart diagram for writing data to the AD7711.
Figures 17, 18 and 19 show some typical interface circuits.
The flowchart of Figure 15 is for continuous read operations
from the AD7711 output register. In the example shown, the
DRDY line is continuously polled. Depending on the microprocessor configuration, the DRDY line may come to an interrupt
input in which case the DRDY will automatically generate an
interrupt without being polled. The reading of the serial buffer
could be anything from one read operation up to three read
operations (where 24 bits of data are read into an 8-bit serial
register). A read operation to the control/calibration registers is
similar but in this case the status of DRDY can be ignored. The
A0 line is brought low when the RFS line is brought low when
reading from the control register.
The flowchart also shows the bits being reversed after they have
been read in from the serial port. This depends on whether the
microprocessor expects the MSB of the word first or the LSB of
the word first. The AD7711 outputs the MSB first.
BRING
RFS HIGH
REVERSE
ORDER OF BITS
Figure 15. Flowchart for Continuous Read Operations to
the AD7711
The flowchart for Figure 16 is for a single 24-bit write operation
to the AD7711 control or calibration registers. This shows data
being transferred from data memory to the accumulator before
being written to the serial buffer. Some microprocessor systems
will allow data to be written directly to the serial buffer from
data memory. The writing of data to the serial buffer from the
accumulator will generally consist of either two or three write
operations, depending on the size of the serial buffer.
The flowchart also shows the option of the bits being reversed
before being written to the serial buffer. This depends on
whether the first bit transmitted by the microprocessor is the
MSB or the LSB. The AD7711 expects the MSB as the first bit
in the data stream. In cases where the data is being read or
being written in bytes and the data has to be reversed, the bits
will have to be reversed for every byte.
–24–
REV. F
AD7711
Table VII shows some typical 8XC51 code used for a single 24bit read from the output register of the AD7711. Table VIII
shows some typical code for a single write operation to the control register of the AD7711. The 8XC51 outputs the LSB first
in a write operation while the AD7711 expects the MSB first so
the data to be transmitted has to be rearranged before being
written to the output serial register. Similarly, the AD7711
outputs the MSB first during a read operation while the 8XC51
expects the LSB first. Therefore, the data which is read into the
serial buffer needs to be rearranged before the correct data word
from the AD7711 is available in the accumulator.
START
CONFIGURE AND
INITIALIZE mC/mP
SERIAL PORT
BRING
RFS, TFS & A0 HIGH
Table VII. 8XC51 Code for Reading from the AD7711
LOAD DATA FROM
ADDRESS TO
ACCUMULATOR
REVERSE
ORDER OF
BITS
BRING
TFS & A0 LOW
X3
WRITE DATA FROM
ACCUMULATOR TO
SERIAL BUFFER
BRING
TFS & A0 HIGH
END
Figure 16. Flowchart for Single Write Operation to the
AD7711
AD7711–8051 Interface
Figure 17 shows an interface between the AD7711 and the
8XC51 microcontroller. The AD7711 is configured for its external clocking mode while the 8XC51 is configured in its Mode
0 serial interface mode. The DRDY line from the AD7711 is
connected to the Port P1.2 input of the 8XC51 so the DRDY
line is polled by the 8XC51. The DRDY line can be connected
to the INT1 input of the 8XC51 if an interrupt driven system is
preferred.
DVDD
SYNC
P1.0
8XC51
RFS
P1.1
TFS
P1.2
DRDY
P1.3
A0
P3.0
SDATA
P3.1
SCLK
AD7711
MOV SCON,#00010001B; Configure 8051 for MODE 0
Operation
MOV IE,#00010000B;
Disable All Interrupts
SETB 90H;
Set P1.0, Used as RFS
SETB 91H;
Set P1.1, Used as TFS
SETB 93H;
Set P1.3, Used as A0
MOV R1,#003H;
Sets Number of Bytes to Be Read in
A Read Operation
MOV R0,#030H;
Start Address for Where Bytes Will
Be Loaded
MOV R6,#004H;
Use P1.2 as DRDY
WAIT:
NOP;
MOV A,P1;
Read Port 1
ANL A,R6;
Mask Out All Bits Except DRDY
JZ READ;
If Zero Read
SJMP WAIT;
Otherwise Keep Polling
READ:
CLR 90H;
Bring RFS Low
CLR 98H;
Clear Receive Flag
POLL:
JB 98H, READ1
Tests Receive Interrupt Flag
SJMP POLL
READ 1:
MOV A,SBUF;
Read Buffer
RLC A;
Rearrange Data
MOV B.0,C;
Reverse Order of Bits
RLC A; MOV B.1,C; RLC A; MOV B.2,C;
RLC A; MOV B.3,C; RLC A; MOV B.4,C;
RLC A; MOV B.5,C; RLC A; MOV B.6,C;
RLC A; MOV B.7,C;
MOV A,B;
MOV @R0,A;
Write Data to Memory
INC R0;
Increment Memory Location
DEC R1
Decrement Byte Counter
MOV A,R1
JZ END
Jump if Zero
JMP WAIT
Fetch Next Byte
END:
SETB 90H
Bring RFS High
FIN:
SJMP FIN
MODE
Figure 17. AD7711 to 8XC51 Interface
REV. F
–25–
2
AD7711
Table VIII. 8XC51 Code for Writing to the AD7711
MOV SCON,#00000000B;
MOV IE,#10010000B;
MOV IP,#00010000B;
SETB 91H;
SETB 90H;
MOV R1,#003H;
MOV R0,#030H;
MOV A,#00H;
MOV SBUF,A;
WAIT:
JMP WAIT;
INT ROUTINE:
NOP;
MOV A,R1;
JZ FIN;
DEC R1;
MOV A,@R;
INC R0;
RLC A;
DVDD
DVDD
Configure 8051 for MODE 0
Operation & Enable Serial Reception
Enable Transmit Interrupt
Prioritize the Transmit Interrupt
Bring TFS High
Bring RFS High
Sets Number of Bytes to Be Written
in a Write Operation
Start Address in RAM for Bytes
Clear Accumulator
Initialize the Serial Port
SYNC
SS
RFS
PC0
68HC11
PC1
TFS
PC2
DRDY
PC3
A0
SCK
SCLK
MISO
SDATA
MOSI
MODE
AD7711
Figure 18. AD7711 to 68HC11 Interface
Wait for Interrupt
AD7711-ADSP-2105 Interface
An interface circuit between the AD7711 and the ADSP-2105
microprocessor is shown in Figure 19. In this interface, the
AD7711 is configured for its self-clocking mode while the RFS
and TFS pins of the ADSP-2105 are configured as inputs and
the ADSP-2105 serial clock line is also configured as an input.
When the ADSP-2105’s serial clock is configured as an input it
needs a couple of clock pulses to initialize itself correctly before
accepting data. Therefore, the first read from the AD7711 may
not read correct data. In the interface shown, a read operation
to the AD7711 accesses either the output register or the calibration registers. Data cannot be read from the control register.
A write operation always writes to the control or calibration
registers.
Interrupt Subroutine
Load R1 to Accumulator
If Zero Jump to FIN
Decrement R1 Byte Counter
Move Byte into the Accumulator
Increment Address
Rearrange Data—From LSB First
to MSB First
MOV B.0,C; RLC A; MOV B.1,C; RLC A;
MOV B.2,C; RLC A; MOV B.3,C; RLC A;
MOV B.4,C; RLC A; MOV B.5,C; RLC A;
MOV B.6,C; RLC A; MOV B.7,C; MOV A,B;
CLR 93H;
Bring A0 Low
CLR 91H;
Bring TFS Low
MOV SBUF,A;
Write to Serial Port
RETI;
Return from Subroutine
FIN:
SETB 91H;
Set TFS High
SETB 93H;
Set A0 High
RETI;
Return from Interrupt Subroutine
DRDY is used as the frame synchronization pulse for read
operations from the output register and it is decoded with A0 to
drive the RFS inputs of both the AD7711 and the ADSP-2105.
The latched A0 line drives the TFS inputs of both the AD7711
and the ADSP-2105 as well as the AD7711 A0 input.
DVDD
AD7711–68HC11 Interface
MODE
Figure 18 shows an interface between the AD7711 and the
68HC11 microcontroller. The AD7711 is configured for its
external clocking mode while the SPI port is used on the
68HC11 which is in its single chip mode. The DRDY line from
the AD7711 is connected to the Port PC0 input of the 68HC11
so the DRDY line is polled by the 68HC11. The DRDY line
can be connected to the IRQ input of the 68HC11 if an interrupt driven system is preferred. The 68HC11 MOSI and MISO
lines should be configured for wired-or operation. Depending
on the interface configuration, it may be necessary to provide
bidirectional buffers between the 68HC11’s MOSI and MISO
lines.
RFS
RFS
DRDY
TFS
ADSP-2105
AD7711
A0
D
Q
74HC74
DMWR
Q
DR
A0
TFS
SDATA
DT
SCLK
SCLK
Figure 19. AD7711 to ADSP-2105 Interface
The 68HC11 is configured in the master mode with its CPOL
bit set to a logic zero and its CPHA bit set to a logic one. With a
10 MHz master clock on the AD7711, the interface will operate
with all four serial clock rates of the 68HC11.
–26–
REV. F
AD7711
APPLICATIONS
Four-Wire RTD Configurations
Figure 20 shows a four-wire RTD application where the RTD
transducer is interfaced directly to the AD7711. In the four-wire
configuration, there are no errors associated with lead resistances as no current flows in the measurement leads connected
to AIN1(+) and AIN1(–). One of the RTD current sources is
used to provide the excitation current for the RTD. A common
nominal resistance value for the RTD is 100 Ω and, therefore,
the RTD will generate a 20 mV signal which can be handled
directly by the analog input of the AD7711. In the circuit
shown, the second RTD excitation current is used to generate
the reference voltage for the AD7711. This reference voltage is
developed across RREF and applied to the differential reference
inputs. For the nominal reference voltage of +2.5 V, RREF is
12.5 kΩ. This scheme ensures that the analog input voltage span
remains ratiometric to the reference voltage. Any errors in the
analog input voltage due to the temperature drift of the RTD
current source is compensated for by the variation in the reference voltage. The typical matching between the two RTD current sources is less than 3 ppm/°C.
equal (the leads would normally be of the same material and of
equal length) and RTD1 and RTD2 match, then the error voltage across RL2 equals the error voltage across RL1 and no error
voltage is developed between AIN1(+) and AIN1(–). Twice the
voltage is developed across RL3 but since this is a commonmode voltage it will not introduce any errors. The circuit of
Figure 21 shows the reference voltage for the AD7711 derived
from the parts own internal reference.
ANALOG +5V SUPPLY
AVDD
DVDD
REF OUT
2.5V
REFERENCE
200mA
RTD1
RL1
AIN1(+)
RTD
AIN1(–)
INTERNAL
CIRCUITRY
PGA
A = 1–128
RL2
RTD2
200mA
RL3
AD7711
AGND
+5V
AVDD
REF IN(+)
REF IN(–)
DVDD
VSS
DGND
200mA
RTD2
Figure 21. Three-Wire RTD Application with the AD7711
REF IN(+)
RREF
INTERNAL
CIRCUITRY
REF IN(–)
200mA
RTD1
AIN1(+)
RTD
PGA
AIN1(–)
A = 1–128
The circuit of Figure 22 shows an alternate three-wire configuration. In this case, the circuit has the same benefits in terms of
eliminating lead resistance errors as outlined in Figure 21, but it
has the additional benefit that the reference voltage is derived
from one of the current sources. This gives all the benefits of
eliminating RTD tempco errors as outlined in Figure 20. The
voltage on either RTD input can go to within 2 V of the AVDD
supply. The circuit is shown for a +2.5 V reference.
AGND
AD7711
VSS
AVDD DVDD
REF IN(–)
REF IN(+)
DGND
RTD1
12.5kV
Figure 20. Four-Wire RTD Application with the AD7711
Three-Wire RTD Configurations
One possible three-wire configuration using the AD7711 is
outlined in Figure 21. In the three-wire configuration, the lead
resistances will result in errors if only one current source is used
as the 200 µA will flow through RL1 developing a voltage error
between AIN1(+) and AIN1(–). In the scheme outlined below,
the second RTD current source is used to compensate for the
error introduced by the 200 µA flowing through RL1. The second RTD current flows through RL2. Assuming RL1 and RL2 are
INTERNAL
CIRCUITRY
200mA
RL1
AIN1(+)
RTD
AIN1(–)
PGA
A = 1–128
RL2
RTD2
200mA
RL3
AD7711
AGND
DGND
VSS
Figure 22. Alternate Three-Wire Configuration
REV. F
–27–
2
AD7711
OUTLINE DIMENSIONS
Dimensions are shown in inches and (mm).
C1655e–0–7/98
Plastic DIP (N-24)
1.275 (32.30)
1.125 (28.60)
24
13
1
12
PIN 1
0.210
(5.33)
MAX
0.200 (5.05)
0.125 (3.18)
0.280 (7.11)
0.240 (6.10) 0.325 (8.25)
0.300 (7.62) 0.195 (4.95)
0.115 (2.93)
0.060 (1.52)
0.015 (0.38)
0.150
(3.81)
MIN
0.100 (2.54)
BSC
0.022 (0.558)
0.014 (0.356)
0.015 (0.381)
0.008 (0.204)
0.070 (1.77) SEATING
0.045 (1.15) PLANE
Cerdip (Q-24)
0.005 (0.13) MIN
0.098 (2.49) MAX
24
13
1
12
0.310 (7.87)
0.220 (5.59)
0.320 (8.13)
0.290 (7.37)
PIN 1
0.060 (1.52)
0.015 (0.38)
1.280 (32.51) MAX
0.200 (5.08)
MAX
0.150
(3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36)
0.100 (2.54)
BSC
0.070 (1.78) SEATING
0.030 (0.76) PLANE
15°
0°
0.015 (0.38)
0.008 (0.20)
SOIC (R-24)
1
12
0.1043 (2.65)
0.0926 (2.35)
PIN 1
0.0118 (0.30)
0.0040 (0.10)
0.0500
(1.27)
BSC
88
0.0192 (0.49)
08
SEATING
0.0138 (0.35) PLANE 0.0125 (0.32)
0.0091 (0.23)
–28–
PRINTED IN U.S.A.
13
0.4193 (10.65)
0.3937 (10.00)
24
0.2992 (7.60)
0.2914 (7.40)
0.6141 (15.60)
0.5985 (15.20)
0.0291 (0.74)
3 458
0.0098 (0.25)
0.0500 (1.27)
0.0157 (0.40)
REV. F