AD AD8224

Precision, Dual-Channel, JFET Input
Rail-to-Rail Instrumentation Amplifier
AD8224
OUT1
OUT2
–VS
16
15
14
13
AD8224
1
12
–IN2
RG1
2
11
RG2
RG1
3
10
RG2
+IN1
4
9
+IN2
7
8
06286-001
6
–VS
5
REF2
–IN1
Figure 1.
Table 1. In Amps and Difference Amplifiers by Category
High
Perform.
AD82201
AD8221
AD8222
Low
Cost
AD85531
AD6231
APPLICATIONS
Medical instrumentation
Precision data acquisition
Transducer interface
Differential drive for high resolution input ADCs
Remote sensors
+VS
FUNCTIONAL BLOCK DIAGRAM
+VS
Two channels in a small 4 mm × 4 mm LFCSP
Low input currents
25 pA maximum input bias current
2 pA maximum input offset current
High CMRR
94 dB CMRR (minimum), G = 10
84 dB CMRR (minimum) to 10 kHz, G = 10
Excellent ac specifications and low power
1.5 MHz bandwidth (G = 1)
14 nV/√Hz input noise (1 kHz)
Slew rate 2 V/μs
750 μA quiescent current per amplifier (maximum)
Versatility
Rail-to-rail output
Input voltage range to below negative supply rail
4 kV ESD protection
4.5 V to 36 V single supply
±2.25 V to ±18 V dual supply
Gain set with single resistor (G = 1 to 1000)
REF1
FEATURES
1
High
Voltage
AD628
AD629
Mil
Grade
AD620
AD621
AD524
AD526
AD624
Low
Power
AD6271
Digital
Gain
AD82311
AD8250
AD8251
AD85551
AD85561
AD85571
Rail-to-rail output.
GENERAL DESCRIPTION
The AD8224 is the first single-supply junction field effect
transistor (JFET) input instrumentation amplifier available in
the space-saving 16-lead, 4 mm × 4 mm LFCSP. It requires the
same board area as a typical single instrumentation amplifier
yet doubles the channel density and offers a lower cost per
channel without compromising performance.
Designed to meet the needs of high performance, portable
instrumentation, the AD8224 has a minimum common-mode
rejection ratio (CMRR) of 78 dB at dc and a minimum CMRR
of 74 dB at 10 kHz for G = 1. Maximum input bias current is
25 pA and typically remains below 300 pA over the entire
industrial temperature range. Despite the JFET inputs, the
AD8224 typically has a noise corner of only 10 Hz.
With the proliferation of mixed-signal processing, the number of
power supplies required in each system has grown.
Designed to alleviate this problem, the AD8224 can operate on a
±18 V dual supply, as well as on a single +5 V supply. The device’s
rail-to-rail output stage maximizes dynamic range on the low
voltage supplies common in portable applications. Its ability to run
on a single 5 V supply eliminates the need for higher voltage, dual
supplies. The AD8224 draws a maximum of 750 μA of quiescent
current per amplifier, making it ideal for battery-powered devices.
In addition, the AD8224 can be configured as a single-channel,
differential output instrumentation amplifier. Differential
outputs provide high noise immunity, which can be useful when
the output signal must travel through a noisy environment, such
as with remote sensors. The configuration can also be used to
drive differential input ADCs.
For a single-channel version, use the AD8220.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2007 Analog Devices, Inc. All rights reserved.
AD8224
TABLE OF CONTENTS
Features .............................................................................................. 1
Layout .......................................................................................... 20
Applications....................................................................................... 1
Solder Wash................................................................................. 21
Functional Block Diagram .............................................................. 1
Input Bias Current Return Path ............................................... 21
General Description ......................................................................... 1
Input Protection ......................................................................... 21
Revision History ............................................................................... 2
RF Interference ........................................................................... 21
Specifications..................................................................................... 3
Common-Mode Input Voltage Range ..................................... 22
Absolute Maximum Ratings............................................................ 9
Applications Information .............................................................. 23
Thermal Resistance ...................................................................... 9
Driving an Analog-to-Digital Converter ................................ 23
ESD Caution.................................................................................. 9
Differential Output .................................................................... 23
Pin Configuration and Function Descriptions........................... 10
Driving a Differential Input ADC............................................ 24
Typical Performance Characteristics ........................................... 11
Driving Cabling .......................................................................... 24
Theory of Operation ...................................................................... 19
Outline Dimensions ....................................................................... 25
Gain Selection ............................................................................. 19
Ordering Guide .......................................................................... 25
Reference Terminal .................................................................... 20
REVISION HISTORY
1/07—Revision 0: Initial Version
Rev. 0 | Page 2 of 28
AD8224
SPECIFICATIONS
VS+ = +15 V, VS− = −15 V, VREF = 0 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 2 displays the specifications for an
individual instrumentation amplifier configured for a single-ended output or dual instrumentation amplifiers configured for differential
outputs as shown in Figure 59.
Table 2. Individual Amplifier—Single-Ended Configuration or Dual Amplifiers—Differential Output Configuration 2 , VS = ±15 V
Parameter
COMMON-MODE REJECTION RATIO (CMRR)
CMRR DC to 60 Hz with 1 kΩ Source Imbalance
G=1
G = 10
G = 100
G = 1000
CMRR at 10 kHz
G=1
G = 10
G = 100
G = 1000
NOISE
Voltage Noise, 1 kHz
Input Voltage Noise, eni
Output Voltage Noise, eno
RTI, 0.1 Hz to 10 Hz
G=1
Test Conditions
Min
Current Noise
Max
Unit
VCM = ±10 V
78
94
94
94
dB
dB
dB
dB
74
84
84
84
dB
dB
dB
dB
VCM = ±10 V
RTI noise = √(eni2 + (eno/G)2)
VIN+, VIN− = 0 V
VIN+, VIN− = 0 V
14
90
G = 1000
VOLTAGE OFFSET
Input Offset, VOSI
Average TC
Output Offset, VOSO
Average TC
Offset RTI vs. Supply (PSR)
G=1
G = 10
G = 100
G = 1000
INPUT CURRENT (PER CHANNEL)
Input Bias Current
Over Temperature 3
Input Offset Current
Over Temperature3
REFERENCE INPUT
RIN
IIN
Voltage Range
Gain to Output
A Grade
Typ
f = 1 kHz
nV√Hz
nV√Hz
5
μV p-p
0.8
μV p-p
1
fA/√Hz
RTI VOS = (VOSI) + (VOSO/G)
300
10
1200
10
T = −40°C to +85°C
T = −40°C to +85°C
±5 V to ±15 V
86
96
96
96
dB
dB
dB
dB
25
T = −40°C to +85°C
300
2
T = −40°C to +85°C
5
40
VIN+, VIN− = 0 V
70
+VS
−VS
1 ± 0.0001
Rev. 0 | Page 3 of 28
μV
μV/°C
μV
μV/°C
pA
pA
pA
pA
kΩ
μA
V
V/V
AD8224
Parameter
GAIN
Gain Range
Gain Error
G=1
G = 10
G = 100
G = 1000
Gain Nonlinearity
G=1
G = 10
G = 100
G = 1000
G=1
G = 10
G = 100
G=1000
Gain vs. Temperature
G=1
G > 10
INPUT
Impedance (Pin to Ground) 4
Input Operating Voltage Range 5
Over Temperature
OUTPUT
Output Swing
Over Temperature
Output Swing
Over Temperature
Short-Circuit Current
POWER SUPPLY (PER AMPLIFIER)
Operating Range
Quiescent Current
Over Temperature
TEMPERATURE RANGE
For Specified Performance
Operational 7
Test Conditions
G = 1 + (49.4 kΩ/RG)
Min
A Grade
Typ
Max
Unit
1000
V/V
0.06
0.3
0.3
0.3
%
%
%
%
8
5
15
100
15
12
35
180
15
10
25
150
20
20
50
250
ppm
ppm
ppm
ppm
ppm
ppm
ppm
ppm
3
10
−50
ppm/°C
ppm/°C
GΩ||pF
V
V
1
VOUT = ±10 V
VOUT = −10 V to +10 V
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
104||5
VS = ±2.25 V to ±18 V for dual supplies
T = −40°C to +85°C
−VS − 0.1
−VS − 0.1
+VS − 2
+VS − 2.1
RL = 2 kΩ
T = −40°C to +85°C
RL = 10 kΩ
T = −40°C to +85°C
−14.25
−14.3
−14.7
−14.6
+14.25
+14.1
+14.7
+14.6
V
V
V
V
mA
±2.25 6
±18
750
850
V
μA
μA
−40
−40
+85
+125
°C
°C
15
T = −40°C to +85°C
1
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
Refers to the differential configuration shown in Figure 59.
3
Refer to Figure 11 and Figure 12 for the relationship between input current and temperature.
4
Differential and common-mode input impedance can be calculated from the pin impedance: ZDIFF = 2(ZPIN); ZCM = ZPIN/2.
5
The AD8224 can operate up to a diode drop below the negative supply, however the bias current increases sharply. The input voltage range reflects the maximum
allowable voltage where the input bias current is within the specification.
6
At this supply voltage, ensure that the input common-mode voltage is within the input voltage range specification.
7
The AD8224 is characterized from −40°C to +125°C. See the Typical Performance Characteristics section for expected operation in this temperature range.
2
Rev. 0 | Page 4 of 28
AD8224
VS+ = +15 V, VS− = −15 V, VREF = 0 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 3 displays the specifications for the
dynamic performance of each individual instrumentation amplifier.
Table 3. Dynamic Performance of Each Individual Amplifier—Single-Ended Output Configuration, VS = ±15 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
1
Conditions
Min
A Grade
Typ
Max
Unit
1500
800
120
14
kHz
kHz
kHz
kHz
5
4.3
8.1
58
μs
μs
μs
μs
6
4.6
9.6
74
μs
μs
μs
μs
ΔVo = ±10 V step
ΔVo = ±10 V step
2
V/μs
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
VS+ = +15 V, VS− = −15 V, VREF = 0 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 4 displays the specifications for the
dynamic performance of both amplifiers when used in the differential output configuration shown in Figure 59.
Table 4. Dynamic Performance of Both Amplifiers—Differential Output Configuration 2 , VS = ±15 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
Conditions
Min
A Grade
Typ
Max
Unit
1500
800
120
14
kHz
kHz
kHz
kHz
5
4.3
8.1
58
μs
μs
μs
μs
6
4.6
9.6
74
μs
μs
μs
μs
ΔVo = ±10 V step
ΔVo = ±10 V step
2
1
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 k.
2
Refers to the differential configuration shown in Figure 59.
Rev. 0 | Page 5 of 28
V/μs
AD8224
VS + = 5 V, VS− = 0 V, VREF = 2.5 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 5 displays the specifications for an
individual instrumentation amplifier configured for a single-ended output or dual instrumentation amplifiers configured for differential
outputs as shown in Figure 59.
Table 5. Individual Amplifier—Single-Ended Configuration or Dual Amplifiers—Differential Output Configuration 2 , VS =+5 V
Parameter
COMMON-MODE REJECTION RATIO (CMRR)
CMRR DC to 60 Hz with 1 kΩ Source Imbalance
G=1
G = 10
G = 100
G = 1000
CMRR at 10 kHz
G=1
G = 10
G = 100
G = 1000
NOISE
Voltage Noise, 1 kHz
Input Voltage Noise, eni
Output Voltage Noise, eno
RTI, 0.1 Hz to 10 Hz
G=1
G = 1000
Current Noise
VOLTAGE OFFSET
Input Offset, VOSI
Average TC
Output Offset, VOSO
Average TC
Offset RTI vs. Supply (PSR)
G=1
G = 10
G = 100
G = 1000
INPUT CURRENT (PER CHANNEL)
Input Bias Current
Over Temperature 3
Input Offset Current
Over Temperature3
REFERENCE INPUT
RIN
IIN
Voltage Range
Gain to Output
Test Conditions
Min
A Grade
Typ
Max
Unit
VCM = 0 to 2.5 V
78
94
94
94
dB
dB
dB
dB
74
84
84
84
dB
dB
dB
dB
RTI noise = √(eni2 + (eno/G)2)
VIN+, VIN− = 0 V, VREF = 0 V
VIN+, VIN− = 0 V, VREF = 0 V
f = 1 kHz
RTI VOS = (VOSI) + (VOSO/G)
14
90
nV√Hz
nV√Hz
5
0.8
1
μV p-p
μV p-p
fA/√Hz
300
10
1200
10
T = −40°C to +85°C
T = −40°C to +85°C
86
96
96
96
dB
dB
dB
dB
25
T = −40°C to +85°C
300
T = −40°C to +85°C
5
2
40
VIN+, VIN− = 0 V
70
+VS
−VS
1 ± 0.0001
Rev. 0 | Page 6 of 28
μV
μV/°C
μV
μV/°C
pA
pA
pA
pA
kΩ
μA
V
V/V
AD8224
Parameter
GAIN
Gain Range
Gain Error
G=1
G = 10
G = 100
G = 1000
Nonlinearity
G=1
G = 10
G = 100
G = 1000
G=1
G = 10
G = 100
G = 1000
Gain vs. Temperature
G=1
G > 10
INPUT
Impedance (Pin to Ground) 4
Input Voltage Range 5
Over Temperature
OUTPUT
Output Swing
Over Temperature
Output Swing
Over Temperature
Short-Circuit Current
POWER SUPPLY (PER AMPLIFIER)
Operating Range
Quiescent Current
Over Temperature
TEMPERATURE RANGE
For Specified Performance
Operational 6
Test Conditions
G = 1 + (49.4 kΩ/RG)
Min
A Grade
Typ
Max
Unit
1000
V/V
0.06
0.3
0.3
0.3
%
%
%
%
35
35
50
90
35
35
50
175
50
50
75
115
50
50
75
200
ppm
ppm
ppm
ppm
ppm
ppm
ppm
ppm
3
10
−50
ppm/°C
ppm/°C
1
VOUT = 0.3 V to 2.9 V for G = 1
VOUT = 0.3 V to 3.8 V for G > 1
VOUT = 0.3 V to 2.9 V for G = 1
VOUT = 0.3 V to 3.8 V for G > 1
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
104||6
T = −40°C to +85°C
−0.1
−0.1
RL = 2 kΩ
T = −40°C to +85°C
RL = 10 kΩ
T = −40°C to +85°C
0.25
0.3
0.15
0.2
GΩ||pF
+VS − 2 V
+VS − 2.1 V
4.75
4.70
4.85
4.80
V
V
V
V
mA
+4.5
+36
750
850
V
μA
μA
−40
−40
+85
+125
°C
°C
15
T = −40°C to +85°C
1
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
Refers to the differential configuration shown in Figure 59.
3
Refer to Figure 11 and Figure 12 for the relationship between input current and temperature.
4
Differential and common-mode impedance can be calculated from the pin impedance: ZDIFF = 2(ZPIN); ZCM = ZPIN/2.
5
The AD8224 can operate up to a diode drop below the negative supply, but the bias current increases sharply. The input voltage range reflects the maximum
allowable voltage where the input bias current is within the specification.
6
The AD8224 is characterized from −40°C to +125°C. See the Typical Performance Characteristics section for expected operation in that temperature range.
2
Rev. 0 | Page 7 of 28
AD8224
VS + = 5 V, VS− = 0 V, VREF = 2.5 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 6 displays the specifications for the
dynamic performance of each individual instrumentation amplifier.
Table 6. Dynamic Performance of Each Individual Amplifier—Single-Ended Output Configuration, VS = +5 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
1
Conditions
Min
A Grade
Typ
Max
Unit
1500
800
120
14
kHz
kHz
kHz
kHz
ΔVo = 3 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
2.5
2.5
7.5
60
μs
μs
μs
μs
ΔVo = 3 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
3.5
3.5
8.5
75
μs
μs
μs
μs
2
V/μs
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
VS + = 5 V, VS− = 0 V, VREF = 2.5 V, TA = 25°C, G = 1, RL = 2 kΩ 1 unless otherwise noted. Table 7 displays the specifications for the
dynamic performance of both amplifiers when used in the differential output configuration shown in Figure 59.
Table 7. Dynamic Performance of Both Amplifiers—Differential Output Configuration 2 , VS = +5 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
Conditions
Min
A Grade
Typ
Max
Unit
1500
800
120
14
kHz
kHz
kHz
kHz
ΔVo = 3 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
2.5
2.5
7.5
60
μs
μs
μs
μs
ΔVo = 3 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
ΔVo = 4 V Step
3.5
3.5
8.5
75
μs
μs
μs
μs
2
1
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
2
Refers to the differential configuration shown in Figure 59.
Rev. 0 | Page 8 of 28
V/μs
AD8224
ABSOLUTE MAXIMUM RATINGS
Maximum Power Dissipation
Table 8.
Parameter
Supply Voltage
Power Dissipation
Output Short Circuit Current
Input Voltage (Common Mode)
Differential Input Voltage
Storage Temperature
Operating Temperature Range2
Lead Temperature Range (Soldering, 10 sec)
Junction Temperature
Rating
±18 V
See Figure 2
Indefinite1
±Vs
±Vs
−65°C to +130°C
−40°C to +125°C
300°C
130°C
Package Glass Transition Temperature
ESD (Human Body Model)
ESD (Charge Device Model)
ESD (Machine Model)
130°C
4 kV
1 kV
0.4 kV
4.0
3.5
Assumes the load is referenced to mid supply.
Temperature for specified performance is −40°C to +85°C. For performance
to +125°C, see the Typical Performance Characteristics section.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions may affect device reliability.
3.0
2.5
2.0
1.5
1.0
0.5
THERMAL RESISTANCE
θJA = 86°C/W WHEN THERMAL PAD
IS NOT SOLDERED TO BOARD
0
–60
Table 9.
Thermal Pad
Soldered to Board
Not Soldered to Board
θJA = 48°C/W WHEN THERMAL PAD
IS SOLDERED TO BOARD
θJA
48
86
–40
–20
0
20
40
60
80
AMBIENT TEMPERATURE (°C)
Unit
°C/W
°C/W
Figure 2. Maximum Power Dissipation
ESD CAUTION
The θJA values in Table 9 assume a 4-layer JEDEC standard
board. If the thermal pad is soldered to the board, then it is
also assumed it is connected to a plane. θJC at the exposed pad
is 4.4°C/W.
Rev. 0 | Page 9 of 28
100
120
140
06286-002
2
Figure 2 shows the maximum safe power dissipation in the
package vs. the ambient temperature for the LFCSP on a 4-layer
JEDEC standard board.
MAX POWER (W)
1
The maximum safe power dissipation for the AD8224 is limited
by the associated rise in junction temperature (TJ) on the die. At
approximately 130°C, which is the glass transition temperature,
the plastic changes its properties. Even temporarily exceeding
this temperature limit may change the stresses that the package
exerts on the die, permanently shifting the parametric performance
of the amplifiers. Exceeding a temperature of 130°C for an
extended period can result in a loss of functionality.
AD8224
11 RG2
10 RG2
9 +IN2
–VS 8
TOP VIEW
+VS 5
+IN1 4
AD8224
REF1 6
REF2 7
RG1 3
12 –IN2
06286-003
PIN 1
INDICATOR
–IN1 1
RG1 2
15 OUT1
14 OUT2
13 –VS
16 +VS
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 3. Pin Configuration
Table 10. Pin Function Descriptions
Pin Number
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
Mnemonic
−IN1
RG1
RG1
+IN1
+VS
REF1
REF2
−VS
+IN2
RG2
RG2
−IN2
−VS
OUT2
OUT1
+VS
Description
Negative Input Instrumentation Amplifier (In-Amp) 1.
Gain Resistor In-Amp 1.
Gain Resistor In-Amp 1.
Positive Input In-Amp 1.
Positive Supply.
Reference Adjust In-Amp 1.
Reference Adjust In-Amp 2.
Negative Supply.
Positive Input In-Amp 2.
Gain Resistor In-Amp 2.
Gain Resistor In-Amp 2.
Negative Input In-Amp 2.
Negative Supply.
Output In-Amp 2.
Output In-Amp 1.
Positive Supply.
Rev. 0 | Page 10 of 28
AD8224
TYPICAL PERFORMANCE CHARACTERISTICS
@ 25°C, Vs = ±15 V, RL=10 kΩ, unless otherwise noted.
1000
4.5
4.0
3.5
GAIN = 100 BANDWIDTH ROLL-OFF
DELTA VOSI (µV)
100
(nV/ Hz)
GAIN = 1
GAIN = 10
GAIN = 100/GAIN = 1000
10
3.0
2.5
2.0
1.5
1.0
GAIN = 1000 BANDWIDTH ROLL-OFF
1
10
100
1k
100k
10k
FREQUENCY (Hz)
0
0.1
06286-009
1
Figure 4. Voltage Spectral Density vs. Frequency
1
10
TIME (s)
100
1000
06286-012
0.5
Figure 7. Change in Input Offset Voltage vs. Warmup Time
XX
150
GAIN = 1000
130
PSRR (dB)
XXX (X)
BANDWIDTH
LIMITED
GAIN = 100
110
GAIN = 10
90
GAIN = 1
70
50
30
XX
XXX (X)
10
1
10
100
1k
10k
100k
1M
06286-013
XX
XX
1M
06286-014
1s/DIV
06286-010
5µV/DIV
FREQUENCY (Hz)
Figure 5. 0.1 Hz to 10 Hz RTI Voltage Noise (G = 1)
Figure 8. Positive PSRR vs. Frequency, RTI
XX
150
130
XXX (X)
PSRR (dB)
110
GAIN = 1000
90
GAIN = 1
70
GAIN = 10
50
GAIN = 100
30
1s/DIV
XX
XXX (X)
06286-011
1µV/DIV
XX
XX
Figure 6. 0.1 Hz to 10 Hz RTI Voltage Noise (G = 1000)
10
1
10
100
1k
10k
100k
FREQUENCY (Hz)
Figure 9. Negative PSRR vs. Frequency, RTI
Rev. 0 | Page 11 of 28
AD8224
0.3
–15.1V
3
INPUT BIAS
CURRENT ±15
–0.1
INPUT BIAS
CURRENT ±5
–0.3
–5.1V
1
–1
–16
–12
–8
–4
0
BANDWIDTH
LIMITED
GAIN = 100
120
GAIN = 10
100
GAIN = 1
80
60
4
8
–0.5
16
12
COMMON-MODE VOLTAGE (V)
40
10
100
1000
10000
100000
FREQUENCY (Hz)
Figure 10. Input Current vs. Common-Mode Voltage
06286-018
5
140 GAIN = 1000
CMRR (dB)
7
0.1
INPUT OFFSET CURRENT (pA)
INPUT
OFFSET
CURRENT ±5
06286-068
INPUT BIAS CURRENT (pA)
160
INPUT OFFSET
CURRENT ±15
9
Figure 13. CMRR vs. Frequency
160
140 GAIN = 1000
1n
100p
10p
GAIN = 10
100 GAIN = 1
BANDWIDTH
LIMITED
80
IOS
1p
0
25
50
75
100
125
150
TEMPERATURE (°C)
06286-016
40
–25
1
10
100
1000
10000
100000
FREQUENCY (Hz)
06286-019
60
0.1p
–50
GAIN = 100
120
IBIAS
CMRR (dB)
INPUT BIAS CURRENT (A)
10n
Figure 14. CMRR vs. Frequency, 1 kΩ Source Imbalance
Figure 11. Input Bias Current and Offset Current Temperature,
VS = ±15 V, VREF = 0 V
7
10n
6
5
CMRR (µV/V)
IBIAS
100p
10p
IOS
1p
0.1p
3
1
–25
0
25
50
75
100
125
150
TEMPERATURE (°C)
0
–50
–30
–10
10
30
50
70
90
110
TEMPERATURE (°C)
Figure 15. Change in CMRR vs. Temperature, G = 1
Figure 12. Input Bias Current and Offset Current vs. Temperature,
VS = 5 V, VREF = 2.5 V
Rev. 0 | Page 12 of 28
130
06286-020
–50
4
2
06286-017
CURRENT (A)
1n
AD8224
70
60
40
GAIN = 100
20
XXX
GAIN (dB)
30
GAIN = 10
10
0
GAIN = 1
–10
NONLINEARITY (20ppm/DIV)
GAIN = 1000
50
RLOAD = 2kΩ
RLOAD = 10kΩ
VS = ±15V
–30
10k
100k
10M
1M
–10
FREQUENCY (Hz)
–2
XXX
0
2
0
2
4
6
8
10
6
8
10
RLOAD = 2kΩ
RLOAD = 10kΩ
06286-022
VS = ±15V
–4
–2
NONLINEARITY (100ppm/DIV)
NONLINEARITY (10ppm/DIV)
RLOAD = 10kΩ
–6
–4
Figure 19. Gain Nonlinearity, G = 100
RLOAD = 2kΩ
–8
–6
OUTPUT VOLTAGE (V)
Figure 16. Gain vs. Frequency
–10
–8
4
6
8
06286-025
1k
06286-021
–40
100
XXX
06286-024
–20
VS = ±15V
10
–10
–8
–6
VIN (V)
–4
–2
0
2
4
OUTPUT VOLTAGE (V)
Figure 17. Gain Nonlinearity, G = 1
Figure 20. Gain Nonlinearity, G = 1000
RLOAD = 10kΩ
VS = ±15V
–10
–8
–6
–4
–2
0
2
4
6
8
10
+13V
12
6
0
±15V SUPPLIES
–14.8V, +5.5V
+14.9V, +5.5V
+3V
–4.8V, +0.6V
+4.95V, +0.6V
±5V SUPPLIES
–4.8V, –3.3V
–6
+4.95V, –3.3V
–14.8V, –8.3V
+14.9V, –8.3V
–5.3V
–12
–18
–16
–15.3V
–12
–8
–4
0
4
8
12
16
OUTPUT VOLTAGE (V)
VIN (V)
Figure 18. Gain Nonlinearity, G = 10
Figure 21. Input Common-Mode Voltage Range vs. Output Voltage,
G = 1, VREF = 0 V
Rev. 0 | Page 13 of 28
06286-026
INPUT COMMON-MODE VOLTAGE (V)
RLOAD = 2kΩ
06286-023
XXX
NONLINEARITY (10ppm/DIV)
18
AD8224
+3V
3
2
+0.1V, +1.7V
+4.9V, +1.7V
+5V SINGLE SUPPLY,
VREF = +2.5V
1
+0.1V, +0.5V
+4.9V, +0.5V
0
–0.3V
0
1
2
3
4
5
6
OUTPUT VOLTAGE (V)
+14.9V, +5.4V
–4.9V, +0.4V
+4.9V, +0.5V
–4.9V, –4.1V
+4.9V, –4.1V
–5.3V
–14.8V, –9V
+14.9V, –9V
–12
–15.3V
+125°C
2
4
6
8
10
12
–12
–8
–4
0
–40°C
–2
14
16
18
4
8
12
16
+85°C
+25°C
–3
+125°C
–4
+4
+3
+2
+125°C
+1
OUTPUT VOLTAGE (V)
VS –
2
4
6
8
10
+85°C
12
+25°C
14
–40°C
16
18
DUAL SUPPLY VOLTAGE (±V)
Figure 26. Output Voltage Swing vs. Supply Voltage, RL = 2 kΩ, G = 10,
VREF = 0 V
Figure 23. Input Common-Mode Voltage Range vs. Output Voltage,
G = 100, VREF = 0 V
VS+
4
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
–0.2
+3V
3
2
+0.1V, +1.7V
+4.9V, +1.7V
+5V SINGLE SUPPLY,
VREF = +2.5V
1
0
+0.1V, –0.5V
0
1
+4.9V, –0.5V
–0.3V
2
3
4
+125°C
–0.4
+0.4
+125°C
+85°C
+25°C
–40°C
+85°C
+25°C
–40°C
+0.2
5
6
OUTPUT VOLTAGE (V)
Figure 24. Input Common-Mode Voltage Range vs. Output Voltage,
G = 100, VS = 5 V, VREF = 2.5 V
VS –
06286-029
INPUT COMMON-MODE VOLTAGE (V)
+85°C
06286-031
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
+3V
–14.9V, +5.4V
–6
–1
–1
+25°C
–1
±15V SUPPLIES
06286-028
INPUT COMMON-MODE VOLTAGE (V)
–40°C
VS–
VS+
±5V SUPPLIES
–18
–16
+1
Figure 25. Input Voltage Limit vs. Supply Voltage, G = 1, VREF =0 V
+13V
0
+25°C
+85°C
NOTES
1. THE AD8224 CAN OPERATE UP TO A VBE BELOW
THE NEGATIVE SUPPLY, BUT THE BIAS CURRENT
WILL INCREASE SHARPLY.
VOLTAGE SUPPLY (V)
18
6
+125°C
–2
–1
Figure 22. Input Common-Mode Voltage Range vs. Output Voltage,
G = 1, VS = 5 V, VREF = 2.5 V
12
–40°C
2
4
6
8
10
12
14
DUAL SUPPLY VOLTAGE (±V)
16
18
06286-032
–1
–1
–1
06286-030
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
VS+
06286-027
INPUT COMMON-MODE VOLTAGE (V)
4
Figure 27. Output Voltage Swing vs. Supply Voltage, RL = 10 kΩ, G = 10,
VREF = 0 V
Rev. 0 | Page 14 of 28
AD8224
VS+
+25°C
+85°C
5
+125°C
0
+125°C
–5
+85°C
–10
+25°C
–40°C
–15
100
10k
1k
RLOAD (Ω)
Figure 28. Output Voltage Swing vs. Load Resistance VS = ±15 V, VREF = 0 V
–1
+85°C
–2
+2
+125°C
+1
+85°C
+25°C
–40°C
0
2
4
6
8
10
12
14
16
IOUT (mA)
Figure 31. Output Voltage Swing vs. Output Current, VS = 5 V, VREF = 2.5 V
XX
5
–40°C
+25°C
4
NO LOAD
+85°C
47pF
100pF
+125°C
3
XXX (X)
OUTPUT VOLTAGE SWING (V)
+25°C
+125°C
VS –
06286-033
OUTPUT VOLTAGE SWING (V)
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
–40°C
10
06286-036
15
2
+125°C
1
+25°C
+85°C
–40°C
RLOAD (Ω)
20mV/DIV
XX
XX
Figure 29. Output Voltage Swing vs. Load Resistance VS = 5 V, VREF = 2.5 V
XX
XXX (X)
Figure 32. Small Signal Pulse Response for Various Capacitive Loads,
VS = ±15 V, VREF = 0 V
VS+
XX
–40°C
–1
+125°C
–2
47pF
100pF
NO LOAD
+85°C
+25°C
–3
–4
XXX (X)
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
5µs/DIV
06286-037
10k
1k
06286-034
0
100
+4
+3
+2
+125°C
+85°C
+25°C
+1
2
4
6
8
IOUT (mA)
10
12
14
16
20mV/DIV
XX
XX
Figure 30. Output Voltage Swing vs. Output Current, VS = ±15 V, VREF = 0 V
5µs/DIV
XX
XXX (X)
06286-038
–40°C
0
06286-035
VS –
Figure 33. Small Signal Pulse Response for Various Capacitive Loads,
VS = 5 V, VREF = 2.5 V
Rev. 0 | Page 15 of 28
AD8224
30
25
XX
GAIN = 10, 100, 1000
GAIN = 1
5V/DIV
20
XXX (X)
OUTPUT VOLTAGE SWING (V p-p)
35
15
0.002%/DIV
10
8.1μs TO 0.01%
9.6μs TO 0.001%
5
10k
100k
10M
1M
FREQUENCY (Hz)
06286-039
1k
XX
XX
06286-042
20µs/DIV
0
100
XX
XXX (X)
Figure 34. Output Voltage Swing vs. Large Signal Frequency Response
Figure 37. Large Signal Pulse Response and Settle Time, G = 100,
RL = 10 kΩ, VS = ±15 V, VREF = 0 V
XX
XX
5V/DIV
XXX (X)
XXX (X)
5V/DIV
5µs TO 0.01%
6µs TO 0.001%
0.002%/DIV
58μs TO 0.01%
74μs TO 0.001%
200µs/DIV
20µs/DIV
XX
XXX (X)
XX
XX
06286-040
XX
XX
06286-043
0.002%/DIV
XX
XXX (X)
Figure 35. Large Signal Pulse Response and Settle Time, G = 1,
RL = 10 kΩ, VS = ±15 V, VREF = 0 V
Figure 38. Large Signal Pulse Response and Settle Time, G = 1000,
RL = 10 kΩ, VS = ±15 V, VREF = 0 V
XX
XXX
XXX (X)
5V/DIV
0.002%/DIV
4.3μs TO 0.01%
4.6μs TO 0.001%
20mV/DIV
Figure 36. Large Signal Pulse Response and Settle Time, G = 10,
RL = 10 kΩ, VS = ±15 V, VREF = 0 V
4µs/DIV
XXX
06286-044
XX
XXX (X)
06286-041
20µs/DIV
XX
XX
Figure 39. Small Signal Pulse Response, G = 1, RL = 2 kΩ, CL = 100 pF,
VS = ±15 V, VREF = 0 V
Rev. 0 | Page 16 of 28
XXX
XXX
AD8224
06286-045
4µs/DIV
XXX
Figure 43. Small Signal Pulse Response, G = 1, RL = 2 kΩ, CL = 100 pF,
VS = 5 V, VREF = 2.5 V
XXX
XXX
Figure 40. Small Signal Pulse Response, G = 10, RL = 2 kΩ, CL = 100 pF,
VS = ±15 V, VREF = 0 V.
4µs/DIV
XXX
06286-048
20mV/DIV
20mV/DIV
4µs/DIV
4µs/DIV
Figure 41. Small Signal Pulse Response, G = 100, RL = 2 kΩ, C L= 100 pF,
VS = ±15 V, VREF =0 V
Figure 44. Small Signal Pulse Response, G = 10, RL = 2 kΩ, CL = 100 pF,
VS = 5 V, VREF = 2.5 V
XXX
XXX
XXX
XXX
06286-049
20mV/DIV
06286-046
20mV/DIV
20mV/DIV
Figure 42. Small Signal Pulse Response, G = 1000, RL = 2 kΩ, CL = 100 pF,
VS = ±15 V, VREF = 0 V
4µs/DIV
XXX
06286-050
40µs/DIV
XXX
06286-047
20mV/DIV
Figure 45. Small Signal Pulse Response, G = 100, RL = 2 kΩ, CL = 100 pF,
VS = 5 V, VREF = 2.5 V
Rev. 0 | Page 17 of 28
AD8224
180
SOURCE
VOUT = 20V p-p
SOURCE V OUT
SMALLER TO
AVOID SLEW
RATE LIMIT
GAIN=1000
XXX
CHANNEL SEPARATION (dB)
160
20mV/DIV
THERMAL CROSSTALK
VARIES WITH LOAD
140
120
GAIN=1
100
80
XXX
40
06286-051
40µs/DIV
Figure 46. Small Signal Pulse Response, G = 1000,RL = 2 kΩ, CL = 100 pF,
VS = 5 V, VREF = 2.5 V
1
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
06286-069
60
Figure 49. Channel Separation vs. Frequency, RL = 2 kΩ, Source Channel at G = 1
15
60
GAIN = 1000
GAIN = 100
10
SETTLED TO 0.001%
GAIN (dB)
SETTLING TIME (µs)
40
20
GAIN = 10
0
SETTLED TO 0.01%
5
GAIN = 1
0
5
10
15
20
OUTPUT VOLTAGE STEP SIZE (V)
–40
100
06286-052
0
1k
10k
100k
1M
10M
FREQUENCY (Hz)
06286-055
–20
Figure 50. Differential Output Configuration: Gain vs. Frequency
Figure 47. Settling Time vs. Step Size (G = 1) ±15 V, VREF = 0 V
100
100
CMROUT = 20 log
90
VDIFF_OUT
VCM_OUT
70
SETTLED TO 0.001%
CMROUT (dB)
SETTLING TIME (µs)
80
10
SETTLED TO 0.01%
LIMITED BY
MEASUREMENT
SYSTEM
60
50
40
30
20
1
10
100
1000
GAIN (V/V)
Figure 48. Settling Time vs. Gain for a 10 V Step, VS = ±15 V, VREF = 0 V
Rev. 0 | Page 18 of 28
0
1
10
100
1k
10k
100k
FREQUENCY (Hz)
Figure 51. Differential Output Configuration:
Common-Mode Output vs. Frequency
1M
06286-056
1
06286-053
10
AD8224
THEORY OF OPERATION
+VS
+VS
NODE A
R1
24.7kΩ
+VS
+VS
NODE B
RG
–VS
20kΩ
R2
24.7kΩ
–VS
NODE F
+VS
20kΩ
OUTPUT
20kΩ
+VS
+VS
NODE C
J1 Q1
+IN
–VS
A3
VPINCH
NODE E
NODE D
C1
C2
A1
A2
Q2
–IN
J2
VPINCH
+VS
–VS
REF
20kΩ
–VS
–VS
VB
I
06286-057
I
–VS
Figure 52. Simplified Schematic
The AD8224 is a JFET input, monolithic instrumentation amplifier
based on the classic three op amp topology (see Figure 52). Input
Transistor J1 and Input Transistor J2 are biased at a fixed current so
that any input signal forces the output voltages of A1 and A2 to
change accordingly. The input signal creates a current through RG
that flows in R1 and R2 such that the outputs of A1 and A2 provide
the correct, gained signal. Topologically, J1, A1, R1 and J2, A2, R2
can be viewed as precision current feedback amplifiers with a gain
bandwidth of 1.5 MHz. The common-mode voltage and amplified
differential signal from A1 and A2 are applied to a difference
amplifier that rejects the common-mode voltage, but amplifies the
differential signal. The difference amplifier employs 20 kΩ laser
trimmed resistors that result in an in-amp with gain error less than
0.06%. New trim techniques were developed to ensure that CMRR
exceeds 78 dB (G = 1).
Using JFET transistors, the AD8224 offers extremely high input
impedance, extremely low bias currents of 25 pA maximum,
low offset current of 2 pA maximum, and no input bias current
noise. In addition, input offset is less than 300 μV and drift is
less than 10 μV/°C. Ease of use and robustness were considered.
A common problem for instrumentation amplifiers is that at
high gains, when the input is overdriven, an excessive
milliampere input bias current can result and the output can
undergo phase reversal.
Overdriving the input at high gains refers to when the input
signal is within the supply voltages, but the amplifier cannot
output the gained signal. For example, at a gain of 100, driving
the amplifier with 10 V on ±15 V constitutes overdriving the
inputs because the amplifier cannot output 100 V.
The AD8224 has none of these problems; its input bias current
is limited to less than 10 μA and the output does not phase
reverse under overdrive fault conditions.
The AD8224 has extremely low load induced nonlinearity. All
amplifiers that comprise the AD8224 have rail-to-rail output
capability for enhanced dynamic range. The input of the AD8224
can amplify signals with wide common-mode voltages even
slightly lower than the negative supply rail. The AD8224 operates
over a wide supply voltage range. It can operate from either a
single +4.5 V to +36 V supply or a dual ±2.25 V to ±18 V. The
transfer function of the AD8224 is
G =1+
49.4 kΩ
RG
Users can easily and accurately set the gain using a single,
standard resistor. Since the input amplifiers employ a current
feedback architecture, the AD8224 gain bandwidth product
increases with gain, resulting in a system that does not experience
as much bandwidth loss as voltage feedback architectures at
higher gains.
GAIN SELECTION
Placing a resistor across the RG terminals sets the gain of the
AD8224. This is calculated by referring to Table 11 or by using
the following gain equation:
Rev. 0 | Page 19 of 28
RG =
49.4 kΩ
G −1
AD8224
Table 11. Gains Achieved Using 1% Resistors
LAYOUT
1% Standard Table Value of RG (Ω)
49.9 k
12.4 k
5.49 k
2.61 k
1.00 k
499
249
100
49.9
The AD8224 is a high precision device. To ensure optimum
performance at the PC board level, care must be taken in the
design of the board layout. The AD8224 pinout is arranged in
a logical manner to aid in this task.
Calculated Gain
1.990
4.984
9.998
19.93
50.40
100.0
199.4
495.0
991.0
Package Considerations
The AD8224 is available in a 16-lead, 4 mm × 4 mm LFCSP.
Blindly copying the footprint from another 4 mm × 4 mm
LFCSP part is not recommended; it may not have the same
thermal pad size and leads. Refer to the Outline Dimensions
section to verify that the PCB symbol has the correct dimensions.
Space between the leads and thermal pad should be kept as
wide as possible for the best bias current performance. To
maintain the AD8224 ultralow bias current performance, the
thermal pad area can be reduced to extend the gap between the
leads and the pad.
The AD8224 defaults to G = 1 when no gain resistor is used.
The tolerance and gain drift of the RG resistor should be added
to the AD8224 specifications to determine the total gain
accuracy of the system. When the gain resistor is not used,
gain error and gain drift are kept to a minimum.
REFERENCE TERMINAL
The output voltage of the AD8224 is developed with respect to
the potential on the reference terminal. This is useful when the
output signal needs to be offset to a precise midsupply level. For
example, a voltage source can be tied to the REF1 or REF2 pin
to level-shift the output so that the AD8224 can drive a singlesupply ADC. Pin REFx is protected with ESD diodes and
should not exceed either +VS or −VS by more than 0.5 V.
For best performance, source impedance to the REF terminal
should be kept below 1 Ω. As shown in Figure 52 the reference
terminal, REF, is at one end of a 20 kΩ resistor. Additional
impedance at the REF terminal adds to this 20 kΩ resistor and
results in amplification of the signal connected to the positive
input. The amplification from the additional RREF can be
computed by
40 kΩ + RREF
AD8224
The AD8224 has a higher CMRR over frequency than typical
in-amps, which gives it greater immunity to disturbances, such
as line noise and its associated harmonics. A well-implemented
layout is required to maintain this high performance. Input
source impedances should be matched closely. Source resistance
should be placed close to the inputs so that it interacts with as
little parasitic capacitance as possible.
CORRECT
AD8224
VREF
AD8224
VREF
+
OP2177
AD8224
–
–
Figure 53. Driving the Reference Pin
06286-058
VREF
+
To preserve maximum pin compatibility with other dual
instrumentation amplifiers, such as the AD8222, leave the pad
unconnected. This can be done by not soldering the paddle at
all or by soldering the part to a landing that is a not connected
to any other net. For high vibration applications, a landing is
recommended.
Common-Mode Rejection over Frequency
Only the positive signal path is amplified; the negative path is
unaffected. This uneven amplification degrades the amplifier’s
CMRR.
CORRECT
The AD8224 4 mm × 4 mm LFCSP comes with a thermal pad.
This pad is connected internally to +VS. The pad can either be
left unconnected or connected to the positive supply rail.
Because the AD8224 dissipates little power, heat dissipation is
rarely an issue. If improved heat dissipation is desired (for example,
when driving heavy loads), connect the thermal pad to the
positive supply rail. For the best heat dissipation performance,
the positive supply rail should be a plane in the board. See
the Thermal Resistance section for more information.
2 (20 kΩ + RREF )
INCORRECT
Thermal Pad
Parasitics at the RGx pins can also affect CMRR over frequency.
The PCB should be laid out so that the parasitic capacitances at
each pin match. Traces from the gain setting resistor to the RGx
pins should be kept short to minimize parasitic inductance.
Reference
Errors introduced at the reference terminal feed directly to
the output. Take care to tie the REFx pins to the appropriate
local ground.
Rev. 0 | Page 20 of 28
AD8224
Power Supplies
INPUT BIAS CURRENT RETURN PATH
A stable dc voltage should be used to power the instrumentation
amplifier. Noise on the supply pins can adversely affect
performance.
The input bias current of the AD8224 must have a return path
to common. When the source, such as a transformer, cannot
provide a return current path, one should be created, as shown
in Figure 55.
The AD8224 has two positive supply pins (Pin 5 and Pin 16)
and two negative supply pins (Pin 8 and Pin 13). While the part
functions with only one pin from each supply pair connected,
both pins should be connected for specified performance and
optimum reliability.
The AD8224 should be decoupled with 0.1 μF bypass capacitors,
one for each supply. Place the positive supply decoupling
capacitor near Pin 16, and the negative supply decoupling
capacitor near Pin 8. Each supply should also be decoupled with
a 10 μF tantalum capacitor. The tantalum capacitor can be
placed further away from the AD8224 and can generally be
shared by other precision integrated circuits. Figure 54 shows an
example layout.
INPUT PROTECTION
All terminals of the AD8224 are protected against ESD. ESD
protection is guaranteed to 4 kV (human body model). In
addition, the input structure allows for dc overload conditions
a diode drop above the positive supply and a diode drop below
the negative supply. Voltages beyond a diode drop of the
supplies cause the ESD diodes to conduct and enable current to
flow through the diode. Therefore, an external resistor should
be used in series with each of the inputs to limit current for
voltages above +Vs. In either scenario, the AD8224 safely
handles a continuous 6 mA current at room temperature.
For applications where the AD8224 encounters extreme
overload voltages, as in cardiac defibrillators, external series
resistors and low leakage diode clamps, such as BAV199L,
FJH1100, or SP720, should be used.
0.1µF
INCORRECT
CORRECT
+VS
15
14
13
AD8224
AD8224
1
12
2
11
3
10
4
9
RG
RG
5
6
7
AD8224
REF
REF
–VS
–VS
TRANSFORMER
TRANSFORMER
+VS
+VS
C
8
C
R
1
fHIGH-PASS = 2πRC
AD8224
C
REF
AD8224
C
REF
R
0.1µF
06286-059
–VS
–VS
CAPACITIVELY COUPLED
CAPACITIVELY COUPLED
Figure 55. Creating an IBIAS Path
Figure 54. Example Layout
SOLDER WASH
RF INTERFERENCE
The solder process can leave flux and other contaminants on
the board. When these contaminants are between the AD8224
leads and thermal pad, they can create leakage paths that are
larger than the AD8224 bias currents. A thorough washing
process removes these contaminants and restores the device’s
excellent bias current performance.
RF rectification is often a problem in applications where there are
large RF signals. The problem appears as a small dc offset voltage.
The AD8224 by its nature has a 5 pF gate capacitance (CG) at its
inputs. Matched series resistors form a natural low-pass filter that
reduces rectification at high frequency (see Figure 56).
Rev. 0 | Page 21 of 28
06286-060
16
+VS
AD8224
The relationship between external, matched series resistors and the
internal gate capacitance is expressed as follows:
FilterFreqCM =
1
2πRCG
1
2πRCG
+15V
+15V
0.1µF
0.1µF
10µF
+
CC
10µF
1nF
R
+IN
4.02kΩ
CD
–IN
CG
–VS
CC
VOUT
AD8224
–VS
R
REF
–IN
4.02kΩ
CG
VOUT
AD8224
10nF
R
+IN
R
1nF
10µF
0.1µF
REF
+
+
–15V
06286-062
FilterFreq DIFF =
Mismatched CC capacitors result in mismatched low-pass filters.
The imbalance causes the AD8224 to treat what would have
been a common-mode signal as a differential signal. To reduce
the effect of mismatched external CC capacitors, select a value of
CD greater than 10 times CC. This sets the differential filter
frequency lower than the common-mode frequency.
Figure 57. RFI Suppression
10µF
COMMON-MODE INPUT VOLTAGE RANGE
+
–15V
06286-061
0.1µF
Figure 56. RFI Filtering Without External Capacitors
To eliminate high frequency common-mode signals while using
smaller source resistors, a low-pass R-C network can be placed
at the input of the instrumentation amplifier (see Figure 57).
The filter limits the input signal bandwidth according to the
following relationship:
FilterFreq DIFF =
FilterFreqCM =
The 3-op amp architecture of the AD8224 applies gain and then
removes the common-mode voltage. Therefore, internal nodes
in the AD8224 experience a combination of both the gained
signal and the common-mode signal. This combined signal can
be limited by the voltage supplies even when the individual input
and output signals are not. Figure 21 through Figure 24 show the
allowable common-mode input voltage ranges for various
output voltages, supply voltages, and gains.
1
2πR(2 CD + CC + CG )
1
2πR(CC + CG )
Rev. 0 | Page 22 of 28
AD8224
APPLICATIONS INFORMATION
+IN
An instrumentation amplifier is often used in front of an analog-todigital converter to provide CMRR and additional conditioning
such as a voltage level shift and gain (see Figure 58). In this
example, a 2.7 nF capacitor and a 500 Ω resistor create an antialiasing filter for the AD7685. The 2.7 nF capacitor also serves to
store and deliver necessary charge to the switched capacitor input
of the ADC. The 500 Ω series resistor reduces the burden of the
2.7 nF load from the amplifier. However, large source impedance in
front of the ADC can degrade total harmonic distortion (THD).
ADR435
4.7µF
+IN
±50mV
REF
2.7nF
AD7685
–IN
06286-063
+2.5V
Figure 58. Driving an ADC in a Low Frequency Application
REF2
–OUT
Figure 59. Differential Circuit Schematic
Setting the Common-Mode Voltage
The output common-mode voltage is set by the average of +IN2
and REF2. The transfer function is
VCM_OUT = (V+OUT + V−OUT)/2 = (V+IN2 + VREF2)/2
Another differential output topology is shown in Figure 60.
Instead of a second in-amp, ½ of a dual OP2177 op amp creates
the inverted output. Because the OP2177 comes in an MSOP,
this configuration allows the creation of a dual channel,
precision differential output in-amp with little board area.
The differential configuration of the AD8224 has the same
excellent dc precision specifications as the single-ended output
configuration and is recommended for applications in the
frequency range of dc to 1 MHz.
The circuit configuration, outlined in Table 4 and Table 7, refer
to the configuration shown in Figure 59 only. The circuit includes
an RC filter that maintains the stability of the loop.
The transfer function for the differential output is
VDIFF_OUT = V+OUT − V−OUT = (V+IN − V−IN) × G
A common application sets the common-mode output voltage
to the midscale of a differential ADC. In this case, the ADC
reference voltage is sent to the +IN2 terminal, and ground is
connected to the REF2 terminal. This produces a commonmode output voltage of half the ADC reference voltage.
2-Channel Differential Output Using a Dual Op Amp
DIFFERENTIAL OUTPUT
Errors from the op amp are common to both outputs and are,
thus, common mode. Errors from mismatched resistors also
create a common-mode dc offset. Because these errors are
common mode, they are likely to be rejected by the next device
in the signal chain.
where:
G =1+
+IN2
500Ω
AD8224
1.07kΩ
33pF
+IN2 and REF2 have different properties that allow the
reference voltage to be easily set for a wide variety of applications.
+IN2 has high impedance, but cannot swing to the positive
supply rail. REF2 must be driven with a low impedance, but can
go 300 mV beyond the supply rails.
0.1µF
+5V
AD8224
+IN
AD8224
49.4 kΩ
+OUT
–IN
RG
REF
4.99kΩ
4.99kΩ
VREF
+
–
OP2177
–OUT
Figure 60. Differential Output Using Op Amp
Rev. 0 | Page 23 of 28
06286-065
10µF
+OUT
20kΩ
–
+
+5V
+
–IN
+
AD8224
–
For applications where THD performance is critical, the series
resistor needs to be small. At worst, a small series resistor can
load the AD8224, potentially causing the output to overshoot or
ring. In such cases, a buffer amplifier, such as the AD8615 should
be used after the AD8224 to drive the ADC.
RG
06286-064
DRIVING AN ANALOG-TO-DIGITAL CONVERTER
AD8224
+12V
10µF
+
0.1µF
+5V
100pF
NPO
1kΩ
5%
+IN
0.1µF
+OUT
1000pF
AD8224
–IN
+IN2
100pF
NPO
5%
–OUT
REF2
IN–
2.7nF
2.7nF
+
AD7688
GND
REF
10µF
X5R
+12V
+5V REF
10µF
VDD
IN+
806Ω
(DIFF OUT)
1kΩ
806Ω
0.1µF
0.1µF
VIN
–12V
+5V REF
VOUT
0.1µF
ADR435
06286-066
GND
Figure 61. Driving a Differential ADC
DRIVING A DIFFERENTIAL INPUT ADC
However, other converters have less robust inputs and may need
the added protection.
The AD8224 can be configured in differential output mode
to drive a differential analog-to-digital converter. Figure 61
illustrates several of the concepts.
Reference
The 1 kΩ resistor, 1000 pF capacitor, and 100 pF capacitors in
front of the in-amp form a 76 kHz filter. This is the first of two
antialiasing filters in the circuit and helps to reduce the noise of
the system. The 100 pF capacitors protect against commonmode RFI signals. Note that they are 5% COG/NPO types.
These capacitors match well over time and temperature, which
keeps the system’s CMRR high over frequency.
Second Antialiasing Filter
An 806 Ω resistor and a 2.7 nF capacitor are located between
each AD8224 output and ADC input. These components create
a 73 kHz low-pass filter for another stage of antialiasing
protection.
These four elements also isolate the ADC from loading the
AD8224. The 806 Ω resistor shields the AD8224 from the
ADC’s switched capacitor input, which looks like a timevarying load. The 2.7 nF capacitor provides a charge to the
switched capacitor front end of the ADC. If the application
requires a lower frequency antialiasing filter, increase the value
of the capacitor rather than the resistor.
The ADR435 supplies a reference voltage to both the ADC and
the AD8224. Because REF2 on the AD8224 is grounded, the
common-mode output voltage is precisely half the reference
voltage, exactly where it needs to be for the ADC.
DRIVING CABLING
All cables have a certain capacitance per unit length, which
varies widely with cable type. The capacitive load from the cable
may cause peaking in the AD8224 output response. To reduce
peaking, use a resistor between the AD8224 and the cable.
Because cable capacitance and desired output response vary
widely, this resistor is best determined empirically. A good
starting point is 50 Ω.
The AD8224 operates at a low enough frequency that
transmission line effects are rarely an issue; therefore, the
resistor need not match the characteristic impedance of
the cable.
The 806 Ω resistors can also protect an ADC from overvoltages.
Because the AD8224 runs on wider supply voltages than a
typical ADC, there is a possibility of overdriving the ADC. This
is not an issue with a PulSAR® converter, such as the AD7688.
Its input can handle a 130 mA overdrive, which is much higher
than the short-circuit limit of the AD8224.
AD8224
(DIFF OUT)
AD8224
(SINGLE OUT)
06286-067
First Antialiasing Filter
Figure 62. Driving a Cable
Rev. 0 | Page 24 of 28
AD8224
OUTLINE DIMENSIONS
4.00
BSC SQ
0.60 MAX
12 13
3.75
BSC SQ
TOP VIEW
12° MAX
1.00
0.85
0.80
SEATING
0.30
PLANE
0.23
0.18
1
16
EXPOSED
PAD
0.65
BSC
4
8
9
PIN 1
INDICATOR
2.65
2.50 SQ
2.35
5
0.25 MIN
1.95 BCS
0.80 MAX
0.65 TYP
BOTTOM VIEW
0.05 MAX
0.02 NOM
COPLANARITY
0.20 REF
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC.
031006-A
PIN 1
INDICATOR
0.50
0.40
0.30
Figure 63. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-16-13)
Dimensions are shown in millimeters
ORDERING GUIDE
Model
AD8224ACPZ-R7 1
AD8224ACPZ-RL1
AD8224ACPZ-WP1
AD8224-EVALZ1
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Product Description
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
Evaluation Board
Z = Pb-free part.
Rev. 0 | Page 25 of 28
Package Option
CP-16-13
CP-16-13
CP-16-13
AD8224
NOTES
Rev. 0 | Page 26 of 28
AD8224
NOTES
Rev. 0 | Page 27 of 28
AD8224
NOTES
©2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06286-0-1/07(0)
Rev. 0 | Page 28 of 28