a FEATURES High Accuracy 0.02% Max Nonlinearity, 0 V to 2 V RMS Input 0.10% Additional Error to Crest Factor of 3 Wide Bandwidth 8 MHz at 2 V RMS Input 600 kHz at 100 mV RMS Computes: True RMS Square Mean Square Absolute Value dB Output (60 dB Range) Chip Select-Power Down Feature Allows: Analog “3-State” Operation Quiescent Current Reduction from 2.2 mA to 350 A Side-Brazed DIP, Low Cost Cerdip and SOIC PRODUCT DESCRIPTION The AD637 is a complete high accuracy monolithic rms-to-dc converter that computes the true rms value of any complex waveform. It offers performance that is unprecedented in integrated circuit rms-to-dc converters and comparable to discrete and modular techniques in accuracy, bandwidth and dynamic range. A crest factor compensation scheme in the AD637 permits measurements of signals with crest factors of up to 10 with less than 1% additional error. The circuit’s wide bandwidth permits the measurement of signals up to 600 kHz with inputs of 200 mV rms and up to 8 MHz when the input levels are above 1 V rms. As with previous monolithic rms converters from Analog Devices, the AD637 has an auxiliary dB output available to the user. The logarithm of the rms output signal is brought out to a separate pin allowing direct dB measurement with a useful range of 60 dB. An externally programmed reference current allows the user to select the 0 dB reference voltage to correspond to any level between 0.1 V and 2.0 V rms. A chip select connection on the AD637 permits the user to decrease the supply current from 2.2 mA to 350 µA during periods when the rms function is not in use. This feature facilitates the addition of precision rms measurement to remote or hand-held applications where minimum power consumption is critical. In addition when the AD637 is powered down the output goes to a high impedance state. This allows several AD637s to be tied together to form a wide-band true rms multiplexer. The input circuitry of the AD637 is protected from overload voltages that are in excess of the supply levels. The inputs will not be damaged by input signals if the supply voltages are lost. High Precision, Wide-Band RMS-to-DC Converter AD637 FUNCTIONAL BLOCK DIAGRAMS Ceramic DIP (D) and Cerdip (Q) Packages SOIC (R) Package BUFFER BUFFER AD637 1 AD637 1 16 14 ABSOLUTE VALUE 2 ABSOLUTE VALUE 2 13 14 3 BIAS SECTION 12 3 BIAS SECTION 4 SQUARER/DIVIDER 4 11 SQUARER/DIVIDER 13 25kV 12 5 25kV 15 10 5 25kV 9 7 7 11 6 25kV 6 FILTER FILTER 10 8 8 9 The AD637 is available in two accuracy grades (J, K) for commercial (0°C to +70°C) temperature range applications; two accuracy grades (A, B) for industrial (–40°C to +85°C) applications; and one (S) rated over the –55°C to +125°C temperature range. All versions are available in hermetically-sealed, 14-lead side-brazed ceramic DIPs as well as low cost cerdip packages. A 16-lead SOIC package is also available. PRODUCT HIGHLIGHTS 1. The AD637 computes the true root-mean-square, mean square, or absolute value of any complex ac (or ac plus dc) input waveform and gives an equivalent dc output voltage. The true rms value of a waveform is more useful than an average rectified signal since it relates directly to the power of the signal. The rms value of a statistical signal is also related to the standard deviation of the signal. 2. The AD637 is laser wafer trimmed to achieve rated performance without external trimming. The only external component required is a capacitor which sets the averaging time period. The value of this capacitor also determines low frequency accuracy, ripple level and settling time. 3. The chip select feature of the AD637 permits the user to power down the device down during periods of nonuse, thereby, decreasing battery drain in remote or hand-held applications. 4. The on-chip buffer amplifier can be used as either an input buffer or in an active filter configuration. The filter can be used to reduce the amount of ac ripple, thereby, increasing the accuracy of the measurement. REV. E Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1999 AD637–SPECIFICATIONS (@ +25ⴗC, and ⴞ15 V dc unless otherwise noted) Model AD637J/A Typ Min VOUT = TRANSFER FUNCTION CONVERSION ACCURACY Total Error, Internal Trim1 (Fig. 2) T MIN to TMAX vs. Supply, + VIN = +300 mV vs. Supply, – VIN = –300 mV DC Reversal Error at 2 V Nonlinearity 2 V Full Scale2 Nonlinearity 7 V Full Scale Total Error, External Trim Input Offset Voltage Input Current Input Resistance Output Current CHIP SELECT PROVISION (CS) RMS “ON” Level RMS “OFF” Level IOUT of Chip Select CS “LOW” CS “HIGH” On Time Constant Off Time Constant POWER SUPPLY Operating Voltage Range Quiescent Current Standby Current TRANSISTOR COUNT VOUT = avg . (VIN ) ⴞ0.5 ⴞ 0.2 ⴞ2.0 ⴞ 0.3 150 300 0.1 0.02 0.05 ± 0.25 ± 0.05 ⴞ1 ⴞ 0.5 ⴞ6 ⴞ 0.7 150 300 0.25 0.04 0.05 ± 0.5 ± 0.1 Specified Accuracy ± 0.1 ± 1.0 25 25 25 0 to 7 0 to 4 6.4 8 0 to 7 ±15 0 to 4 ±6 ±15 9.6 ±0.5 6.4 8 0 to 7 ± 15 0 to 4 ±6 ± 15 9.6 ± 0.2 6.4 8 Units 2 Specified Accuracy ± 0.1 ± 1.0 mV ± % of Reading mV ± % of Reading µV/V µV/V % of Reading % of FSR % of FSR mV ± % of Reading % of Reading % of Reading ms/µF CAV ± 15 V rms V p-p ±6 V rms V p-p ± 15 9.6 ± 0.5 V p-p kΩ mV 11 66 200 11 66 200 11 66 200 kHz kHz kHz 150 1 8 150 1 8 150 1 8 kHz MHz MHz ±0.05 ⴞ1 ⴞ0.089 ± 0.04 ⴞ0.5 ⴞ0.056 ± 0.04 ⴞ1 ⴞ0.07 mV mV/°C 0 to +12.0 +13.5 0 to +12.0 +13.5 0 to +12.0 +13.5 V 0 to +2 6 0 to +2 6 0 to +2 6 V mA mA Ω kΩ 5 1 –VS to (+V S – 2.5 V) +2.2 +2.2 20 0.5 100 20 0.5 100 ±0.5 –3 +0.33 –0.033 20 ± 0.3 –3 +0.33 –0.033 20 ± 0.5 –3 +0.33 –0.033 20 ±0.8 ±2 108 80 100 5 1 –VS to (+V S – 2.5 V) ⴞ2 ⴞ10 ± 0.5 ±2 108 80 100 0 to +10 25 ±0.2 5 1 –VS to (+VS – 2.5 V) ⴞ1 ⴞ5 (+5 mA, –130 µA) 20 1 5 20 +2.2 20 0.5 100 (+5 mA, –130 µA) Short Circuit Current Small Signal Bandwidth Slew Rate5 DENOMINATOR INPUT Input Range Input Resistance Offset Voltage 2 Max Specified Accuracy ±0.1 ±1.0 ±0.5 ± 0.1 dB OUTPUT Error, VIN 7 mV to 7 V rms, 0 dB = 1 V rms Scale Factor Scale Factor Temperature Coefficient BUFFER AMPLIFIER Input Output Voltage Range ⴞ1 ⴞ 0.5 ⴞ3.0 ⴞ 0.6 150 300 0.25 0.04 0.05 avg . (VIN ) AD637S Typ Min 30 100 FREQUENCY RESPONSE4 Bandwidth for 1% Additional Error (0.09 dB) VIN = 20 mV VIN = 200 mV VIN = 2 V ±3 dB Bandwidth VIN = 20 mV VIN = 200 mV VIN = 2 V IREF for 0 dB = 1 V rms IREF Range VOUT = Max 30 100 AVERAGING TIME CONSTANT OUTPUT CHARACTERISTICS Offset Voltage vs. Temperature Voltage Swing, ±15 V Supply, 2 kΩ Load Voltage Swing, ±3 V Supply, 2 kΩ Load Output Current Short Circuit Current Resistance, Chip Select “High” Resistance, Chip Select “Low” AD637K/B Typ Min 2 30 100 ERROR VS. CREST FACTOR3 Crest Factor 1 to 2 Crest Factor = 3 Crest Factor = 10 INPUT CHARACTERISTICS Signal Range, ±15 V Supply Continuous RMS Level Peak Transient Input Signal Range, ±5 V Supply Continuous rms Level Peak Transient Input Maximum Continuous Nondestructive Input Level (All Supply Voltages) Input Resistance Input Offset Voltage avg . (VIN ) Max 0 to +10 25 ± 0.2 20 ⴞ2 ⴞ10 V mV nA Ω (+5 mA, –130 µA) 20 1 5 30 ±0.5 ± 0.8 ±2 108 80 100 dB mV/dB % of Reading/°C dB/°C µA µA 30 ± 0.5 20 20 1 5 mA MHz V/µs 0 to +10 25 30 ± 0.2 ± 0.5 V kΩ mV Open or +2.4 V < VC < +V S VC < +0.2 V Open or +2.4 V < VC < +VS VC < +0.2 V Open or +2.4 V < VC < +VS VC < +0.2 V 10 Zero 10 µs + ((25 kΩ) × CAV ) 10 µs + ((25 kΩ) × CAV ) 10 Zero 10 µs + ((25 kΩ) × CAV ) 10 µs + ((25 kΩ) × CAV ) 10 Zero 10 µs + ((25 kΩ) × CAV ) 10 µs + ((25 kΩ) × CAV ) µA ⴞ3.0 V mA µA ⴞ3.0 2.2 350 ⴞ18 3 450 ⴞ3.0 2.2 350 107 107 –2– ⴞ18 3 450 2.2 350 ⴞ18 3 450 107 REV. E AD637 NOTES 1 Accuracy specified 0-7 V rms dc with AD637 connected as shown in Figure 2. 2 Nonlinearity is defined as the maximum deviation from the straight line connecting the readings at 10 mV and 2 V. 3 Error vs. crest factor is specified as additional error for 1 V rms. 4 Input voltages are expressed in volts rms. % are in % of reading. 5 With external 2 kΩ pull down resistor tied to –V S . Specifications subject to change without notice. Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to calculate outgoing quality levels. All min and max specifications are guaranteed, although only those shown in boldface are tested on all production units. ABSOLUTE MAXIMUM RATINGS ORDERING GUIDE ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 500 V Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V dc Internal Quiescent Power Dissipation . . . . . . . . . . . . 108 mW Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C Lead Temperature Range (Soldering 10 secs) . . . . . . . +300°C Rated Operating Temperature Range AD637J, K . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C AD637A, B . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C AD637S, 5962-8963701CA . . . . . . . . . . . –55°C to +125°C Model Temperature Range Package Description Package Option AD637AR AD637BR AD637AQ AD637BQ AD637JD AD637JD/+ AD637KD AD637KD/+ AD637JQ AD637KQ AD637JR AD637JR-REEL AD637JR-REEL7 AD637KR AD637SD AD637SD/883B AD637SQ/883B AD637SCHIPS 5962-8963701CA* –40°C to +85°C –40°C to +85°C –40°C to +85°C –40°C to +85°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C 0°C to +70°C –55°C to +125°C –55°C to +125°C –55°C to +125°C 0°C to +70°C –55°C to +125°C SOIC SOIC Cerdip Cerdip Side Brazed Ceramic DIP Side Brazed Ceramic DIP Side Brazed Ceramic DIP Side Brazed Ceramic DIP Cerdip Cerdip SOIC SOIC SOIC SOIC Side Brazed Ceramic DIP Side Brazed Ceramic DIP Cerdip Die Cerdip R-16 R-16 Q-14 Q-14 D-14 D-14 D-14 D-14 Q-14 Q-14 R-16 R-16 R-16 R-16 D-14 D-14 Q-14 Q-14 *A standard microcircuit drawing is available. FILTER/AMPLIFIER BUFF OUT ONE QUADRANT SQUARER/DIVIDER BUFF IN CAV 24kV +VS BUFFER AMPLIFIER A5 RMS OUT A4 I4 dB OUT I1 24kV COM Q4 Q1 ABSOLUTE VALUE VOLTAGE – CURRENT CONVERTER 6kV Q5 Q2 6kV Q3 BIAS I3 A3 24kV A2 12kV 125V VIN CS DEN INPUT OUTPUT OFFSET AD637 A1 –VS Figure 1. Simplified Schematic CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD637 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. E –3– WARNING! ESD SENSITIVE DEVICE AD637 the AD637 can be ac coupled through the addition of a nonpolar capacitor in series with the input as shown in Figure 2. FUNCTIONAL DESCRIPTION The AD637 embodies an implicit solution of the rms equation that overcomes the inherent limitations of straightforward rms computation. The actual computation performed by the AD637 follows the equation BUFFER AD637 1 V 2 V rms = Avg IN V rms Figure 1 is a simplified schematic of the AD637, it is subdivided into four major sections; absolute value circuit (active rectifier), square/divider, filter circuit and buffer amplifier. The input voltage VIN which can be ac or dc is converted to a unipolar current I1 by the active rectifier A1, A2. I1 drives one input of the squarer divider which has the transfer function 2 I I4 = 1 I3 The output current of the squarer/divider, I4 drives A4 which forms a low-pass filter with the external averaging capacitor. If the RC time constant of the filter is much greater than the longest period of the input signal than A4s output will be proportional to the average of I4. The output of this filter amplifier is used by A3 to provide the denominator current I3 which equals Avg. I4 and is returned to the squarer/divider to complete the implicit rms computation. 14 NC ABSOLUTE VALUE 2 3 12 BIAS SECTION 4 SQUARER/DIVIDER OPTIONAL AC COUPLING VIN CAPACITOR 13 NC 11 +VS 10 –VS 25kV 5 25kV 6 VO = 9 VIN3 CAV FILTER 7 8 Figure 2. Standard RMS Connection The performance of the AD637 is tolerant of minor variations in the power supply voltages, however, if the supplies being used exhibit a considerable amount of high frequency ripple it is advisable to bypass both supplies to ground through a 0.1 µF ceramic disc capacitor placed as close to the device as possible. The output signal range of the AD637 is a function of the supply voltages, as shown in Figure 3. The output signal can be used buffered or nonbuffered depending on the characteristics of the load. If no buffer is needed, tie buffer input (Pin 1) to common. The output of the AD637 is capable of driving 5 mA into a 2 kΩ load without degrading the accuracy of the device. I 2 I 4 = Avg 1 = I1 rms I 4 and VOUT = VIN rms If the averaging capacitor is omitted, the AD637 will compute the absolute value of the input signal. A nominal 5 pF capacitor should be used to insure stability. The circuit operates identically to that of the rms configuration except that I3 is now equal to I4 giving 20 2 I1 I4 MAX VOUT – Volts 2kV Load I4 = I 4 = I1 The denominator current can also be supplied externally by providing a reference voltage, VREF, to Pin 6. The circuit operates identically to the rms case except that I3 is now proportional to VREF. Thus: 2 I I 4 = Avg 1 I3 and 2 V V O = IN V DEN This is the mean square of the input signal. 15 10 5 0 0 63 65 610 615 SUPPLY VOLTAGE – DUAL SUPPLY – Volts 618 Figure 3. AD637 Max VOUT vs. Supply Voltage CHIP SELECT STANDARD CONNECTION The AD637 includes a chip select feature which allows the user to decrease the quiescent current of the device from 2.2 mA to 350 µA. This is done by driving the CS, Pin 5, to below 0.2 V dc. Under these conditions, the output will go into a high impedance state. In addition to lowering power consumption, this feature permits bussing the outputs of a number of AD637s to form a wide bandwidth rms multiplexer. If the chip select is not being used, Pin 5 should be tied high. The AD637 is simple to connect for a majority of rms measurements. In the standard rms connection shown in Figure 2, only a single external capacitor is required to set the averaging time constant. In this configuration, the AD637 will compute the true rms of any input signal. An averaging error, the magnitude of which will be dependent on the value of the averaging capacitor, will be present at low frequencies. For example, if the filter capacitor CAV, is 4 µF this error will be 0.1% at 10 Hz and increases to 1% at 3 Hz. If it is desired to measure only ac signals, –4– REV. E AD637 functions of input signal frequency f, and the averaging time constant τ (τ: 25 ms/µF of averaging capacitance). As shown in Figure 6, the averaging error is defined as the peak value of the ac component, ripple, plus the value of the dc error. OPTIONAL TRIMS FOR HIGH ACCURACY The AD637 includes provisions to allow the user to trim out both output offset and scale factor errors. These trims will result in significant reduction in the maximum total error as shown in Figure 4. This remaining error is due to a nontrimmable input offset in the absolute value circuit and the irreducible nonlinearity of the device. The trimming procedure on the AD637 is as follows: The peak value of the ac ripple component of the averaging error is defined approximately by the relationship: 50 in % of reading where (t > 1/f) 6.3 τf l. Ground the input signal, VIN and adjust R1 to give 0 V output from Pin 9. Alternatively R1 can be adjusted to give the correct output with the lowest expected value of VIN. EO IDEAL EO DC ERROR = AVERAGE OF OUTPUT–IDEAL 2. Connect the desired full scale input to VIN, using either a dc or a calibrated ac signal, trim R3 to give the correct output at Pin 9, i.e., 1 V dc should give l.000 V dc output. Of course, a 2 V peak-to-peak sine wave should give 0.707 V dc output. Remaining errors are due to the nonlinearity. AVERAGE ERROR DOUBLE-FREQUENCY RIPPLE TIME 5.0 Figure 6. Typical Output Waveform for a Sinusoidal Input AD637K MAX ERROR – mV 2.5 This ripple can add a significant amount of uncertainty to the accuracy of the measurement being made. The uncertainty can be significantly reduced through the use of a post filtering network or by increasing the value of the averaging capacitor. INTERNAL TRIM AD637K EXTERNAL TRIM 0 The dc error appears as a frequency dependent offset at the output of the AD637 and follows the equation: 1 in % of reading 0.16 + 6.4τ 2 f 2 Since the averaging time constant, set by CAV , directly sets the time that the rms converter “holds” the input signal during computation, the magnitude of the dc error is determined only by CAV and will not be affected by post filtering. 2.5 AD637K: 0.5mV 60.2% 0.25mV 60.05% EXTERNAL 5.0 0 0.5 1.0 INPUT LEVEL – Volts 2.0 1.5 Figure 4. Max Total Error vs. Input Level AD637K Internal and External Trims DC ERROR OR RIPPLE % OF READING BUFFER 100 AD637 14 1 R4 147V ABSOLUTE VALUE 2 3 +VS OUTPUT R1 OFFSET 50kV ADJUST –VS VIN 12 BIAS SECTION R2 1MV 13 4 SQUARER/DIVIDER 11 +VS 10 –VS 25kV 5 9 FILTER 7 + 8 CAV V rms OUT 1.0 DC ERROR 100 1k SINEWAVE INPUT FREQUENCY – Hz 10k Figure 7. Comparison of Percent DC Error to the Percent Peak Ripple over Frequency Using the AD637 in the Standard RMS Connection with a 1 × µ F CAV R3 1kV The ac ripple component of averaging error can be greatly reduced by increasing the value of the averaging capacitor. There are two major disadvantages to this: first, the value of the averaging capacitor will become extremely large and second, the settling time of the AD637 increases in direct proportion to the value of the averaging capacitor (Ts = 115 ms/µF of averaging capacitance). A preferable method of reducing the ripple is through the use of the post filter network, shown in Figure 8. This network can be used in either a one or two pole configuration. For most applications the single pole filter will give the best overall compromise between ripple and settling time. SCALE FACTOR ADJUST, 62% Figure 5. Optional External Gain and Offset Trims CHOOSING THE AVERAGING TIME CONSTANT The AD637 will compute the true rms value of both dc and ac input signals. At dc the output will track the absolute value of the input exactly; with ac signals the AD637’s output will approach the true rms value of the input. The deviation from the ideal rms value is due to an averaging error. The averaging error is comprised of an ac and dc component. Both components are REV. E PEAK RIPPLE 0.1 10 25kV 6 10 –5– AD637 100 100 +VS 10 –VS 25kV 9 FILTER 8 + CAV 1.0 1.0 VALUES FOR CAV AND 1% SETTLING TIME 0.1 FOR STATED % OF READING AVERAGING ERROR* ACCURACY 62% DUE TO COMPONENT TOLERANCE 0.1 * %dc ERROR + %RIPPLE (Peak) 0.01 1 10 100 1k INPUT FREQUENCY – Hz 10k 0.01 100k Figure 9a. 24kV Figure 8. Two Pole Sallen-Key Filter 10 * %dc ERROR + % PEAK RIPPLE ACCURACY 620% DUE TO COMPONENT TOLERANCE R O R R ER O R % 01 R ER 0. O R R 1% 0. ER RO ER 1.0 1.0 5% Figure 9b shows the relationship between averaging error, signal frequency settling time and averaging capacitor value. This graph is drawn for filter capacitor values of 3.3 times the averaging capacitor value. This ratio sets the magnitude of the ac and dc errors equal at 50 Hz. As an example, by using a 1 µF averaging capacitor and a 3.3 µF filter capacitor, the ripple for a 60 Hz input signal will be reduced from 5.3% of reading using the averaging capacitor alone to 0.15% using the single pole filter. This gives a factor of thirty reduction in ripple and yet the settling time would only increase by a factor of three. The values of CAV and C2, the filter capacitor, can be calculated for the desired value of averaging error and settling time by using Figure 9b. 10 1% Figure 9a shows values of CAV and the corresponding averaging error as a function of sine-wave frequency for the standard rms connection. The 1% settling time is shown on the right side of the graph. 100 VALUES OF CAV, C2 AND 1% SETTLING TIME FOR STATED % OF READING AVERAGING ERROR* FOR 1 POLE POST FILTER FOR 1 POLE FILTER, SHORT RX AND REMOVE C3 0.1 0.1 0.01 1 10 100 1k INPUT FREQUENCY – Hz 10k 0.01 100k Figure 9b. R O R R ER O % R 01 R ER 0. O R R ER RO ER 1.0 0.1 0.01 Figure 9c can be used to determine the required value of CAV, C2 and C3 for the desired level of ripple and settling time. * %dc ERROR + % PEAK RIPPLE ACCURACY 620% DUE TO COMPONENT TOLERANCE 10 1.0 1% 0. For applications that are extremely sensitive to ripple, the two pole configuration is suggested. This configuration will minimize capacitor values and settling time while maximizing performance. 10 5% The symmetry of the input signal also has an effect on the magnitude of the averaging error. Table I gives practical component values for various types of 60 Hz input signals. These capacitor values can be directly scaled for frequencies other than 60 Hz, i.e., for 30 Hz double these values, for 120 Hz they are halved. 100 VALUES OF CAV, C2 AND C3 AND 1% SETTLING TIME FOR STATED % OF READING AVERAGING ERROR* 2 POLL SALLEN-KEY FILTER 1% REQUIRED CAV (AND C2 + C3) C2 = C3 = 2.2 3 CAV 100 1 10 100 1k INPUT FREQUENCY – Hz 0.1 10k FOR 1% SETTLING TIME IN SECONDS MULTIPLY READING BY 0.365 + C2 FOR 1% SETTLING TIME IN SECONDS MULTIPLY READING BY 0.400 100 REQUIRED CAV (AND C2) C2 = 3.3 3 CAV RX 24kV R O R ER 11 R O R ER 7 C3 25kV CHIP SELECT 5 dB SQUARER/DIVIDER 10 R O R ER 4 DENOMINATOR 6 INPUT + 12 NC BIAS SECTION R O R ER 3 % 10 OUTPUT OFFSET 1% 0. ANALOG COM 10 SIGNAL INPUT 13 1% ABSOLUTE VALUE NC 2 RMS OUTPUT FOR 1% SETTLING TIME IN SECONDS MULTIPLY READING BY 0.115 14 1 BUFFER OUTPUT % 01 0. BUFFER INPUT AD637 REQUIRED CAV – mF BUFFER 0.01 100k Figure 9c. –6– REV. E AD637 Table I. Practical Values of CAV and C2 for Various Input Waveforms Input Waveform and Period Absolute Value Circuit Waveform and Period Recommended CAV and C2 Values for 1% Averaging Minimum Error@60Hz with T = 16.6ms R 3 CAV 1% Recommended Recommended Settling Time Standard Standard Time Constant Value C Value C2 AV 1/2T T A 0V 1/2T 0.47mF 1.5mF 181ms T 0.82mF 2.7mF 325ms Symmetrical Sine Wave T T B AC MEASUREMENT ACCURACY AND CREST FACTOR Crest factor is often overlooked in determining the accuracy of an ac measurement. Crest factor is defined as the ratio of the peak signal amplitude to the rms value of the signal (C.F. = Vp/ V rms). Most common waveforms, such as sine and triangle waves, have relatively low crest factors (≤2). Waveforms which resemble low duty cycle pulse trains, such as those occurring in switching power supplies and SCR circuits, have high crest factors. For example, a rectangular pulse train with a 1% duty cycle has a crest factor of 10 (C.F. = 1 η ). T 0V Sine Wave with dc Offset T C 0 T T2 Vp T2 0V 10(T – T2) 6.8mF 22mF h = DUTY CYCLE = CF = 1/ e0 eIN (rms) = 1 Volt rms 100mF 2.67sec 100ms T h 10 Pulse Train Waveform T T T2 T2 CAV = 22mF 10(T – 2T2) 5.6mF 18mF 2.17sec 0V INCREASE IN ERROR – % D FREQUENCY RESPONSE The frequency response of the AD637 at various signal levels is shown in Figure 10. The dashed lines show the upper frequency limits for 1%, 10% and ± 3 dB of additional error. For example, note that for 1% additional error with a 2 V rms input the highest frequency allowable is 200 kHz. A 200 mV signal can be measured with 1% error at signal frequencies up to 100 kHz. 1.0 CF = 10 0.1 CF = 3 0.01 10 VOUT – Volts 1000 Figure 12 is a curve of additional reading error for the AD637 for a 1 volt rms input signal with crest factors from 1 to 11. A rectangular pulse train (pulsewidth 100 µs) was used for this test since it is the worst-case waveform for rms measurement (all 1V RMS INPUT 1% 0.1 10 100 PULSEWIDTH – ms Figure 11. AD637 Error vs. Pulsewidth Rectangular Pulse 7V RMS INPUT 2V RMS INPUT 1 1 10% 100mV RMS INPUT 63dB +1.5 1k 10mV RMS INPUT 10k 100k 1M INPUT FREQUENCY – Hz +1.0 INCREASE IN ERROR – % 0.01 10M Figure 10. Frequency Response To take full advantage of the wide bandwidth of the AD637 care must be taken in the selection of the input buffer amplifier. To insure that the input signal is accurately presented to the converter, the input buffer must have a –3 dB bandwidth that is wider than that of the AD637. A point that should not be overlooked is the importance of slew rate in this application. For example, the minimum slew rate required for a 1 V rms 5 MHz sine-wave input signal is 44 V/µs. The user is cautioned that this is the minimum rising or falling slew rate and that care must be exercised in the selection of the buffer amplifier as some amplifiers exhibit a two-to-one difference between rising and falling slew rates. The AD845 is recommended as a precision input buffer. REV. E +0.5 0 +0.5 POSITIVE INPUT PULSE CAV = 22mF –1.0 –1.5 1 2 3 4 5 6 7 CREST FACTOR 8 9 10 Figure 12. Additional Error vs. Crest Factor –7– 11 MAGNITUDE OF ERROR – % OF rms LEVEL AD637 2.0 DB CALIBRATION 1.8 1. 2. 3. 4. 1.6 1.4 1.2 CF = 10 1.0 Set VIN = 1.00 V dc or 1.00 V rms Adjust R1 for 0 dB out = 0.00 V Set VIN = 0.1 V dc or 0.10 V rms Adjust R2 for dB out = – 2.00 V Any other dB reference can be used by setting VIN and R1 accordingly. 0.8 CF = 7 0.6 LOW FREQUENCY MEASUREMENTS 0.4 0.2 If the frequencies of the signals to be measured are below 10 Hz, the value of the averaging capacitor required to deliver even 1% averaging error in the standard rms connection becomes extremely large. The circuit shown in Figure 15 shows an alternative method of obtaining low frequency rms measurements. The averaging time constant is determined by the product of R and CAV1, in this circuit 0.5 s/µF of CAV. This circuit permits a 20:1 reduction in the value of the averaging capacitor, permitting the use of high quality tantalum capacitors. It is suggested that the two pole Sallen-Key filter shown in the diagram be used to obtain a low ripple level and minimize the value of the averaging capacitor. CF = 3 0.0 0.5 1.0 VIN – V rms 1.5 2.0 Figure 13. Error vs. RMS Input Level for Three Common Crest Factors the energy is contained in the peaks). The duty cycle and peak amplitude were varied to produce crest factors from l to 10 while maintaining a constant 1 volt rms input amplitude. CONNECTION FOR dB OUTPUT Another feature of the AD637 is the logarithmic or decibel output. The internal circuit which computes dB works well over a 60 dB range. The connection for dB measurement is shown in Figure 14. The user selects the 0 dB level by setting R1 for the proper 0 dB reference current (which is set to exactly cancel the log output current from the squarer/divider circuit at the desired 0 dB point). The external op amp is used to provide a more convenient scale and to allow compensation of the +0.33%/°C temperature drift of the dB circuit. The special T.C. resistor R3 is available from Tel Labs in Londenderry, New Hampshire (model Q-81) and from Precision Resistor Inc., Hillside, N.J. (model PT146). If the frequency of interest is below 1 Hz, or if the value of the averaging capacitor is still too large, the 20:1 ratio can be increased. This is accomplished by increasing the value of R. If this is done it is suggested that a low input current, low offset voltage amplifier like the AD548 be used instead of the internal buffer amplifier. This is necessary to minimize the offset error introduced by the combination of amplifier input currents and the larger resistance. R2 33.2kV SIGNAL INPUT dB SCALE FACTOR ADJUST 5kV +VS BUFFER BUFFER INPUT AD637 1 14 ABSOLUTE VALUE NC 2 ANALOG COM OUTPUT OFFSET 3 4 SQUARER/DIVIDER * 1kV AD707JN COMPENSATED dB OUTPUT + 100mV/dB –VS 11 +VS 25kV CHIP SELECT 5 dB SIGNAL INPUT 12 NC BIAS SECTION DENOMINATOR 6 INPUT 13 R3 60.4V BUFFER OUTPUT 10 –VS RMS OUTPUT 25kV 9 FILTER 7 8 + 1mF CAV 10kV +VS R1 500kV +2.5 VOLTS *1kV + 3500ppm TC RESISTOR TEL LAB Q81 PRECISION RESISTOR PT146 OR EQUIVALENT AD508J 0dB ADJUST Figure 14. dB Connection –8– REV. E AD637 1mF NOTE: VALUES CHOSEN TO GIVE 0.1% AVERAGING ERROR @ 1Hz BUFFER 3.3MV 1 OUTPUT OFFSET 50kV ADJUST AD548JN FILTERED V rms OUTPUT 14 ABSOLUTE VALUE NC 2 +VS 3.3MV 1mF AD637 +VS 3 –VS SIGNAL INPUT 13 6.8MV 12 NC BIAS SECTION 1MV 4 SQUARER/DIVIDER 1000pF 11 +VS 10 –VS 25kV 5 –VS 25kV 9 6 FILTER 7 499kV CAV1 3.3mF 8 + VIN2 V rms 100mF CAV 1% R Figure 15. AD637 as a Low Frequency RMS Converter VECTOR SUMMATION EXPANDABLE Vector summation can be accomplished through the use of two AD637s as shown in Figure 16. Here the averaging capacitors are omitted (nominal 100 pF capacitors are used to insure stability of the filter amplifier), and the outputs are summed as shown. The output of the circuit is 2 V O = V X +V Y BUFFER 14 ABSOLUTE VALUE 2 3 2 13 VX IN 12 BIAS SECTION This concept can be expanded to include additional terms by feeding the signal from Pin 9 of each additional AD637 through a 10 kΩ resistor to the summing junction of the AD711, and tying all of the denominator inputs (Pin 6) together. 4 SQUARER/DIVIDER 11 +VS 10 –VS 25kV 5 25kV 9 6 If CAV is added to IC1 in this configuration, the output is 100pF FILTER 7 2 AD637 1 8 5pF 2 V X +V Y . If the averaging capacitor is included on both 10kV BUFFER IC1 and IC2, the output will be 2 2 V X +V Y . AD637 1 This circuit has a dynamic range of 10 V to 10 mV and is limited only by the 0.5 mV offset voltage of the AD637. The useful bandwidth is 100 kHz. 10kV 14 ABSOLUTE VALUE 2 3 13 VX IN AD711K 12 BIAS SECTION 4 SQUARER/DIVIDER 11 10kV +VS 20kV 25kV 5 10 –VS 25kV 9 6 100pF 7 FILTER 8 VOUT = Figure 16. AD637 Vector Sum Configuration REV. E –9– VX2 + VV2 AD637 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). TO-116 Package (D-14) 0.005 (0.13) MIN 0.098 (2.49) MAX 14 8 0.098 (2.49) MAX 14 8 0.310 (7.87) 0.220 (5.59) 1 0.320 (8.13) 0.290 (7.37) 0.060 (1.52) 0.015 (0.38) 7 PIN 1 0.785 (19.94) MAX 0.200 (5.08) MAX 0.150 (3.81) MAX 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.014 (0.36) 0.015 (0.38) 0.008 (0.20) 0.100 0.070 (1.78) SEATING (2.54) 0.030 (0.76) PLANE BSC 0.060 (1.52) 0.015 (0.38) 0.150 (3.81) MIN 0.100 0.070 (1.78) SEATING (2.54) 0.030 (0.76) PLANE BSC 0.320 (8.13) 0.290 (7.37) 15° 0° 0.015 (0.38) 0.008 (0.20) SOIC Package (R-16) 0.4133 (10.50) 0.3977 (10.00) 16 9 0.2992 (7.60) 0.2914 (7.40) 1 PIN 1 0.4193 (10.65) 0.3937 (10.00) 8 0.050 (1.27) BSC 0.0118 (0.30) 0.0040 (0.10) 0.1043 (2.65) 0.0926 (2.35) 88 0.0192 (0.49) SEATING 08 0.0125 (0.32) 0.0138 (0.35) PLANE 0.0091 (0.23) 0.0291 (0.74) 3 458 0.0098 (0.25) 0.0500 (1.27) 0.0157 (0.40) PRINTED IN U.S.A. 0.200 (5.08) MAX 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.014 (0.36) 0.310 (7.87) 0.220 (5.59) 1 7 PIN 1 0.785 (19.94) MAX C804f–0–12/99 (rev. E) 0.005 (0.13) MIN Cerdip Package (Q-14) –10– REV. E