LT1373 250kHz Low Supply Current High Efficiency 1.5A Switching Regulator U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LT ®1373 is a low supply current high frequency current mode switching regulator. It can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and “Cuk.” A 1.5A high efficiency switch is included on the die, along with all oscillator, control and protection circuitry. All functions of the LT1373 are integrated into 8-pin SO/PDIP packages. 1mA IQ at 250kHz Uses Small Inductors: 15µH All Surface Mount Components Only 0.6 Square Inch of Board Space Low Minimum Supply Voltage: 2.7V Constant Frequency Current Mode Current Limited Power Switch: 1.5A Regulates Positive or Negative Outputs Shutdown Supply Current: 12µA Typ Easy External Synchronization 8-Pin SO or PDIP Packages Compared to the 500kHz LT1372, which draws 4mA of quiescent current, the LT1373 switches at 250kHz, typically consumes only 1mA and has higher efficiency. High frequency switching allows for small inductors to be used. All surface mount components consume less than 0.6 square inch of board space. U APPLICATIO S ■ ■ ■ ■ ■ New design techniques increase flexibility and maintain ease of use. Switching is easily synchronized to an external logic level source. A logic low on the shutdown pin reduces supply current to 12µA. Unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recovery. Oscillator frequency shifting protects external components during overload conditions. Boost Regulators CCFL Backlight Driver Laptop Computer Supplies Multiple Output Flyback Supplies Inverting Supplies , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATIO 12V Output Efficiency 5V-to-12V Boost Converter 5 OFF + VIN ON 4 S/S 8 VSW 6, 7 R1 215k 1% C4** 22µF 2 FB GND VOUT† 12V + LT1373 C1** 22µF VIN = 5V f = 250kHz D1 MBRS120T3 L1* 22µH † VC 1 C2 0.01µF R2 24.9k 1% MAX IOUT L1 IOUT 15µH 0.3A 22µH 0.35A *SUMIDA CD75-220KC (22µH) OR COILCRAFT D03316-153 (15µH) **AVX TPSD226M025R0200 R3 5k LT1373 • TA01 90 EFFICIENCY (%) 5V 100 80 70 60 50 1 10 100 OUTPUT CURRENT (mA) 1000 LT1373 • TA02 1 LT1373 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Supply Voltage ....................................................... 30V Switch Voltage LT1373 ............................................................... 35V LT1373HV .......................................................... 42V S/S Pin Voltage ....................................................... 30V Feedback Pin Voltage (Transient, 10ms) .............. ±10V Feedback Pin Current ........................................... 10mA Negative Feedback Pin Voltage (Transient, 10ms) ............................................. ±10V Operating Junction Temperature Range Commercial ........................................ 0°C to 125°C* Industrial ......................................... – 40°C to 125°C Short Circuit ......................................... 0°C to 150°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER TOP VIEW VC 1 8 VSW FB 2 7 GND NFB 3 6 GND S S/S 4 5 VIN N8 PACKAGE 8-LEAD PDIP S8 PACKAGE 8-LEAD PLASTIC SO LT1373CN8 LT1373HVCN8 LT1373CS8 LT1373HVCS8 LT1373IN8 LT1373HVIN8 LT1373IS8 LT1373HVIS8 S8 PART MARKING TJMAX = 125°C, θJA = 100°C/ W (N8) TJMAX = 125°C, θJA = 120°C/ W (S8) 1373 1373I 1373H 1373HI Consult factory for Military grade parts. *Units shipped prior to Date Code 9552 are rated at 100°C maximum operating temperature. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted. SYMBOL PARAMETER CONDITIONS VREF Reference Voltage Measured at Feedback Pin VC = 0.8V IFB Feedback Input Current ● MIN TYP MAX UNITS 1.230 1.225 1.245 1.245 1.260 1.265 V V 50 150 275 nA nA %/V VFB = VREF ● VNFB INFB gm AV f Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V Negative Feedback Reference Voltage Measured at Negative Feedback Pin Feedback Pin Open, VC = 0.8V Negative Feedback Input Current 0.01 0.03 ● – 2.51 – 2.55 – 2.45 – 2.45 – 2.39 – 2.35 V V VNFB = VNFR ● – 12 –7 –2 µA Negative Feedback Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V ● 0.01 0.05 %/V Error Amplifier Transconductance ∆IC = ±5µA 250 150 375 ● 500 600 µmho µmho 25 50 90 µA 850 1500 µA 1.95 0.40 2.30 0.52 V V Error Amplifier Source Current VFB = VREF – 150mV, VC = 1.5V ● Error Amplifier Sink Current VFB = VREF + 150mV, VC = 1.5V ● Error Amplifier Clamp Voltage High Clamp, VFB = 1V Low Clamp, VFB = 1.5V 1.70 0.25 Error Amplifier Voltage Gain 250 V/ V VC Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V Switching Frequency 2.7V ≤ VIN ≤ 25V 0°C ≤ TJ ≤ 125°C – 40°C ≤ TJ ≤ 0°C (I Grade) ● 225 210 200 250 250 275 290 290 kHz kHz kHz ● 90 500 ns Maximum Switch Duty Cycle Switch Current Limit Blanking Time 2 ● 95 340 % LT1373 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, unless otherwise noted. SYMBOL PARAMETER CONDITIONS BV Output Switch Breakdown Voltage LT1373 LT1373HV 0°C ≤ TJ ≤ 125°C – 40°C ≤ TJ ≤ 0°C (I Grade) MIN TYP ● 35 47 V ● 42 40 47 V V UNITS 0.5 0.85 Ω 1.9 1.7 2.7 2.5 A A Supply Current Increase During Switch On-Time 10 20 mA/A Control Voltage to Switch Current Transconductance 2 VSAT Output Switch “On” Resistance ISW = 1A ● ILIM Switch Current Limit Duty Cycle = 50% Duty Cycle = 80% (Note 2) ● ● ∆IIN ∆ISW Minimum Input Voltage IQ MAX 1.5 1.3 A/V ● 2.4 2.7 V Supply Current 2.7V ≤ VIN ≤ 25V ● 1 1.5 mA Shutdown Supply Current 2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V 0°C ≤ TJ ≤ 125°C – 40°C ≤ TJ ≤ 0°C (I Grade) ● 12 30 50 µA µA 2.7V ≤ VIN ≤ 25V ● 0.6 1.3 2 V ● 5 12 100 µs ● – 10 15 µA ● 300 340 kHz Shutdown Threshold Shutdown Delay 0V ≤ VS/S ≤ 5V S/S Pin Input Current Synchronization Frequency Range Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired. Note 2: For duty cycles (DC) between 50% and 90%, minimum guaranteed switch current is given by ILIM = 0.667 (2.75 – DC). U W TYPICAL PERFOR A CE CHARACTERISTICS Switch Saturation Voltage vs Switch Current Switch Current Limit vs Duty Cycle 3.0 150°C 100°C 0.9 25°C SWITCH CURRENT LIMIT (A) 0.8 0.7 0.6 0.5 –55°C 0.4 0.3 0.2 3.0 2.5 2.8 25°C AND 125°C 2.0 –55°C 1.5 1.0 INPUT VOLTAGE (V) 1.0 SWITCH SATURATION VOLTAGE (V) Minimum Input Voltage vs Temperature 2.6 2.4 2.2 2.0 0.5 0.1 0 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 SWITCH CURRENT (A) LT1373 • G01 0 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) LT1373 • G02 1.8 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1373 • G03 3 LT1373 U W TYPICAL PERFOR A CE CHARACTERISTICS Shutdown Delay and Threshold vs Temperature 1.8 14 1.4 1.2 12 1.0 10 SHUTDOWN DELAY 8 0.8 6 0.6 4 0.4 2 0.2 0 –50 –25 0 SHUTDOWN THRESHOLD (V) 1.6 SHUTDOWN THRESHOLD 0 25 50 75 100 125 150 TEMPERATURE (°C) 3.0 100 fSYNC = 330kHz 2.5 2.0 1.5 1.0 0.5 0 –50 –25 0 1 0 –1 –2 –3 –4 –1 0 1 2 3 4 5 6 S/S PIN VOLTAGE (V) 7 8 gm = 80 70 60 50 40 30 0 VC THRESHOLD 0.8 0.4 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1373 • G10 4 0 25 50 75 100 125 150 TEMPERATURE (°C) Negative Feedback Input Current vs Temperature 0 350 300 250 200 150 100 50 0.6 100 LT1373 • G09 NEGATIVE FEEDBACK INPUT CURRENT (µA) 1.4 200 Feedback Input Current vs Temperature FEEDBACK INPUT CURRENT (nA) VC PIN VOLTAGE (V) 1.6 300 0 –50 –25 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 FEEDBACK PIN VOLTAGE (V) VFB = VREF VC HIGH CLAMP ∆I (VC) ∆V (FB) 400 LT1373 • G08 2.2 0.1 20 400 1.8 VREF –0.2 –0.1 FEEDBACK PIN VOLTAGE (V) Error Amplifier Transconductance vs Temperature 90 10 9 2.4 1.0 –50 100 VC Pin Threshold and High Clamp Voltage vs Temperature 1.2 –25 500 LT1373 • G07 2.0 0 LT1373 • G06 TRANSCONDUCTANCE (µmho) SWITCHING FREQUENCY (% OF TYPICAL) S/S PIN INPUT CURRENT (µA) 2 125°C 25 –0.3 110 3 –5 50 Switching Frequency vs Feedback Pin Voltage VIN = 5V 25°C –55°C LT1373 • G05 S/S Pin Input Current vs Voltage 4 75 –75 25 50 75 100 125 150 TEMPERATURE (°C) LT1373 • G04 5 Error Amplifier Output Current vs Feedback Pin Voltage ERROR AMPLIFIER OUTPUT CURRENT (µA) 2.0 18 MINIMUM SYNCHRONIZATION VOLTAGE (VP-P) 20 16 SHUTDOWN DELAY (µs) Minimum Synchronization Voltage vs Temperature 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1373 • G11 –2 VNFB = VNFR –4 –6 –8 –10 –12 –14 –16 –18 –20 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1373 • G12 LT1373 U U U PI FU CTIO S floating. To synchronize switching, drive the S/S pin between 300kHz and 340kHz. VC (Pin 1): Compensation Pin. The VC pin is used for frequency compensation, current limiting and soft start. It is the output of the error amplifier and the input of the current comparator. Loop frequency compensation can be performed with an RC network connected from the VC pin to ground. VIN (Pin 5): Input Supply Pin. Bypass VIN with 10µF or more. The part goes into undervoltage lockout when VIN drops below 2.5V. Undervoltage lockout stops switching and pulls the VC pin low. FB (Pin 2): The feedback pin is used for positive output voltage sensing and oscillator frequency shifting. It is the inverting input to the error amplifier. The noninverting input of this amplifier is internally tied to a 1.245V reference. Load on the FB pin should not exceed 100µA when the NFB pin is used. See Applications Information. GND S (Pin 6): The ground sense pin is a “clean” ground. The internal reference, error amplifier and negative feedback amplifier are referred to the ground sense pin. Connect it to ground. Keep the ground path connection to the output resistor divider and the VC compensation network free of large ground currents. NFB (Pin 3): The negative feedback pin is used for negative output voltage sensing. It is connected to the inverting input of the negative feedback amplifier through a 400k source resistor. GND (Pin 7): The ground pin is the emitter connection of the power switch and has large currents flowing through it. It should be connected directly to a good quality ground plane. S/S (Pin 4): Shutdown and Synchronization Pin. The S/S pin is logic level compatible. Shutdown is active low and the shutdown threshold is typically 1.3V. For normal operation, pull the S/S pin high, tie it to VIN or leave it VSW (Pin 8): The switch pin is the collector of the power switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. W BLOCK DIAGRA VIN SHUTDOWN DELAY AND RESET S/S SYNC SW LOW DROPOUT 2.3V REG 250kHz OSC ANTI-SAT LOGIC DRIVER SWITCH 5:1 FREQUENCY SHIFT + 400k NFB NEGATIVE FEEDBACK AMP – COMP 200k – FB + ERROR + AMP 1.245V REF GND SENSE CURRENT AMP VC AV ≈ 6 0.08Ω – GND LT1373 • BD 5 LT1373 U OPERATIO The LT1373 is a current mode switcher. This means that switch duty cycle is directly controlled by switch current rather than by output voltage. Referring to the Block Diagram, the switch is turned “On” at the start of each oscillator cycle. It is turned “Off” when switch current reaches a predetermined level. Control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. This technique has several advantages. First, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. Second, it reduces the 90° phase shift at mid-frequencies in the energy storage inductor. This greatly simplifies closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low dropout design allows input voltage to vary from 2.7V to 25V with virtually no change in device performance. A 250kHz oscillator is the basic clock for all internal timing. It turns “On” the output switch via the logic and driver circuitry. Special adaptive anti-sat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. This minimizes driver dissipation and provides very rapid turn-off of the switch. A 1.245V bandgap reference biases the positive input of the error amplifier. The negative input of the amplifier is brought out for positive output voltage sensing. The error amplifier has nonlinear transconductance to reduce out- put overshoot on start-up or overload recovery. When the feedback voltage exceeds the reference by 40mV, error amplifier transconductance increases ten times, which reduces output overshoot. The feedback input also invokes oscillator frequency shifting, which helps protect components during overload conditions. When the feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle. Unique error amplifier circuitry allows the LT1373 to directly regulate negative output voltages. The negative feedback amplifier’s 400k source resistor is brought out for negative output voltage sensing. The NFB pin regulates at – 2.45V while the amplifier output internally drives the FB pin to 1.245V. This architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. (Consult Linear Technology Marketing for units that can regulate down to – 1.25V.) The error signal developed at the amplifier output is brought out externally. This pin (VC) has three different functions. It is used for frequency compensation, current limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low output current) and 1.9V (high output current). The error amplifier is a current output (gm) type, so this voltage can be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft start. Switch duty cycle goes to zero if the VC pin is pulled below the control pin threshold, placing the LT1373 in an idle mode. U W U U APPLICATIO S I FOR ATIO Positive Output Voltage Setting The LT1373 develops a 1.245V reference (VREF) from the FB pin to ground. Output voltage is set by connecting the FB pin to an output resistor divider (Figure 1). The FB pin bias current represents a small error and can usually be ignored for values of R2 up to 25k. The suggested value for R2 is 24.9k. The NFB pin is normally left open for positive output applications. 6 VOUT R1 FB PIN R1 = R2 R2 ( ) ( ) VOUT = VREF 1 + R1 R2 VOUT –1 1.245 VREF LT1373 • F01 Figure 1. Positive Output Resistor Divider LT1373 U W U U APPLICATIO S I FOR ATIO Negative Output Voltage Setting The LT1373 develops a – 2.45V reference (VNFR) from the NFB pin to ground. Output voltage is set by connecting the NFB pin to an output resistor divider (Figure 2). The – 7µA NFB pin bias current (INFB) can cause output voltage errors and should not be ignored. This has been accounted for in the formula in Figure 2. The suggested value for R2 is 2.49k. The FB pin is normally left open for negative output applications. See Dual Polarity Output Voltage Sensing for limitations of FB pin loading when using the NFB pin. –VOUT INFB R1 NFB PIN ( ) –VOUT = VNFB 1 + R1 + INFB (R1) R2 VOUT – 2.45 R2 VNFR ( ) R1 = 2.45 + (7 • 10 –6) R2 A logic low on the S/S pin activates shutdown, reducing the part’s supply current to 12µA. Typical synchronization range is from 1.05 and 1.8 times the part’s natural switching frequency, but is only guaranteed between 300kHz and 340kHz. A 12µs resetable shutdown delay network guarantees the part will not go into shutdown while receiving a synchronization signal. Caution should be used when synchronizing above 330kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. Higher inductor values will tend to eliminate problems. Thermal Considerations LT1373 • F02 Figure 2. Negative Output Resistor Divider Dual Polarity Output Voltage Sensing Certain applications benefit from sensing both positive and negative output voltages. One example is the Dual Output Flyback Converter with Overvoltage Protection circuit shown in the Typical Applications section. Each output voltage resistor divider is individually set as described above. When both the FB and NFB pins are used, the LT1373 acts to prevent either output from going beyond its set output voltage. For example in this application, if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. The positive output would sag slightly below its set-point voltage. This technique prevents either output from going unregulated high at no load. Please note that the load on the FB pin should not exceed 100µA when the NFB pin is used. This situation occurs when the resistor dividers are used at both FB and NFB. True load on FB is not the full divider current unless the positive output is shorted to ground. See Dual Output Flyback Converter application. Shutdown and Synchronization The dual function S/S pin provides easy shutdown and synchronization. It is logic level compatible and can be pulled high, tied to VIN or left floating for normal operation. Care should be taken to ensure that the worst-case input voltage and load current conditions do not cause excessive die temperatures. The packages are rated at 120°C/W for SO (S8) and 130°C/W for PDIP (N8). Average supply current (including driver current) is: IIN = 1mA + DC (ISW/60 + ISW • 0.004) ISW = switch current DC = switch duty cycle Switch power dissipation is given by: PSW = (ISW)2 • RSW • DC RSW = output switch “On” resistance Total power dissipation of the die is the sum of supply current times supply voltage plus switch power: PD(TOTAL) = (IIN • VIN) + PSW Choosing the Inductor For most applications the inductor will fall in the range of 10µH to 50µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the power switch which has a 1.5A limit. Higher values also reduce input ripple voltage, and reduce core loss. When choosing an inductor you might have to consider maximum load current, core and copper losses, allowable 7 LT1373 U W U U APPLICATIO S I FOR ATIO component height, output voltage ripple, EMI, fault current in the inductor, saturation, and of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts). Keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 1. Assume that the average inductor current (for a boost converter) is equal to load current times VOUT/VIN and decide whether or not the inductor must withstand continuous overload conditions. If average inductor current at maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 1.5A overload condition. Also, be aware that boost converters are not short-circuit protected, and that under output short conditions, inductor current is limited only by the available current of the input supply. 5. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the Linear Technology application department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall in between somewhere. The following formula assumes continuous mode operation, but it errors only slightly on the high side for discontinuous mode, so it can be used for all conditions. IPEAK = IOUT • VOUT VIN (VOUT – VIN) + VIN 2(f)(L)(VOUT) VIN = minimum input voltage f = 250kHz switching frequency 3. Decide if the design can tolerate an “open” core geometry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent EMI problems. One would not want an open core next to a magnetic storage media for instance! This is a tough decision because the rods or barrels are temptingly cheap and small, and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. 4. Start shopping for an inductor which meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating), and fault current, (if the 8 Output Capacitor The output capacitor is normally chosen by its effective series resistance (ESR), because this is what determines output ripple voltage. At 500kHz, any polarized capacitor is essentially resistive. To get low ESR takes volume, so physically smaller capacitors have high ESR. The ESR range for typical LT1373 applications is 0.05Ω to 0.5Ω. A typical output capacitor is an AVX type TPS, 22µF at 25V, with a guaranteed ESR less than 0.2Ω. This is a “D” size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and tested for low ESR, so they give the lowest ESR for a given volume. To further reduce ESR, multiple output capacitors can be used in parallel. The value in microfarads is not particularly critical and values from 22µF to greater than 500µF work well, but you cannot cheat mother nature on ESR. If you find a tiny 22µF solid tantalum capacitor, it will have high ESR and output ripple voltage will be terrible. Table 1 shows some typical solid tantalum surface mount capacitors. Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E CASE SIZE AVX TPS, Sprague 593D AVX TAJ ESR (MAX Ω) RIPPLE CURRENT (A) 0.1 to 0.3 0.7 to 0.9 0.7 to 1.1 0.4 0.1 to 0.3 0.9 to 2.0 0.7 to 1.1 0.36 to 0.24 0.2 (Typ) 1.8 to 3.0 0.5 (Typ) 0.22 to 0.17 2.5 to 10 0.16 to 0.08 D CASE SIZE AVX TPS, Sprague 593D AVX TAJ C CASE SIZE AVX TPS AVX TAJ B CASE SIZE AVX TAJ LT1373 U W U U APPLICATIO S I FOR ATIO Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. This is historically true and type TPS capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. Solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. High discharge surges, such as when the regulator output is dead shorted, do not harm the capacitors. Single inductor boost regulators have large RMS ripple current in the output capacitor, which must be rated to handle the current. The formula to calculate this is: Output Capacitor Ripple Current (RMS) DC IRIPPLE (RMS) = IOUT 1 – DC = IOUT VOUT – VIN VIN Input Capacitors The input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular, and does not contain large squarewave currents as is found in the output capacitor. Capacitors in the range of 10µF to 100µF with an ESR (effective series resistance) of 0.3Ω or less work well up to a full 1.5A switch current. Higher ESR capacitors may be acceptable at low switch currents. Input capacitor ripple current for boost converter is: IRIPPLE = 0.3(VIN)(VOUT – VIN) (f)(L)(VOUT) f = 250kHz switching frequency The input capacitor can see a very high surge current when a battery or high capacitance source is connected “live”, and solid tantalum capacitors can fail under this condition. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (AVX TPS series, for instance), but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor. AVX recommends derating capacitor voltage by 2:1 for high surge applications. Ceramic and aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. Ceramic Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems. Solid tantalum capacitor ESR generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. They are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. Linear Technology plans to issue a Design Note on the use of ceramic capacitors in the near future. Output Diode The suggested output diode (D1) is a 1N5818 Schottky or its Motorola equivalent, MBR130. It is rated at 1A average forward current and 30V reverse voltage. Typical forward voltage is 0.42V at 1A. The diode conducts current only during switch-off time. Peak reverse voltage for boost converters is equal to regulator output voltage. Average forward current in normal operation is equal to output current. Frequency Compensation Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance (≈ 1MΩ) of the error amplifier. The pole falls in the range of 5Hz to 30Hz. The series resistor creates a “zero” at 2kHz to 10kHz, which improves loop stability and transient response. A second capacitor, typically one tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: VC Pin Ripple = 1.245(VRIPPLE)(gm)(RC) VOUT 9 LT1373 U W U U APPLICATIO S I FOR ATIO VRIPPLE = output ripple (VP-P) gm = error amplifier transconductance (≈ 375µmho) RC = series resistor on VC pin VOUT = DC output voltage To prevent irregular switching, VC pin ripple should be kept below 50mVP-P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 0.001µF capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate. The high speed switching current path is shown schematically in Figure 3. Minimum lead length in this path is essential to ensure clean switching and low EMI. The path including the switch, output diode and output capacitor is the only one containing nanosecond rise and fall times. Keep this path as short as possible. L1 SWITCH NODE VOUT HIGH FREQUENCY CIRCULATING PATH VIN LOAD Switch Node Considerations For maximum efficiency, switch rise and fall time are made as short as possible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the switch node is essential. B field (magnetic) radiation is minimized by keeping output diode, switch pin and output bypass capacitor leads as short as possible. E field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. A ground plane should always be used under the switcher circuitry to prevent interplane coupling. LT1373 • F03 Figure 3 More Help For more detailed information on switching regulator circuits, please see AN19. Linear Technology also offers a computer software program, SwitcherCADTM, to assist in designing switching converters. In addition, our applications department is always ready to lend a helping hand. SwitcherCAD is a trademark of Linear Technology Corporation. U TYPICAL APPLICATIONS N Positive-to-Negative Converter with Direct Feedback VIN 2.7V TO 16V + OFF C1 22µF VIN VSW LT1373 NFB VC GND 1 C2 0.01µF R1 5k R2 275k 1% T1* 5 ON 4 S/S Dual Output Flyback Converter with Overvoltage Protection 6, 7 8 2 D2 P6KE-15A D3 1N4148 1 • 4 • † MAX IOUT IOUT 0.3A 0.5A 0.75A VIN 4.75V TO 13V C3 47µF 3 D1 MBRS130LT3 3 + R1 302.6k 1% R2 2.55k 1% –VOUT† –5V R3 2.49k 1% OFF C1 100µF 2 5 VIN 8 VSW FB ON 4 S/S LT1373 VIN 3V 5V 9V *COILTRONICS CTX20-2 (407) 241-7876 + NFB VC GND 1 LT1373 • TA03 MBRS140T3 T1* 2, 3 5 + P6KE-20A • 3 1N4148 6, 7 •4 8 • 1 MBRS140T3 6, 7 C2 0.01µF R3 5k *DALE LPE-4841-100MB (605) 665-9301 10 + VOUT 15V C3 47µF C4 47µF –VOUT –15V R4 12.4k 1% R5 2.49k 1% LT1373 • TA04 LT1373 U TYPICAL APPLICATIO S Low Ripple 5V to – 3V “Cuk” † Converter 2 3 1• •4 R1 1k 1% C2 47µF 16V 5 C1 22µF 10V VOUT –3V 250mA L1* VIN 5V + 4 7 6 VSW VIN 8 + C6 0.1µF S/S LT1373 GND NFB GND S VC 3 1 D1** + R4 5k C4 0.01µF C3 47µF 16V R2 5.49k 1% *SUMIDA CLS62-100L **MOTOROLA MBR0520LT3 † PATENTS MAY APPLY U PACKAGE DESCRIPTION LT1373 • TA05 Dimensions in inches (millimeters) unless otherwise noted. N8 Package 8-Lead PDIP (Narrow 0.300) (LTC DWG # 05-08-1510) 0.300 – 0.325 (7.620 – 8.255) 0.009 – 0.015 (0.229 – 0.381) ( 0.045 – 0.065 (1.143 – 1.651) 0.400* (10.160) MAX 0.130 ± 0.005 (3.302 ± 0.127) 0.065 (1.651) TYP 8 7 6 5 1 2 3 4 0.255 ± 0.015* (6.477 ± 0.381) +0.035 0.325 –0.015 +0.889 8.255 –0.381 ) 0.125 (3.175) 0.020 MIN (0.508) MIN 0.018 ± 0.003 (0.457 ± 0.076) 0.100 (2.54) BSC N8 1098 *THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS. MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm) S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.053 – 0.069 (1.346 – 1.752) 0.008 – 0.010 (0.203 – 0.254) 0°– 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 8 7 6 5 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) SO8 1298 1 2 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of circuits as described herein will not infringe on existing patent rights. 3 4 11 LT1373 U TYPICAL APPLICATIO S 90% Efficient CCFL Supply Two Li-Ion Cells to 5V SEPIC Conveter 5mA MAX LAMP C2 27pF VIN 4V TO 9V D1 1N4148 10 T1 VIN 4.5V TO 30V 5 4 3 2 + 10µF C1 0.1µF OFF + 330Ω Q1 Q2 ON 4 S/S LT1373 FB GND + OFF ON 4 S/S VSW 562Ω* 8 VFB GND 6, 7 + 22k VOUT† 5V R2 75k 1% + C3 100µF 10V R3 24.9k 1% † MAX IOUT 10k C1 = AVX TPSD 336M020R0200 C2 = TOKIN 1E225ZY5U-C203-F C3 = AVX TPSD 107M010R0100 L1 = COILTRONICS CTX33-2, SINGLE INDUCTOR WITH TWO WINDINGS 2 VC • L1B 33µH 2 R1 5k C4 0.01µF D2 1N4148 20k DIMMING LT1373 • 1 5 VIN 8 L1A C2 D1 33µH 2.2µF MBRS130LT3 VC 6, 7 L1 100µH 2.2µF VSW C1 33µF 20V 1N5818 2.7V TO 5.5V 5 VIN 1 IOUT 0.45A 0.55A 0.65A 0.72A VIN 4V 5V 7V 9V LT1373 • TA07 0.1µF 1 1N4148 2µF OPTIONAL REMOTE DIMMING C1 = WIMA MKP-20 L1 = COILCRAFT D03316-104 Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001 T1 = COILTRONICS CTX 110609 * = 1% FILM RESISTOR LT1372 • TA06 CCFL BACKLIGHT APPLICATION CIRCUITS CONTAINED IN THIS DATA SHEET ARE COVERED BY U.S. PATENT NUMBER 5408162 AND OTHER PATENTS PENDING DO NOT SUBSTITUTE COMPONENTS COILTRONICS (407) 241-7876 COILCRAFT (708) 639-6400 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1172 100kHz 1.25A Boost Switching Regulator Also for Flyback, Buck and Inverting Configurations 13V 1.2A Monolithic Buck Converter Includes PMOS Switch On-Chip ® LTC 1265 LT1302 Micropower 2A Boost Converter Converts 2V to 5V at 600mA LT1308A/LT1308B 600kHz 2A Switch DC/DC Converter 5V at 1A from a Single Li-Ion Cell LT1370 500kHz High Efficiency 6A Boost Converter 6A, 0.065Ω Internal Switch LT1372 500kHz 1.5A Boost Switching Regulator Also Regulates Negative Flyback Outputs LT1374 4.5A, 500kHz Step-Down Converter 4.5A, 0.07Ω Internal Switch LT1376 500kHz 1.5A Buck Switching Regulator Handles Up to 25V Inputs LT1377 1MHz 1.5A Boost Switching Regulator Only 1MHz Integrated Switching Regulator Available LT1613 1.4MHz Switching Regulator in 5-Lead SOT-23 5V at 200mA from 4.4V Input LT1615 Micropower Step-Up DC/DC in 5-Lead SOT-23 20µA IQ, 36V, 350mA Switch LT1949 600kHz, 1A Switch PWM DC/DC Converter 1.1A, 0.5Ω, 30V Internal Switch, VIN as Low as 1.5V 12 Linear Technology Corporation 1373fb LT/TP 0200 2K REV B • PRINTED IN THE USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1995