LTC1872 Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23 U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO High Efficiency: Over 90% High Output Currents Easily Achieved Wide VIN Range: 2.5V to 9.8V VOUT Limited Only by External Components Constant Frequency 550kHz Operation Burst ModeTM Operation at Light Load Current Mode Operation for Excellent Line and Load Transient Response Low Quiescent Current: 270µA Shutdown Mode Draws Only 8µA Supply Current ±2.5% Reference Accuracy Tiny 6-Lead SOT-23 Package U APPLICATIO S ■ ■ ■ ■ The LTC1872 boasts a ±2.5% output voltage accuracy and consumes only 270µA of quiescent current. For applications where efficiency is a prime consideration, the LTC1872 is configured for Burst Mode operation, which enhances efficiency at low output current. In shutdown, the device draws a mere 8µA. The high 550kHz constant operating frequency allows the use of a small external inductor. The LTC1872 is available in a small footprint 6-lead SOT-23. Lithium-Ion-Powered Applications Cellular Telephones Wireless Modems Portable Computers Scanners , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U ■ The LTC®1872 is a constant frequency current mode stepup DC/DC controller providing excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the LTC1872 when the input voltage falls below 2.0V. TYPICAL APPLICATION 220pF 80.6k 1 VIN ITH/RUN 5 100 3 GND SENSE – VFB NGATE VOUT 5V 1A 4 6 M1 D1 VIN = 3.3V VOUT = 5V 95 L1 4.7µH LTC1872 2 Efficiency vs Load Current + C2 2× 22µF 6.3V 422k 90 EFFICIENCY (%) 147k VIN 3.3V C1 10µF 10V R1 0.03Ω 85 80 75 C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: MURATA GRM42-2X5R226K6.3 D1: IR10BQ015 L1: MURATA LQN6C4R7M04 M1: IRLMS2002 R1: DALE 0.25W 70 65 1872 TA01 Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter 1 10 100 LOAD CURRENT (mA) 1000 1872 TA01b 1 LTC1872 W U U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) Input Supply Voltage (VIN).........................– 0.3V to 10V SENSE –, NGATE Voltages ............ – 0.3V to (VIN + 0.3V) VFB, ITH/RUN Voltages ..............................– 0.3V to 2.4V NGATE Peak Output Current (< 10µs) ....................... 1A Storage Ambient Temperature Range ... – 65°C to 150°C Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Note 3) ............................. 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW LTC1872ES6 6 NGATE ITH/RUN 1 5 VIN GND 2 4 SENSE – VFB 3 S6 PART MARKING S6 PACKAGE 6-LEAD PLASTIC SOT-23 LTMK TJMAX = 150°C, θJA = 230°C/ W Consult factory for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2) PARAMETER CONDITIONS Input DC Supply Current Normal Operation Sleep Mode Shutdown UVLO Typicals at VIN = 4.2V (Note 4) 2.4V ≤ VIN ≤ 9.8V 2.4V ≤ VIN ≤ 9.8V 2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V VIN < UVLO Threshold Undervoltage Lockout Threshold VIN Falling VIN Rising Shutdown Threshold (at ITH/RUN) Start-Up Current Source VITH/RUN = 0V Regulated Feedback Voltage 0°C to 70°C(Note 5) – 40°C to 85°C(Note 5) VFB Input Current (Note 5) MIN TYP MAX UNITS 270 230 8 6 420 370 22 10 µA µA µA µA ● 1.55 1.85 2.00 2.10 2.35 2.40 V V ● 0.15 0.35 0.55 V 0.25 0.5 0.85 µA ● ● 0.780 0.770 0.800 0.800 0.820 0.830 V V 10 50 nA 550 650 kHz Oscillator Frequency VFB = 0.8V Gate Drive Rise Time CLOAD = 3000pF 40 ns Gate Drive Fall Time CLOAD = 3000pF 40 ns Peak Current Sense Voltage (Note 6) 120 mV Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC1872E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • θJA°C/W) 2 500 114 Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC1872 is tested in a feedback loop that servos VFB to the output of the error amplifier. Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense voltage is VREF/6.67 at duty cycle <40%, and decreases as duty cycle increases due to slope compensation as shown in Figure 2. LTC1872 U W TYPICAL PERFORMANCE CHARACTERISTICS Reference Voltage vs Temperature 10 VIN = 4.2V VFB VOLTAGE (mV) 815 810 805 800 795 790 785 2.24 VIN = 4.2V 8 2.20 6 4 2 0 –2 –4 –6 780 5 25 45 65 85 105 125 TEMPERATURE (°C) 5 25 45 65 85 105 125 TEMPERATURE (°C) 2.04 2.00 1.96 1.84 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1872 G02 Maximum Current Sense Trip Voltage vs Duty Cycle 1872 G03 Shutdown Threshold vs Temperature 600 VIN = 4.2V TA = 25°C 120 2.08 1.88 –10 –55 –35 –15 1872 G01 130 2.12 1.92 –8 775 –55 –35 –15 VIN FALLING 2.16 UVLO TRIP VOLTAGE (V) 820 NORMALIZED FREQUENCY (%) 825 Undervoltage Lockout Trip Voltage vs Temperature Normalized Oscillator Frequency vs Temperature 560 VIN = 4.2V ITH/RUN VOLTAGE (mV) VIN – VSENSE – (mV) 520 110 100 90 80 70 480 440 400 360 320 280 60 240 50 20 30 40 50 60 70 80 DUTY CYCLE (%) 90 100 1872 G04 200 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1872 G05 U U U PIN FUNCTIONS ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V causes the device to be shut down. In shutdown all functions are disabled and the NGATE pin is held low. GND (Pin 2): Ground Pin. SENSE – (Pin 4): The Negative Input to the Current Comparator. VIN (Pin 5): Supply Pin. Must be closely decoupled to GND Pin 2. NGATE (Pin 6): Gate Drive for the External N-Channel MOSFET. This pin swings from 0V to VIN. VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output. 3 LTC1872 W FUNCTIONAL DIAGRA U VIN SENSE – 5 4 U + ICMP – VIN RS SLOPE COMP OSC NGATE SWITCHING LOGIC AND BLANKING CIRCUIT R Q S 6 – FREQ FOLDBACK BURST CMP + 0.3V SLEEP – + 0.15V OVP + – VREF + 60mV + VREF 0.8V VIN EAMP 0.5µA VFB + – 1 ITH/RUN 3 VIN VIN 0.3V – 0.35V VOLTAGE REFERENCE + SHDN CMP VREF 0.8V – GND SHDN UV 2 UNDERVOLTAGE LOCKOUT 1.2V 1872FD U OPERATIO (Refer to Functional Diagram) Main Control Loop The LTC1872 is a constant frequency current mode switching regulator. During normal operation, the external N-channel power MOSFET is turned on each cycle by the oscillator and turned off when the current comparator (ICMP) resets the RS latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the ITH/RUN pin, which is the output of the error amplifier EAMP. An external resistive divider connected between VOUT and ground allows the EAMP to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 4 0.8V reference, which in turn causes the ITH/RUN voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling the ITH/RUN pin low. Releasing ITH/RUN allows an internal 0.5µA current source to charge up the external compensation network. When the ITH/RUN pin reaches 0.35V, the main control loop is enabled with the ITH/RUN voltage then pulled up to its zero current level of approximately 0.7V. As the external compensation network continues to charge up, the corresponding output current trip level follows, allowing normal operation. LTC1872 U OPERATIO (Refer to Functional Diagram) Comparator OVP guards against transient overshoots > 7.5% by turning off the external N-channel power MOSFET and keeping it off until the fault is removed. Burst Mode Operation The LTC1872 enters Burst Mode operation at low load currents. In this mode, the peak current of the inductor is set as if VITH/RUN = 1V (at low duty cycles) even though the voltage at the ITH/RUN pin is at a lower value. If the inductor’s average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the ITH/RUN voltage goes below 0.85V, the sleep signal goes high, turning off the external MOSFET. The sleep signal goes low when the ITH/RUN voltage goes above 0.925V and the LTC1872 resumes normal operation. The next oscillator cycle will turn the external MOSFET on and the switching cycle repeats. Undervoltage Lockout To prevent operation of the N-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorporated into the LTC1872. When the input supply voltage drops below approximately 2.0V, the N-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes. Overvoltage Protection The overvoltage comparator in the LTC1872 will turn the external MOSFET off when the feedback voltage has risen 7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV. Slope Compensation and Inductor’s Peak Current The inductor’s peak current is determined by: IPK = VITH − 0.7 ( 10 RSENSE ) when the LTC1872 is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves in Figure 2. Short-Circuit Protection Since the power switch in a boost converter is not in series with the power path from input to load, turning off the switch provides no protection from a short-circuit at the output. External means such as a fuse in series with the boost inductor must be employed to handle this fault condition. 110 100 SF = IOUT/IOUT(MAX) (%) 90 80 70 60 50 IRIPPLE = 0.4IPK AT 5% DUTY CYCLE IRIPPLE = 0.2IPK AT 5% DUTY CYCLE 40 30 20 VIN = 4.2V 10 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 1872 F02 Figure 2. Maximum Output Current vs Duty Cycle 5 LTC1872 U W U U APPLICATIONS INFORMATION The basic LTC1872 application circuit is shown in Figure␣ 1. External component selection is driven by the load requirement and begins with the selection of L1 and RSENSE (= R1). Next, the power MOSFET and the output diode D1 is selected followed by CIN(= C1) and COUT(= C2). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the inductor’s peak current. The output current the LTC1872 can provide is given by: 0.12 I VIN IOUT = − RIPPLE 2 VOUT + VD RSENSE where IRIPPLE is the inductor peak-to-peak ripple current (see Inductor Value Calculation section) and VD is the forward drop of the output diode at the full rated output current. A reasonable starting point for setting ripple current is: ( )( ) VOUTVIN+ VD IRIPPLE = O.4 IOUT Rearranging the above equation, it becomes: RSENSE = 1 VIN 10 IOUT VOUT + VD ( )( ) for Duty Cycle < 40% However, for operation that is above 40% duty cycle, slope compensation’s effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using the scaling factor (SF, in %) in Figure 2, the value of RSENSE is: RSENSE = 6 SF VIN 100 VOUT + VD (10)(IOUT )( ) Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current. However, this is at the expense of efficiency due to an increase in MOSFET gate charge losses. The inductance value also has a direct effect on ripple current. The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VOUT. The inductor’s peak-to-peak ripple current is given by: IRIPPLE = VIN VOUT + VD − VIN f L VOUT + VD () where f is the operating frequency. Accepting larger values of IRIPPLE allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is: V + VD IRIPPLE = 0.4 IOUT (MAX) OUT V IN In Burst Mode operation, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peak-to-peak ripple current must not exceed: IRIPPLE ≤ 0.03 RSENSE This implies a minimum inductance of: LMIN = VOUT + VD − VIN 0.03 VOUT + VD f RSENSE VIN A smaller value than L MIN could be used in the circuit; however, the inductor current will not be continuous during burst periods. LTC1872 U W U U APPLICATIONS INFORMATION Inductor Selection When selecting the inductor, keep in mind that inductor saturation current has to be greater than the current limit set by the current sense resistor. Also, keep in mind that the DC resistance of the inductor will affect the efficiency. Off the shelf inductors are available from Murata, Coilcraft, Toko, Panasonic, Coiltronics and many other suppliers. It is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings. Schottky diodes are recommended for low forward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation. Power MOSFET Selection CIN and COUT Selection The main selection criteria for the power MOSFET are the threshold voltage VGS(TH), the “on” resistance RDS(ON), reverse transfer capacitance CRSS and total gate charge. To prevent large input voltage ripple, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current for a boost converter is approximately equal to: Since the LTC1872 is designed for operation down to low input voltages, a logic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1872 is less than the absolute maximum VGS rating, typically 8V. The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation given by: R DS(ON) ≅ PP (DC )IIN (1+ δp) 2 where PP is the allowable power dissipation and δp is the temperature dependency of RDS(ON). (1 + δp) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. DC is the maximum operating duty cycle of the LTC1872. Output Diode Selection Under normal load conditions, the average current conducted by the diode in a boost converter is equal to the output load current: ID(avg) = IOUT ( ) C IN Required IRMS ≈ 0.3 IRIPPLE where IRIPPLE is as defined in the Inductor Value Calculation section. Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC1872, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: V + VD IRIPPLE ∆VOUT ≈ IO • OUT + • 2 VIN 1 2 2 1 2 ESR + 2π fC OUT 7 LTC1872 U W U U APPLICATIONS INFORMATION where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. The output capacitor RMS current is approximately equal to: Setting Output Voltage The LTC1872 develops a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by: R2 VOUT = 0.8V 1 + R1 105 where IPK is the peak inductor current and DC is the switch duty cycle. When using electrolytic output capacitors, if the ripple and ESR requirements are met, there is likely to be far more capacitance than required. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. An excellent choice of tantalum capacitors is the AVX TPS and KEMET T510 series of surface mount tantalum capacitors. Also, ceramic capacitors in X5R pr X7R dielectrics offer excellent performance. Low Supply Operation Although the LTC1872 can function down to approximately 2.0V, the maximum allowable output current is reduced when VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 3 is the effect of VIN on VREF as VIN goes below 2.3V. 8 NORMALIZED VOLTAGE (%) IPK • DC − DC 2 VREF 100 VITH 95 90 85 80 75 2.0 2.2 3.0 2.4 2.6 2.8 INPUT VOLTAGE (V) 1872 F03 Figure 3. Line Regulation of VREF and VITH VOUT LTC1872 3 VFB R2 R1 1872 F04 Figure 4. Setting Output Voltage LTC1872 U W U U APPLICATIONS INFORMATION For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to LTC1872. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (η1 + η2 + η3 + ...) where η1, η2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1872 circuits: 1) LTC1872 DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode. 1. The VIN current is the DC supply current, given in the electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN. 2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is a current out of VIN which is typically much larger than the contoller’s DC supply current. In continuous mode, IGATECHG = f(Qp). 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current sense resistor. The MOSFET RDS(ON) multiplied by duty cycle times the average output current squared can be summed with I2R losses in the inductor ESR in series with the current sense resistor. 4. The output diode is a major source of power loss at high currents. The diode loss is calculated by multiplying the forward voltage by the load current. 5. Transition losses apply to the external MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from: Transition Loss = 2(VIN)2IIN(MAX)CRSS(f) Other losses, including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. 9 LTC1872 U W U U APPLICATIONS INFORMATION PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1872. These items are illustrated graphically in the layout diagram in Figure 5. Check the following in your layout: 1. The Schottky diode should be closely connected between the output capacitor and the drain of the external MOSFET. 2. The (+) plate of CIN should connect to the sense resistor as closely as possible. This capacitor provides AC current to the inductor. 4. Connect the end of RSENSE as close to VIN (Pin 5) as possible. The VIN pin is the SENSE + of the current comparator. 5. The trace from SENSE – (Pin 4) to the Sense resistor should be kept short. The trace should connect close to RSENSE. 6. Keep the switching node NGATE away from sensitive small signal nodes. 7. The VFB pin should connect directly to the feedback resistors. The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. 3. The input decoupling capacitor (0.1µF) should be connected closely between VIN (Pin 5) and ground (Pin 2). 1 ITH/RUN NGATE VIN 6 LTC1872 2 RITH VIN GND RS 5 0.1µF 3 CITH VFB SENSE – 4 + M1 L1 CIN D1 VOUT + R2 COUT R1 BOLD LINES INDICATE HIGH CURRENT PATHS Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist) 10 1872 F05 LTC1872 U TYPICAL APPLICATIO LTC1872 12V/500mA Boost Converter C1 10µF 10V R1 0.033Ω 1 ITH/RUN VIN 5 L1 10µH LTC1872 10k 2 3 220pF GND SENSE – VFB NGATE 4 6 VIN 3V TO 9.8V M1 D1 + C2 47µF 16V VOUT 12V 1.1M 1872 TA02 78.7k C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSE476M016R0150 D1: IR10BQ015 L1: COILTRONICS UP2B-100 M1: Si9804DV R1: DALE 0.25W LTC1872 Three-Cell White LED Driver VIN = 3 AA CELLS ≈ 2.7V TO 4.8V C1 10µF 10V R1 0.27Ω AA AA 1 ITH/RUN VIN 5 LTC1872 10k 2 220pF 3 GND SENSE – VFB NGATE L1 150µH VOUT ≈ 28.8V (WITH 8 LEDs) 4 6 M1 AA D0 + C2 15µF 35V C3 0.1µF CERAMIC 15mA D1 D2 1 TO 8 WHITE LEDs • • • D8 C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSD156M035R0300 D0: MOTOROLA MBR0540 D1-D7: CMD333UWC L1: COILCRAFT DO1608C-154 M1: Si9804 R1: DALE 0.25W Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 53.6Ω 1872 TA04 11 LTC1872 U TYPICAL APPLICATIO LTC1872 – 2.5V to 3.3V/0.5A Boost Converter R1 0.034Ω 10k 220pF ITH/RUN VIN 5 + 1 L1 4.7µH LTC1872 2.7V to 9.8V Input to 3.3V/1.2A Output SEPIC Converter C2 2× 100µF 10V LTC1872 2 3 GND SENSE – VFB NGATE 4 6 M1 D1 332k VOUT 3.3V 0.5A CC1 220pF RC1 10k 1 3 0.1µF CERAMIC + C1 100µF 10V ITH/RUN VIN 5 L1A LTC1872 2 VIN 2.7V TO 9.8V CIN 10µF 10V, X5R RCS 0.03Ω GND SENSE – VFB NGATE L1B 4 D1 MBRM120 Rf1 252k C01 180µF 4V, SP VOUT 3.3V/1.2A CS 4.7µF 10V 6 Rf2 80.6k U1 + M1 1872 TA05 80.6k VIN –2.5V 180k CIN, CS; TOKO, MURATA OR TAIYO YUDEN C01: PANASONIC EEFUE0G181R L1: BH ELECTRONICS 511-1012 M1: IRLMS2002 RCS: DALE OR IRC 1872 TA03 C1, C2: AVX TPSE107M010R0100 D1: MOTOROLA MBR2045CT L1: COILTRONICS UP2B-4R7 M1: Si9804DV R1: DALE 0.25W U1: PANASONIC 2SB709A U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S6 Package 6-Lead Plastic SOT-23 2.6 – 3.0 (0.110 – 0.118) 1.50 – 1.75 (0.059 – 0.069) 0.35 – 0.55 (0.014 – 0.022) FOR VOUT = 5V CHANGE Rf1 TO 427kΩ AND C01 TO 150µF, 6V PANASONIC SP TYPE CAPACITOR (LTC DWG # 05-08-1634) 0.00 – 0.15 (0.00 – 0.006) 0.09 – 0.20 (0.004 – 0.008) (NOTE 2) 0.90 – 1.45 (0.035 – 0.057) 0.35 – 0.50 0.90 – 1.30 (0.014 – 0.020) (0.035 – 0.051) 1.90 (0.074) SIX PLACES (NOTE 2) REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DIMENSIONS ARE INCLUSIVE OF PLATING 3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 4. MOLD FLASH SHALL NOT EXCEED 0.254mm 5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ) 2.80 – 3.00 (0.110 – 0.118) (NOTE 3) 0.95 (0.037) REF S6 SOT-23 0898 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1304 Micropower DC/DC Converter with Low-Battery Detector 120µA Quiescent Current, 1.5V ≤ VIN ≤ 8V LT1610 1.7MHz, Single Cell Micropower DC/DC Converter 30µA Quiescent Current, VIN Down to 1V LT1613 1.4MHz, Single Cell DC/DC Converter in 5-Lead SOT-23 Internally Compensated, VIN Down to 1V LT1619 Low Voltage Current Mode PWM Controller 8-Lead MSOP Package, 1.9V ≤ VIN ≤ 18V LT1680 High Power DC/DC Step-Up Controller Operation Up to 60V, Fixed Frequency Current Mode LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller 8-Pin N-Channel Drive, 3.5V ≤ VIN ≤ 36V LT1615 Micropower Step-Up DC/DC Converter in SOT-23 20µA Quiescent Current, VIN Down to 1V LTC1700 No RSENSE Synchronous Current Mode DC/DC Step-Up Controller 95% Efficient, 0.9V ≤ VIN ≤ 5V, 550kHz Operation LTC1772 Constant Frequency Current Mode Step-Down DC/DC Controller VIN 2.5V to 9.8V, IOUT up to 4A, SOT-23 Package LTC3401/LTC3402 1A/2A, 3MHz Micropower Synchronous Boost Converter 10-Lead MSOP Package, 0.5V ≤ VIN ≤ 5V 12 Linear Technology Corporation 1872f LT/TP 0301 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 2000