LINER LTC4267CGN-1-TR

LTC4267-1
Power over Ethernet
IEEE 802.3af PD Interface with
Integrated Switching Regulator
DESCRIPTIO
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FEATURES
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Complete Power Interface Port for IEEE 802®.3af
Powered Device (PD)
Onboard 100V, UVLO Switch
Precision Dual Level Inrush Current Limit
Integrated Current Mode Switching Regulator
Onboard 25k Signature Resistor with Disable
Programmable Classiﬁcation Current (Class 0-4)
Thermal Overload Protection
Power Good Signal
Integrated Error Ampliﬁer and Voltage Reference
Low Proﬁle 16-Pin SSOP Package
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APPLICATIO S
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IP Phone Power Management
Wireless Access Points
Security Cameras
Power over Ethernet
The LTC®4267-1 combines an IEEE 802.3af compliant
Powered Device (PD) interface with a current mode switching regulator, providing a complete power solution for PD
applications. The LTC4267-1 integrates the 25k signature
resistor, classiﬁcation current source, thermal overload
protection, signature disable and power good signal along
with an undervoltage lockout optimized for use with the
IEEE-required diode bridge. The LTC4267-1 provides an
increased operational current limit, maximizing power
available for class 3 applications.
The current mode switching regulator is designed for
driving a 6V rated N-channel MOSFET and features programmable slope compensation, soft-start, and constant
frequency operation, minimizing noise even with light
loads. The LTC4267-1 includes an onboard error ampliﬁer
and voltage reference allowing use in both isolated and
nonisolated conﬁgurations.
The LTC4267-1 is available in a space saving, low proﬁle
16-pin SSOP package.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
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TYPICAL APPLICATIO
Class 2 PD with 3.3V Isolated Power Supply
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LTC4267-1
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ABSOLUTE
AXI U RATI GS
PIN CONFIGURATION
(Note 1)
5017*&8
VPORTN with Respect to VPORTP Voltage ... 0.3V to –100V
POUT, SIGDISA, ⎯P⎯W⎯R⎯G⎯D
Voltage..................... VPORTN + 100V to VPORTN –0.3V
PVCC to PGND Voltage (Note 2)
Low Impedance Source ........................... –0.3V to 8V
Current Fed .......................................... 5mA into PVCC
RCLASS Voltage .................VPORTN + 7V to VPORTN – 0.3V
⎯P⎯W⎯R⎯G⎯D Current .....................................................10mA
RCLASS Current.....................................................100mA
NGATE to PGND Voltage ...........................–0.3V to PVCC
VFB, ITH/RUN to PGND Voltages ................ –0.3V to 3.5V
SENSE to PGND Voltage .............................. –0.3V to 1V
NGATE Peak Output Current (<10μs) ..........................1A
Operating Ambient Temperature Range
LTC4267C-1 ............................................. 0°C to 70°C
LTC4267I-1 ..........................................–40°C to 85°C
Junction Temperature ........................................... 150°C
Storage Temperature Range...................–65°C to 150°C
Lead Temperature (Soldering, 10 sec) .................. 300°C
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TJMAX = 150°C, θJA = 90°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC4267CGN-1#PBF
LTC4267IGN-1#PBF
LTC4267CGN-1#TRPBF
LTC4267IGN-1#TRPBF
4267-1
4267I-1
16-Lead Narrow Plastic SSOP
16-Lead Narrow Plastic SSOP
0°C to 70°C
–40°C to 85°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC4267CGN-1
LTC4267IGN-1
LTC4267CGN-1#TR
LTC4267IGN-1#TR
4267-1
4267I-1
16-Lead Narrow Plastic SSOP
16-Lead Narrow Plastic SSOP
0°C to 70°C
–40°C to 85°C
Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges.
*For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. (Note 3)
SYMBOL
PARAMETER
CONDITIONS
VPORTN
Supply Voltage
Maximum Operating Voltage
Signature Range
Classiﬁcation Range
UVLO Turn-On Voltage
UVLO Turn-Off Voltage
Voltage with Respect to VPORTP Pin
(Notes 4, 5, 6)
PVCC Turn-On Voltage
PVCC Turn-Off Voltage
PVCC Hysteresis
Voltage with Respect to PGND
Voltage with Respect to PGND
VTURNON – VTURNOFF
VTURNON
VTURNOFF
VHYST
MIN
●
●
●
●
●
●
●
●
–1.5
–12.5
–34.8
–29.3
7.8
4.6
1.5
TYP
–36.0
–30.5
8.7
5.7
3.0
MAX
UNITS
–57
–9.5
–21
–37.2
–31.5
9.2
6.8
V
V
V
V
V
V
V
V
42671f
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LTC4267-1
ELECTRICAL
CHARACTERISTICS The ● denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at T = 25°C. (Note 3)
A
SYMBOL
PARAMETER
CONDITIONS
VCLAMP1mA
PVCC Shunt Regulator Voltage
IPVCC = 1mA, VITH/RUN = 0V, Voltage
with Respect to PGND
VMARGIN
VCLAMP1mA – VTURNON Margin
VPORTN Supply Current when ON
PVCC Supply Current
Normal Operation
Start-Up
IVPORTN_ON
IPVCC_ON
IVPORTN_CLASS VPORTN Supply Current
During Classiﬁcation
ΔICLASS
Current Accuracy
During Classiﬁcation
RSIGNATURE
Signature Resistance
MIN
TYP
MAX
UNITS
●
8.3
9.4
10.3
V
●
0.05
0.6
⎯ W
⎯ ⎯R⎯G⎯D, SIGDISA Floating
VPORTN = –48V, POUT, P
(Note 7)
VITH/RUN – PGND = 1.3V
PVCC – PGND = VTURNON – 100mV
VPORTN = –17.5V, POUT Tied to VPORTP, RCLASS,
SIGDISA Floating (Note 8)
●
10mA < ICLASS < 40mA, –12.5V ≤ VPORTN ≤ –21V
(Note 9)
●
–1.5V ≤ VPORTN ≤ – 9.5V, POUT Tied to VPORTP,
IEEE 802.3af 2-Point Measurement (Notes 4, 5)
●
●
●
●
RINVALID
Invalid Signature Resistance
–1.5V ≤ VPORTN ≤ – 9.5V, SIGDISA and POUT Tied to
VPORTP, IEEE 802.3af 2-Point Measurement
(Notes 4, 5)
●
VIH
Signature Disable
High Level Input Voltage
With Respect to VPORTN
High Level Invalidates Signature (Note 10)
●
VIL
Signature Disable
Low Level Input Voltage
With Respect to VPORTN
Low Level Enables Signature
●
RINPUT
Signature Disable, Input Resistance
Power Good Output Low Voltage
With Respect to VPORTN
I = 1mA VPORTN = –48V,
⎯P⎯W⎯R⎯G⎯D Referenced to VPORTN
VPORTN = –48V, Voltage between VPORTN and POUT
POUT Falling
POUT Rising
VPORTN = 0V, ⎯P⎯W⎯R⎯G⎯D FET Off, V⎯P⎯W⎯R⎯G⎯D = 57V
I = 300mA, VPORTN = –48V, Measured from
VPORTN to POUT
PVCC – PGND = VTURNON + 100mV
VITH/RUN – PGND = 0V, PVCC – PGND = 8V
Referenced to PGND, PVCC – PGND = 8V (Note 11)
PVCC – PGND = 8V (Note 11)
ITH/RUN Pin Load = ±5μA (Note 11)
VTURNOFF < PVCC < VCLAMP (Note 11)
ITH/RUN Sinking 5μA, PVCC – PGND = 8V (Note 11)
ITH/RUN Sourcing 5μA, PVCC – PGND = 8V (Note 11)
VPORTN = 0V, Power MOSFET Off,
POUT = 57V (Note 12)
VPORTN = –48V, POUT = –43V (Note 13, 14)
VPORTN = –48V, POUT = –43V (Note 13, 14)
VITH/RUN – PGND = 1.3V, PVCC – PGND = 8V
VITH/RUN – PGND = 1.3V, VFB – PGND = 0.8V,
PVCC – PGND = 8V
VITH/RUN – PGND = 1.3V, VFB – PGND = 0.8V,
PVCC – PGND = 8V
●
VPG_OUT
Power Good Trip Point
VPG _FALL
VPG_RISE
IPG_LEAK
RON
Power Good Leakage Current
On-Resistance
VITHSHDN
ITHSTART
VFB
IFB
gm
ΔVO(LINE)
ΔVO(LOAD)
Shutdown Threshold (at ITH/RUN)
Start-Up Current Source at ITH/RUN
Regulated Feedback Voltage
VFB Input Current
Error Ampliﬁer Transconductance
Output Voltage Line Regulation
Output Voltage Load Regulation
IPOUT_LEAK
POUT Leakage
ILIM_HI
ILIM_LO
fOSC
DCON(MIN)
Input Current Limit, High Level
Input Current Limit, Low Level
Oscillator Frequency
Minimum Switch On Duty Cycle
DCON(MAX)
Maximum Switch On Duty Cycle
0.35
240
40
0.5
23.25
9
3
1.3
2.7
1.5
3.0
0.15
0.2
0.780
200
●
±3.5
%
26.00
kΩ
11.8
kΩ
57
V
0.45
V
0.5
kΩ
V
200
6
450
205
240
8
mA
mA
kHz
%
80
90
%
0.28
0.3
0.800
10
333
0.05
3
3
●
●
μA
μA
mA
150
1.0
●
●
350
90
0.65
V
V
μA
Ω
Ω
V
μA
V
nA
μA/V
mV/V
mV/μA
mV/μA
μA
●
●
V
mA
100
●
●
●
3
350
90
180
70
1.7
3.3
1
1.6
2
0.45
0.4
0.812
50
500
42671f
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LTC4267-1
ELECTRICAL CHARACTERISTICS
The ● denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. (Note 3)
SYMBOL
PARAMETER
CONDITIONS
tRISE
tFALL
VIMAX
ISLMAX
tSFST
TSHUTDOWN
NGATE Drive Rise Time
NGATE Drive Fall Time
Peak Current Sense Voltage
Peak Slope Compensation Output Current
Soft-Start Time
Thermal Shutdown Trip Temperature
CLOAD = 3000pF, PVCC – PGND = 8V
CLOAD = 3000pF, PVCC – PGND = 8V
RSL = 0, PVCC – PGND = 8V (Note 15)
PVCC – PGND = 8V (Note 16)
PVCC – PGND = 8V
(Notes 13, 17)
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: PVCC internal clamp circuit self regulates to 9.4V with respect to
PGND.
Note 3: The LTC4267-1 operates with a negative supply voltage in the
range of – 1.5V to – 57V. To avoid confusion, voltages for the PD interface
are always referred to in terms of absolute magnitude. Terms such as
“maximum negative voltage” refer to the largest negative voltage and
a “rising negative voltage” refers to a voltage that is becoming more
negative.
Note 4: The LTC4267-1 is designed to work with two polarity protection
diode drops between the PSE and PD. Parameter ranges speciﬁed in the
Electrical Characteristics section are with respect to this product pins and
are designed to meet IEEE 802.3af speciﬁcations when these diode drops
are included. See the Application Information section.
Note 5: Signature resistance is measured via the two-point ΔV/ΔI method
as deﬁned by IEEE 802.3af. The PD signature resistance is offset from
the 25k to account for diode resistance. With two series diodes, the total
PD resistance will be between 23.75k and 26.25k and meet IEEE 802.3af
speciﬁcations. The minimum probe voltages measured at the LTC4267-1
pins are –1.5V and –2.5V. The maximum probe voltages are –8.5V and
–9.5V.
Note 6: The PD interface includes hysteresis in the UVLO voltages to
preclude any start-up oscillation. Per IEEE 802.3af requirements, the PD
will power up from a voltage source with 20Ω series resistance on the ﬁrst
trial.
Note 7: Dynamic Supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 8: IVPORTN_CLASS does not include classiﬁcation current
programmed at the RCLASS pin. Total current in classiﬁcation mode will be
IVPORTN_CLASS + ICLASS (See note 9).
Note 9: ICLASS is the measured current ﬂowing through RCLASS. ΔICLASS
accuracy is with respect to the ideal current deﬁned as ICLASS = 1.237/
MIN
●
90
TYP
40
40
100
5
1.4
140
MAX
115
UNITS
ns
ns
mV
μA
ms
°C
RCLASS. The current accuracy does not include variations in RCLASS
resistance. The total classiﬁcation current for a PD also includes the IC
quiescent current (IVPORTN_CLASS). See Applications Information.
Note 10: To disable the 25k signature, tie SIGDISA to VPORTP or hold
SIGDISA high with respect to VPORTN. See Applications Information.
Note 11: The switching regulator is tested in a feedback loop that servos
VFB to the output of the error ampliﬁer while maintaining ITH/RUN at the
midpoint of the current limit range.
Note 12: IPOUT_LEAK includes current drawn through POUT by the power
good status circuit. This current is compensated for in the 25k signature
resistance and does not affect PD operation.
Note 13: The LTC4267-1 PD Interface includes thermal protection. In
the event of an overtemperature condition, the PD interface will turn off
the switching regulator until the part cools below the overtemperature
limit. The LTC4267-1 is also protected against thermal damage from
incorrect classiﬁcation probing by the PSE. If the LTC4267-1 exceeds the
overtemperature threshold, the classiﬁcation load current is disabled.
Note 14: The PD interface includes dual level input current limit. At turnon, before the POUT load capacitor is charged, the PD current level is set
to a low level. After the load capacitor is charged and the POUT – VPORTN
voltage difference is below the power good threshold, the PD switches to
high level current limit. The PD stays in high level current limit until the
input voltage drops below the UVLO turn-off threshold.
Note 15: Peak current sense voltage is reduced dependent on duty cycle
and an optional external resistor in series with the SENSE pin (RSL). For
details, refer to the programmable slope compensation feature in the
Applications Information section.
Note 16: Guaranteed by design.
Note 17: The PD interface includes overtemperature protection that is
intended to protect the device from momentary overload conditions.
Junction temperature will exceed 125°C when overtemperature protection
is active. Continuous operation above the speciﬁed maximum operating
junction temperature may impair device reliability.
42671f
4
LTC4267-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Input Current vs Input Voltage
25k Detection Range
Input Current vs Input Voltage
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TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage vs
Temperature
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LTC4267-1
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TYPICAL PERFOR A CE CHARACTERISTICS
Start-Up IPVCC Supply Current vs
Temperature
ITH/RUN Start-Up Current Source
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LTC4267-1
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PI FU CTIO S
PGND (Pin 1, 8, 9, 16): Switching Regulator Negative
Supply. This pin is the negative supply rail for the switching
regulator controller and must be tied to POUT.
is high impedance until the voltage reaches the turn-on
UVLO threshold. The output is then current limited. See
the Application Information section.
ITH/RUN (Pin 2): Current Threshold/Run Input. This
pin performs two functions. It serves as the switching
regulator error ampliﬁer compensation point as well as
the run/shutdown control input. Nominal voltage range is
0.7V to 1.9V. Forcing the pin below 0.28V with respect to
PGND causes the controller to shut down.
⎯P⎯W⎯R⎯G⎯D (Pin 11): Power Good Output, Open-Drain.
Indicates that the PD MOSFET is on and the switching
regulator can start operation. Low impedance indicates
power is good. ⎯P⎯W⎯R⎯G⎯D is high impedance during detection, classiﬁcation and in the event of a thermal overload.
⎯P⎯W⎯R⎯G⎯D is referenced to VPORTN.
NGATE (Pin 3): Gate Driver Output. This pin drives the
regulator’s external N-Channel MOSFET and swings from
PGND to PVCC.
SIGDISA (Pin 12): Signature Disable Input. SIGDISA allows the PD to present an invalid signature resistance and
remain inactive. Connecting SIGDISA to VPORTP lowers
the signature resistance to an invalid value and disables
all functions of the LTC4267-1. If unused, tie SIGDISA to
VPORTN.
PVCC (Pin 4): Switching Regulator Positive Supply. This
pin is the positive supply rail for the switching regulator
and must be closely decoupled to PGND.
RCLASS (Pin 5): Class Select Input. Used to set the current
value the PD maintains during classiﬁcation. Connect a
resistor between RCLASS and VPORTN (see Table 2).
NC (Pin 6): No Internal Connection.
VPORTN (Pin 7): Negative Power Input. Tie to the –48V
input port through the input diodes.
POUT (Pin 10): Power Output. Supplies –48V to the switching regulator PGND pin and any additional PD loads through
an internal power MOSFET that limits input current. POUT
VPORTP (Pin 13): Positive Power Input. Tie to the input
port power return through the input diodes.
SENSE (Pin 14): Current Sense. This pin performs two
functions. It monitors the regulator switch current by reading the voltage across an external sense resistor. It also
injects a current ramp that develops a slope compensation
voltage across an optional external programming resistor.
See the Applications Information section.
VFB (Pin 15): Feedback Input. Receives the feedback voltage
from the external resistor divider across the output.
42671f
8
LTC4267-1
BLOCK DIAGRAM
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APPLICATIO S I FOR ATIO
OVERVIEW
The LTC4267-1 is partitioned into two major blocks: a
Powered Device (PD) interface controller and a current
mode ﬂyback switching regulator. The Powered Device
(PD) interface is intended for use as the front end of a
PD adhering to the IEEE 802.3af standard, and includes
a trimmed 25k signature resistor, classiﬁcation current
source, and an input current limit circuit. With these
functions integrated into the LTC4267-1, the signature
and power interface for a PD can be built that meets all
the requirements of the IEEE 802.3af speciﬁcation with a
minimum of external components.
The switching regulator portion of the LTC4267-1 is a
constant frequency current mode controller that is optimized for Power over Ethernet applications. The regulator
is designed to drive a 6V N-channel MOSFET and features
soft-start and programmable slope compensation. The
integrated error ampliﬁer and precision reference give the
PD designer the option of using a nonisolated topology
without the need for an external ampliﬁer or reference. The
LTC4267-1 has been speciﬁcally designed to interface with
both IEEE compliant Power Sourcing Equipment (PSE)
and legacy PSEs which do not meet the inrush current
requirement of the IEEE 802.3af speciﬁcation. By setting
the initial inrush current limit to a low level, a PD using
the LTC4267-1 minimizes the current drawn from the PSE
during start-up. After powering up, the LTC4267-1 switches
to the high level current limit, thereby allowing the PD to
consume up to 12.95W if an IEEE 802.3af PSE is present.
This low level current limit also allows the LTC4267-1 to
charge arbitrarily large load capacitors without exceeding
the inrush limits of the IEEE 802.3af speciﬁcation. This
dual level current limit provides the system designer with
ﬂexibility to design PDs which are compatible with legacy
PSEs while also being able to take advantage of the higher
power available in an IEEE 802.3af system.
Using an LTC4267-1 for the power and signature interface functions of a PD provides several advantages. The
LTC4267-1 current limit circuit includes an onboard 100V
power MOSFET. This low leakage MOSFET is speciﬁed to
42671f
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LTC4267-1
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Table 1. LTC4267-1 Operational Mode
as a Function of Input Voltage
INPUT VOLTAGE
(VPORTN with RESPECT to VPORTP) LTC4267-1 MODE OF OPERATION
0V to – 1.4V
Inactive
–1.5V to –9.5V**
25k Signature Resistor Detection
–9.8V to –12.4V
Classiﬁcation Load Current Ramps up
from 0% to 100%
–12.5V to UVLO*
Classiﬁcation Load Current Active
UVLO* to –57V
Power Applied to Switching Regulator
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The LTC4267-1 PD interface has several modes of operation depending on the applied input voltage as shown in
Figure 1 and summarized in Table 1. These modes satisfy
the requirements deﬁned in the IEEE 802.3af speciﬁcation.
The input voltage is applied to the VPORTN pin and must
be negative relative to the VPORTP pin. Voltages in the data
sheet for the PD interface portion of the LTC4267-1 are
with respect to VPORTP while the voltages for the switching regulator are referenced to PGND. It is assumed that
PGND is tied to POUT. Note the use of different ground
symbols throughout the data sheet.
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avoid corrupting the 25k signature resistor while also saving board space and cost. In addition, the inrush current
limit requirement of the IEEE 802.3af standard can cause
large transient power dissipation in the PD. The LTC4267-1
is designed to allow multiple turn-on sequences without
overheating the miniature 16-lead package. In the event of
excessive power cycling, the LTC4267-1 provides thermal
overload protection to keep the onboard power MOSFET
within its safe operating area.
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Rising input threshold ≅ – 36.0V
Falling input threshold ≅ –30.5V
**Measured at LTC4267-1 pin. The LTC4267-1 meets the IEEE 802.3af 10V
minimum when operating with the required diode bridges.
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Figure 1. Output Voltage, ⎯P⎯W⎯R⎯G⎯D and PD
Current as a Function of Input Voltage
42671f
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Series Diodes
The IEEE 802.3af-deﬁned operating modes for a PD reference the input voltage at the RJ45 connector on the PD.
The PD must be able to accept power of either polarity
at each of its inputs, so it is common to install diode
bridges (Figure 2). The LTC4267-1 takes this into account
by compensating for these diode drops in the threshold
points for each range of operation. A similar adjustment
is made for the UVLO voltages.
Detection
During detection, the PSE will apply a voltage in the
range of –2.8V to –10V on the cable and look for a 25k
signature resistor. This identiﬁes the device at the end of
the cable as a PD. With the terminal voltage in this range,
the LTC4267-1 connects an internal 25k resistor between
the VPORTP and VPORTN pins. This precision, temperature
compensated resistor presents the proper signature to
alert the PSE that a PD is present and desires power to be
applied. The internal low-leakage UVLO switch prevents
the switching regulator circuitry from affecting the detection signature.
The LTC4267-1 is designed to compensate for the voltage
and resistance effects of the IEEE required diode bridge.
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The signature range extends below the IEEE range to accommodate the voltage drop of the two diodes. The IEEE
speciﬁcation requires the PSE to use a ΔV/ΔI measurement
technique to keep the DC offset of these diodes from affecting the signature resistance measurement. However,
the diode resistance appears in series with the signature
resistor and must be included in the overall signature
resistance of the PD. The LTC4267-1 compensates for
the two series diodes in the signature path by offsetting
the resistance so that a PD built using the LTC4267-1 will
meet the IEEE speciﬁcation.
In some applications it is necessary to control whether
or not the PD is detected. In this case, the 25k signature
resistor can be enabled and disabled with the use of the
SIGDISA pin (Figure 3). Disabling the signature via the
SIGDISA pin will change the signature resistor to 9k
(typical) which is an invalid signature per the IEEE 802.3af
speciﬁcation. This invalid signature is present for PD input
voltages from –2.8V to –10V. If the input rises above –10V,
the signature resistor reverts to 25k to minimize power
dissipation in the LTC4267-1. To disable the signature,
tie SIGDISA to VPORTP. Alternately, the SIGDISA pin can
be driven high with respect to VPORTN. When SIGDISA is
high, all functions of the PD interface are disabled.
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Diode Bridges on Main and Spare Inputs
42671f
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Figure 4. IEEE 802.3af Classiﬁcation Probing
Once the PSE has detected a PD, the PSE may optionally classify the PD. Classiﬁcation provides a method for
more efﬁcient allocation of power by allowing the PSE
to identify lower power PDs and allocate less power for
these devices. The IEEE 802.3af speciﬁcation deﬁnes ﬁve
classes (Table 2) with varying power levels. The designer
selects the appropriate classiﬁcation based on the power
consumption of the PD. For each class, there is an associated load current that the PD asserts onto the line
during classiﬁcation probing. The PSE measures the PD
load current to determine the proper classiﬁcation and
PD power requirements.
During classiﬁcation (Figure 4), the PSE presents a ﬁxed
voltage between –15.5V and –20.5V to the PD. With the
input voltage in this range, the LTC4267-1 asserts a load
current from the VPORTP pin through the RCLASS resistor.
The magnitude of the load current is set by the RCLASS
resistor. The resistor values associated with each class are
shown in Table 2. Note that the switching regulator will
not interfere with the classiﬁcation measurement since the
LTC4267-1 has not passed power to the regulator.
Table 2. Summary of IEEE 802.3af Power Classiﬁcations and
LTC4267-1 RCLASS Resistor Selection
Usage
Maximum
Power Levels
at Input of PD
(W)
0
Default
0.44 to 12.95
<5
Open
1
Optional
0.44 to 3.84
10.5
124
2
Optional
3.84 to 6.49
18.5
68.1
3
Optional
6.49 to 12.95
28
45.3
4
Reserved
Reserved*
40
30.9
Class
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Nominal
Classiﬁcation
Load Current
(mA)
LTC4267-1
RCLASS
Resistor
(Ω, 1%)
*Class 4 is currently reserved and should not be used.
The IEEE 802.3af speciﬁcation limits the classiﬁcation
time to 75ms because a signiﬁcant amount of power is
dissipated in the PD. The LTC4267-1 is designed to handle
the power dissipation for this time period. If the PSE probing exceeds 75ms, the LTC4267-1 may overheat. In this
situation, the thermal protection circuit will engage and
disable the classiﬁcation current source in order to protect
the part. The LTC4267-1 stays in classiﬁcation mode until
the input voltage rises above the UVLO turn-on voltage.
VPORTN Undervoltage Lockout
The IEEE speciﬁcation dictates a maximum turn-on voltage
of 42V and a minimum turn-off voltage of 30V for the PD.
In addition, the PD must maintain large on-off hysteresis to
prevent resistive losses in the wiring between the PSE and
the PD from causing start-up oscillation. The LTC4267-1
incorporates an undervoltage lockout (UVLO) circuit that
monitors the line voltage at VPORTN to determine when
to apply power to the integrated switching regulator
(Figure 5). Before the power is applied to the switching
regulator, the POUT pin is high impedance and sitting at
the ground potential since there is no charge on capacitor
C1. When the input voltage rises above the UVLO turn-on
threshold, the LTC4267-1 removes the detection and classiﬁcation loads and turns on the internal power MOSFET.
C1 charges up under the LTC4267-1 current limit control
and the POUT pin transitions from 0V to VPORTN. This
sequence is shown in Figure 1. The LTC4267-1 includes
a hysteretic UVLO circuit on VPORTN that keeps power
applied to the load until the input voltage falls below the
UVLO turn-off threshold. Once the input voltage drops
below –30V, the internal power MOSFET is turned off and
42671f
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LTC4267-1
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APPLICATIO S I FOR ATIO
the classiﬁcation current is reenabled. C1 will discharge
through the PD circuitry and the POUT pin will go to a high
impedance state.
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Input Current Limit
IEEE 802.3af speciﬁes a maximum inrush current and
also speciﬁes a minimum load capacitor between the
VPORTP and POUT pins. To control turn-on surge current
in the system, the LTC4267-1 integrates a dual level current limit circuit with an onboard power MOSFET and
sense resistor to provide a complete inrush control circuit
without additional external components. At turn-on, the
LTC4267-1 will limit the input current to the low level,
allowing the load capacitor to ramp up to the line voltage
in a controlled manner.
The LTC4267-1 has been speciﬁcally designed to interface
with legacy PSEs which do not meet the inrush current
requirement of the IEEE 802.3af speciﬁcation. At turn-on
the LTC4267-1 current limit is set to the lower level. After
C1 is charged up and the POUT – VPORTN voltage difference
is below the power good threshold, the LTC4267-1 switches
to the high level current limit. The dual level current limit
allows legacy PSEs with limited current sourcing capability
to power up the PD while also allowing the PD to draw full
power from an IEEE 802.3af PSE. The dual level current
limit also allows use of arbitrarily large load capacitors.
The IEEE 802.3af speciﬁcation mandates that at turn-on
the PD not exceed the inrush current limit for more than
50ms. The LTC4267-1 is not restricted to the 50ms time
limit because the load capacitor is charged with a current
below the IEEE inrush current limit speciﬁcation.
As the LTC4267-1 switches from the low to high level
current limit, the current will increase momentarily. This
current spike is a result of the LTC4267-1 charging the
last 1.5V at the high level current limit. When charging a
10μF capacitor, the current spike is typically 100μs wide
and 125% of the nominal low level current limit.
The LTC4267-1 stays in the high level current limit mode
until the input voltage drops below the UVLO turn-off
threshold. This dual level current limit provides the system designer with the ﬂexibility to design PDs which are
compatible with legacy PSEs while also being able to take
advantage of the higher power allocation available in an
IEEE 802.3af system.
During the current limited turn on, a large amount of
power is dissipated in the power MOSFET. The LTC4267-1
PD interface is designed to accept this thermal load and
is thermally protected to avoid damage to the onboard
power MOSFET. Note that in order to adhere to the IEEE
802.3af standard, it is necessary for the PD designer to
ensure the PD steady state power consumption falls within
the limits shown in Table 2. In addition, the steady state
current must be less than ILIM_HI.
Power Good
The LTC4267-1 PD Interface includes a power good circuit
(Figure 6) that is used to indicate that load capacitor C1
is fully charged and that the switching regulator can start
operation. The power good circuit monitors the voltage
across the internal UVLO power MOSFET and ⎯P⎯W⎯R⎯G⎯D is
asserted when the voltage falls below 1.5V. The power
good circuit includes hysteresis to allow the LTC4267-1 to
operate near the current limit point without inadvertently
disabling ⎯P⎯W⎯R⎯G⎯D. The MOSFET voltage must increase to
3V before ⎯P⎯W⎯R⎯G⎯D is disabled.
If a sudden increase in voltage appears on the input line,
this voltage step will be transferred through capacitor C1
and appear across the power MOSFET. The response of the
LTC4267-1 will depend on the magnitude of the voltage
step, the rise time of the step, the value of capacitor C1
and the switching regulator load. For fast rising inputs,
42671f
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LTC4267-1
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Figure 6. LTC4267-1 Power Good
the LTC4267-1 will attempt to quickly charge capacitor C1
using an internal secondary current limit circuit. In this
scenario, the PSE current limit should provide the overall
limit for the circuit. For slower rising inputs, the 375mA
current limit in the LTC4267-1 will set the charge rate of
the capacitor C1. In either case, the ⎯P⎯W⎯R⎯G⎯D signal may
go inactive brieﬂy while the capacitor is charged up to the
new line voltage. In the design of a PD, it is necessary
to determine if a step in the input voltage will cause the
⎯P⎯W⎯R⎯G⎯D signal to go inactive and how to respond to this
event. In some designs, it may be desirable to ﬁlter the
⎯P⎯W⎯R⎯G⎯D signal so that intermittent power bad conditions
are ignored. Figure 7 demonstrates a method to insert a
lowpass ﬁlter on the power good interface.
For PD designs that use a large load capacitor and also consume a lot of power, it is important to delay activation of the
⎯ W
⎯ R
⎯ G
⎯ D
⎯ signal. If the regulator
switching regulator with the P
is not disabled during the current-limited turn-on sequence,
the PD circuitry will rob current intended for charging up
the load capacitor and create a slow rising input, possibly
causing the LTC4267-1 to go into thermal shutdown.
The ⎯P⎯W⎯R⎯G⎯D pin connects to an internal open drain, 100V
transistor capable of sinking 1mA. Low impedance to
VPORTN indicates power is good. ⎯P⎯W⎯R⎯G⎯D is high impedance during signature and classiﬁcation probing and in
the event of a thermal overload. During turn-off, ⎯P⎯W⎯R⎯G⎯D
is deactivated when the input voltage drops below 30V.
In addition, ⎯P⎯W⎯R⎯G⎯D may go active brieﬂy at turn-on for
fast rising input waveforms. ⎯P⎯W⎯R⎯G⎯D is referenced to the
VPORTN pin and when active, will be near the VPORTN potential. Connect the ⎯P⎯W⎯R⎯G⎯D pin to the switching regulator
circuitry as shown in Figure 7.
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Figure 7. Power Good Interface Examples
PD Interface Thermal Protection
The LTC4267-1 PD Interface includes thermal overload
protection in order to provide full device functionality
in a miniature package while maintaining safe operating
temperatures. Several factors create the possibility of
signiﬁcant power dissipation within the LTC4267-1. At
turn-on, before the load capacitor has charged up, the
instantaneous power dissipated by the LTC4267-1 can be
as much as 10W. As the load capacitor charges up, the
power dissipation in the LTC4267-1 will decrease until it
reaches a steady-state value dependent on the DC load
current. The size of the load capacitor determines how fast
the power dissipation in the LTC4267-1 will subside. At
room temperature, the LTC4267-1 can typically handle load
capacitors as large as 800μF without going into thermal
shutdown. With large load capacitors, the LTC4267-1 die
temperature will increase by as much as 50°C during a
single turn-on sequence. If for some reason power were
removed from the part and then quickly reapplied so that
the LTC4267-1 had to charge up the load capacitor again,
the temperature rise would be excessive if safety precautions were not implemented.
The LTC4267-1 PD interface protects itself from thermal
damage by monitoring the die temperature. If the die
42671f
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temperature exceeds the overtemperature trip point, the
current is reduced to zero and very little power is dissipated in the part until it cools below the overtemperature
set point. Once the LTC4267-1 has charged up the load
capacitor and the PD is powered and running, there will
be minor residual heating due to the DC load current of
the PD ﬂowing through the internal MOSFET.
During classiﬁcation, excessive heating of the LTC4267-1
can occur if the PSE violates the 75ms probing time limit.
To protect the LTC4267-1, thermal overload circuitry will
disable classiﬁcation current if the die temperature exceeds
the overtemperature trip point. When the die cools down
below the trip point, classiﬁcation current is reenabled.
The PD is designed to operate at a high ambient temperature and with the maximum allowable supply (57V).
However, there is a limit to the size of the load capacitor
that can be charged up before the LTC4267-1 reaches the
overtemperature trip point. Hitting the overtemperature trip
point intermittently does not harm the LTC4267-1, but it
will delay the completion of capacitor charging. Capacitors
up to 200μF can be charged without a problem over the
full operating temperature range.
Switching Regulator Main Control Loop
Due to space limitations, the basics of current mode
DC/DC conversion will not be discussed here. The reader
is referred to the detail treatment in Application Note 19
or in texts such as Abraham Pressman’s Switching Power
Supply Design.
In a Power over Ethernet System, the majority of applications involve an isolated power supply design. This means
that the output power supply does not have any DC electrical path to the PD interface or the switching regulator
primary. The DC isolation is achieved typically through
a transformer in the forward path and an optoisolator in
the feedback path or a third winding in the transformer.
The typical application circuit shown on the front page
of the datasheet represents an isolated design using an
optoisolator. In applications where a nonisolated topology
is desired, the LTC4267-1 features a feedback port and
an internal error ampliﬁer that can be enabled for this
speciﬁc application.
In the typical application circuit (Figure 11), the isolated
topology employs an external resistive voltage divider
to present a fraction of the output voltage to an external
error ampliﬁer. The error ampliﬁer responds by pulling
an analog current through the input LED on an optoisolator. The collector of the optoisolator output presents a
corresponding current into the ITH/RUN pin via a series
diode. This method generates a feedback voltage on the
ITH/RUN pin while maintaining isolation.
The voltage on the ITH/RUN pin controls the pulse-width
modulator formed by the oscillator, current comparator,
and RS latch. Speciﬁcally, the voltage at the ITH/RUN pin
sets the current comparator’s trip threshold. The current
comparator monitors the voltage across a sense resistor
in series with the source terminal of the external N-Channel MOSFET. The LTC4267-1 turns on the external power
MOSFET when the internal free-running 200kHz oscillator
sets the RS latch. It turns off the MOSFET when the current comparator resets the latch or when 80% duty cycle
is reached, whichever happens ﬁrst. In this way, the peak
current levels through the ﬂyback transformer’s primary
and secondary are controlled by the ITH/RUN voltage.
In applications where a nonisolated topology is desirable
(Figure 11), an external resistive voltage divider can present a fraction of the output voltage directly to the VFB pin
of the LTC4267-1. The divider must be designed so when
the output is at its desired voltage, the VFB pin voltage will
equal the 800mV onboard internal reference. The internal
error ampliﬁer responds by driving the ITH/RUN pin. The
LTC4267-1 switching regulator performs in a similar
manner as described previously.
Regulator Start-Up/Shutdown
The LTC4267-1 switching regulator has two shutdown
mechanisms to enable and disable operation: an undervoltage lockout on the PVCC supply pin and a forced
shutdown whenever external circuitry drives the ITH/RUN
pin low. The LTC4267-1 switcher transitions into and out
of shutdown according to the state diagram (Figure 8).
It is important not to confuse the undervoltage lockout
of the PD interface at VPORTN with that of the switching
regulator at PVCC. They are independent functions.
42671f
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Adjustable Slope Compensation
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Figure 8. LTC4267-1 Switching Regulator
Start-Up/Shutdown State Diagram
The undervoltage lockout mechanism on PVCC prevents
the LTC4267-1 switching regulator from trying to drive
the external N-Channel MOSFET with insufﬁcient gate-tosource voltage. The voltage at the PVCC pin must exceed
VTURNON (nominally 8.7V with respect to PGND) at least
momentarily to enable operation. The PVCC voltage must
fall to VTURNOFF (nominally 5.7V with respect to PGND)
before the undervoltage lockout disables the switching
regulator. This wide UVLO hysteresis range supports
applications where a bias winding on the ﬂyback transformer is used to increase the efﬁciency of the LTC4267-1
switching regulator.
The ITH/RUN can be driven below VITHSHDN (nominally 0.28V
with respect to PGND) to force the LTC4267-1 switching
regulator into shutdown. An internal 0.3μA current source
always tries to pull the ITH/RUN pin towards PVCC. When
the ITH/RUN pin voltage is allowed to exceed VITHSHDN and
PVCC exceeds VTURNON, the LTC4267-1 switching regulator
begins to operate and an internal clamp immediately pulls
the ITH/RUN pin to about 0.7V. In operation, the ITH/RUN
pin voltage will vary from roughly 0.7V to 1.9V to represent
current comparator thresholds from zero to maximum.
Internal Soft-Start
An internal soft-start feature is enabled whenever the
LTC4267-1 switching regulator comes out of shutdown.
Speciﬁcally, the ITH/RUN voltage is clamped and is
prevented from reaching maximum until 1.4ms have
passed. This allows the input current of the PD to rise in a
smooth and controlled manner on start-up and stay within
the current limit requirement of the LTC4267-1 interface.
The LTC4267-1 switching regulator injects a 5μA peak
current ramp out through its SENSE pin which can be
used for slope compensation in designs that require it.
This current ramp is approximately linear and begins at
zero current at 6% duty cycle, reaching peak current at
80% duty cycle. Programming the slope compensation
via a series resistor is discussed in the External Interface
and Component Selection section.
EXTERNAL INTERFACE AND COMPONENT SELECTION
Input Interface Transformer
Nodes on an Ethernet network commonly interface to the
outside world via an isolation transformer (Figure 9). For
PoE devices, the isolation transformer must include a
center tap on the media (cable) side. Proper termination
is required around the transformer to provide correct
impedance matching and to avoid radiated and conducted
emissions. Transformer vendors such as Bel Fuse, Coilcraft, Pulse and Tyco (Table 3) can provide assistance with
selection of an appropriate isolation transformer and proper
termination methods. These vendors have transformers
speciﬁcally designed for use in PD applications.
Table 3. Power over Ethernet Transformer Vendors
VENDOR
CONTACT INFORMATION
Bel Fuse Inc.
206 Van Vorst Street
Jersey City, NJ 07302
Tel: 201-432-0463
FAX: 201-432-9542
http://www.belfuse.com
Coilcraft, Inc.
1102 Silver Lake Road
Cary, IL 60013
Tel: 847-639-6400
FAX: 847-639-1469
http://www.coilcraft.com
Pulse Engineering
12220 World Trade Drive
San Diego, CA 92128
Tel: 858-674-8100
FAX: 858-674-8262
http://www.pulseeng.com
Tyco Electronics
308 Constitution Drive
Menlo Park, CA 94025-1164
Tel: 800-227-7040
FAX: 650-361-2508
http://www.circuitprotection.com
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Diode Bridge
IEEE 802.3af allows power wiring in either of two conﬁgurations: on the TX/RX wires or via the spare wire pairs in
the RJ45 connector. The PD is required to accept power in
either polarity on either the main or spare inputs; therefore
it is common to install diode bridges on both inputs in
order to accommodate the different wiring conﬁgurations.
Figure 9 demonstrates an implementation of these diode
bridges. The IEEE 802.3af speciﬁcation also mandates
that the leakage back through the unused bridge be less
than 28μA when the PD is powered with 57V.
The IEEE standard includes an AC impedance requirement
in order to implement the AC disconnect function. Capacitor C14 in Figure 9 is used to meet this AC impedance
requirement. A 0.1μF capacitor is recommended for this
application.
The LTC4267-1 has several different modes of operation based on the voltage present between VPORTN and
VPORTP pins. The forward voltage drop of the input diodes
in a PD design subtracts from the input voltage and will
3+
59
The input diode bridge of a PD can consume over 4%
of the available power in some applications. It may be
desirable to use Schottky diodes in order to reduce power
loss. However, if the standard diode bridge is replaced
with a Schottky bridge, the transition points between the
modes will be affected. Figure 10 shows a technique for
using Schottky diodes while maintaining proper threshold
points to meet IEEE 802.3af compliance. D13 is added to
compensate for the change in UVLO turn-on voltage caused
by the Schottky diodes and consumes little power.
Classiﬁcation Resistor Selection (RCLASS)
The IEEE speciﬁcation allows classifying PDs into four
distinct classes with class 4 being reserved for future use
(Table 2). An external resistor connected from RCLASS to
VPORTN (Figure 4) sets the value of the load current. The
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affect the transition point between modes. When using
the LTC4267-1, it is necessary to pay close attention to
this forward voltage drop. Selection of oversized diodes
will help keep the PD thresholds from exceeding IEEE
speciﬁcations.
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42671f
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Figure 10. PD Front End with Isolation Transformer, 2nd Schottky Diode Bridge
designer should determine which power category the PD
falls into and then select the appropriate value of RCLASS
from Table 2. If a unique load current is required, the value
of RCLASS can be calculated as:
RCLASS = 1.237V/(IDESIRED – IIN_CLASS)
where IIN_CLASS is the LTC4267-1 IC supply current during
classiﬁcation and is given in the electrical speciﬁcations.
The RCLASS resistor must be 1% or better to avoid degrading the overall accuracy of the classiﬁcation circuit.
Resistor power dissipation will be 50mW maximum and
is transient so heating is typically not a concern. In order
to maintain loop stability, the layout should minimize
capacitance at the RCLASS node. The classiﬁcation circuit
can be disabled by ﬂoating the RCLASS pin. The RCLASS pin
should not be shorted to VPORTN as this would force the
LTC4267-1 classiﬁcation circuit to attempt to source very
large currents and quickly go into thermal shutdown.
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Power Good Interface
Load Capacitor
The ⎯P⎯W⎯R⎯G⎯D signal is controlled by a high voltage, opendrain transistor. The designer has the option of using this
signal to enable the onboard switching regulator through
the ITH/RUN or the PVCC pins. Examples of active-high
interface circuits for controlling the switching regulator
are shown in Figure 7.
The IEEE 802.3af speciﬁcation requires that the PD maintain
a minimum load capacitance of 5μF (provided by C1 in
Figure 11). It is permissible to have a much larger load
capacitor and the LTC4267-1 can charge very large load
capacitors before thermal issues become a problem. The
load capacitor must be large enough to provide sufﬁcient
energy for proper operation of the switching regulator.
However, the capacitor must not be too large or the PD
design may violate IEEE 802.3af requirements.
In some applications, it is desirable to ignore intermittent
power bad conditions. This can be accomplished by including capacitor C15 in Figure 7 to form a lowpass ﬁlter.
With the components shown, power bad conditions less
than about 200μs will be ignored. Conversely, in other
applications it may be desirable to delay assertion of
⎯P⎯W⎯R⎯G⎯D to the switching regulator using CPVCC or C17
as shown in Figure 7.
It is recommended that the designer use the power
good signal to enable the switching regulator. Using
⎯P⎯W⎯R⎯G⎯D ensures the capacitor C1 has reached within
1.5V of the ﬁnal value and is ready to accept a load. The
LTC4267-1 is designed with wide power good hysteresis
to handle sudden ﬂuctuations in the load voltage and
current without prematurely shutting off the switching
regulator. Please refer to the Power-Up Sequencing of the
Application Information section.
Signature Disable Interface
To disable the 25k signature resistor, connect SIGDISA pin
to the VPORTP pin. Alternately, SIGDISA pin can be driven
high with respect to VPORTN. An example of a signature
disable interface is shown in Figure 16, option 2. Note that
the SIGDISA input resistance is relatively large and the
threshold voltage is fairly low. Because of high voltages
present on the printed circuit board, leakage currents from
the VPORTP pin could inadvertently pull SIGDISA high. To
ensure trouble-free operation, use high voltage layout
techniques in the vicinity of SIGDISA. If unused, connect
SIGDISA to VPORTN.
If the load capacitor is too large, there can be a problem
with inadvertent power shutdown by the PSE. Consider
the following scenario. If the PSE is running at –57V
(maximum allowed) and the PD has detected and powered
up, the load capacitor will be charged to nearly –57V. If
for some reason the PSE voltage is suddenly reduced to
–44V (minimum allowed), the input bridge will reverse
bias and the PD power will be supplied by the load capacitor. Depending on the size of the load capacitor and the
DC load of the PD, the PD will not draw any power for
a period of time. If this period of time exceeds the IEEE
802.3af 300ms disconnect delay, the PSE will remove
power from the PD. For this reason, it is necessary to
ensure that inadvertent shutdown cannot occur.
Very small output capacitors (≤10μF) will charge very
quickly in current limit. The rapidly changing voltage at
the output may reduce the current limit temporarily, causing the capacitor to charge at a somewhat reduced rate.
Conversely, charging a very large capacitor may cause the
current limit to increase slightly. In either case, once the
output voltage reaches its ﬁnal value, the input current
limit will be restored to its nominal value.
The load capacitor can store signiﬁcant energy when fully
charged. The design of a PD must ensure that this energy
is not inadvertently dissipated in the LTC4267-1. The polarity-protection diode(s) prevent an accidental short on
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the cable from causing damage. However, if the VPORTN
pin is shorted to VPORTP inside the PD while the capacitor
is charged, current will ﬂow through the parasitic body
diode of the internal MOSFET and may cause permanent
damage to the LTC4267-1.
Maintain Power Signature
In an IEEE 802.3af system, the PSE uses the maintain
power signature (MPS) to determine if a PD continues to
require power. The MPS requires the PD to periodically
draw at least 10mA and also have an AC impedance less
than 26.25kΩ in parallel with 0.05μF. If either the DC
current is less than 10mA or the AC impedance is above
26.25kΩ, the PSE may disconnect power. The DC current
must be less than 5mA and the AC impedance must be
above 2MΩ to guarantee power will be removed.
Selecting Feedback Resistor Values
The regulated output voltage of the switching regulator is
determined by the resistor divider across VOUT (R1 and
R2 in Figure 11) and the error ampliﬁer reference voltage
VREF. The ratio of R2 to R1 needed to produce the desired
voltage can be calculated as:
R2 = R1 • (VOUT – VREF)/VREF
In an isolated power supply application, VREF is determined
by the designer’s choice of an external error ampliﬁer.
Commercially available error ampliﬁers or programmable
shunt regulators may include an internal reference of
1.25V or 2.5V. Since the LTC4267-1 internal reference
and error ampliﬁer are not used in an isolated design, tie
the VFB pin to PGND.
In a nonisolated power supply application, the LTC4267-1
onboard internal reference and error ampliﬁer can be
used. The resistor divider output can be tied directly to
the VFB pin. The internal reference of the LTC4267-1 is
0.8V nominal.
Choose resistance values for R1 and R2 to be as large as
possible to minimize any efﬁciency loss due to the static
current drawn from VOUT, but just small enough so that
when VOUT is in regulation, the error caused by the nonzero
input current from the output of the resistor divider to the
error ampliﬁer pin is less than 1%.
Error Ampliﬁer and Optoisolator Considerations
In an isolated topology, the selection of the external error
ampliﬁer depends on the output voltage of the switching
regulator. Typical error ampliﬁers include a voltage reference of either 1.25V or 2.5V. The output of the ampliﬁer
and the ampliﬁer upper supply rail are often tied together
internally. The supply rail is usually speciﬁed with a wide
upper voltage range, but it is not allowed to fall below the
reference voltage. This can be a problem in an isolated
switcher design if the ampliﬁer supply voltage is not properly managed. When the switcher load current decreases
and the output voltage rises, the error ampliﬁer responds
by pulling more current through the LED. The LED voltage
can be as large as 1.5V, and along with RLIM, reduces the
supply voltage to the error ampliﬁer. If the error amp does
not have enough headroom, the voltage drop across the
LED and RLIM may shut the ampliﬁer off momentarily,
causing a lock-up condition in the main loop. The switcher
will undershoot and not recover until the error ampliﬁer
releases its sink current. Care must be taken to select the
reference voltage and RLIM value so that the error ampliﬁer
always has enough headroom. An alternate solution that
avoids these problems is to utilize the LT1431 or LT4430
where the output of the error ampliﬁer and ampliﬁer supply
rail are brought out to separate pins.
The PD designer must also select an optoisolator such
that its bandwidth is sufﬁciently wider than the bandwidth
of the main control loop. If this step is overlooked, the
main control loop may be difﬁcult to stabilize. The output
42671f
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collector resistor of the optoisolator can be selected for
an increase in bandwidth at the cost of a reduction in gain
of this stage.
Output Transformer Design Considerations
Since the external feedback resistor divider sets the
output voltage, the PD designer has relative freedom in
selecting the transformer turns ratio. The PD designer
can use simple ratios of small integers (i.e. 1:1, 2:1, 3:2)
which yields more freedom in setting the total turns and
mutual inductance and may allow the use of an off the
shelf transformer.
Transformer leakage inductance on either the primary or
secondary causes a voltage spike to occur after the output
switch (Q1 in Figure 11) turns off. The input supply voltage plus the secondary-to-primary referred voltage of the
ﬂyback pulse (including leakage spike) must not exceed
the allowed external MOSFET breakdown rating. This spike
is increasingly prominent at higher load currents, where
more stored energy must be dissipated. In some cases,
a “snubber” circuit will be required to avoid overvoltage
breakdown at the MOSFET’s drain node. Application
Note 19 is a good reference for snubber design.
Current Sense Resistor Consideration
The external current sense resistor (RSENSE in Figure 11)
allows the designer to optimize the current limit behavior
for a particular application. As the current sense resistor
is varied from several ohms down to tens of milliohms,
peak swing current goes from a fraction of an ampere to
several amperes. Care must be taken to ensure proper
circuit operation, especially for small current sense resistor values.
Choose RSENSE such that the switching current exercises
the entire range of the ITH/RUN voltage. The nominal voltage
range is 0.7V to 1.9V and RSENSE can be determined by
experiment. The main loop can be temporarily stabilized
by connecting a large capacitor on the power supply. Apply
the maximum load current allowable at the power supply output based on the class of the PD. Choose RSENSE
such that ITH/RUN approaches 1.9V. Finally, exercise the
output load current over the entire operating range and
ensure that ITH/RUN voltage remains within the 0.7V to
1.9V range. Layout is critical around the RSENSE resistor.
For example, a 0.020Ω sense resistor, with one milliohm
(0.001Ω) of parasitic resistance will cause a 5% reduction
in peak switch current. The resistance of printed circuit
copper traces cannot necessarily be ignored and good
layout techniques are mandatory.
Programmable Slope Compensation
The LTC4267-1 switching regulator injects a ramping
current through its SENSE pin into an external slope
compensation resistor (RSL in Figure 11). This current
ramp starts at zero after the NGATE pin has been high for
the LTC4267-1’s minimum duty cycle of 6%. The current
rises linearly towards a peak of 5μA at the maximum duty
cycle of 80%, shutting off once the NGATE pin goes low.
A series resistor (RSL) connecting the SENSE pin to the
current sense resistor (RSENSE) develops a ramping voltage drop. From the perspective of the LTC4267-1 SENSE
pin, this ramping voltage adds to the voltage across the
sense resistor, effectively reducing the current comparator
threshold in proportion to duty cycle. This stabilizes the
control loop against subharmonic oscillation. The amount
of reduction in the current comparator threshold (ΔVSENSE)
can be calculated using the following equation:
ΔVSENSE = 5μA • RSL • [(Duty Cycle – 6%)/74%]
Note: The LTC4267-1 enforces 6% < Duty Cycle < 80%.
Designs not needing slope compensation may replace
RSL with a short-circuit.
Applications Employing a Third Transformer Winding
A standard operating topology may employ a third winding on the transformer’s primary side that provides power
42671f
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42671f
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to the LTC4267-1 switching regulator via its PVCC pin
(Figure 11). However, this arrangement is not inherently
self-starting. Start-up is usually implemented by the use of
an external “trickle-charge” resistor (RSTART) in conjunction with the internal wide hysteresis undervoltage lockout
circuit that monitors the PVCC pin voltage.
RSTART is connected to VPORTP and supplies a current,
typically 100μA, to charge CPVCC. After some time, the
voltage on CPVCC reaches the PVCC turn-on threshold. The
LTC4267-1 switching regulator then turns on abruptly and
draws its normal supply current. The NGATE pin begins
switching and the external MOSFET (Q1) begins to deliver
power. The voltage on CPVCC begins to decline as the
switching regulator draws its normal supply current, which
exceeds the delivery from RSTART. After some time, typically
tens of milliseconds, the output voltage approaches the
desired value. By this time, the third transformer winding
is providing virtually all the supply current required by the
LTC4267-1 switching regulator.
One potential design pitfall is under-sizing the value of
capacitor CPVCC. In this case, the normal supply current
drawn through PVCC will discharge CPVCC rapidly before the
third winding drive becomes effective. Depending on the
particular situation, this may result in either several off-on
cycles before proper operation is reached or permanent
relaxation oscillation at the PVCC node.
Resistor RSTART should be selected to yield a worst-case
minimum charging current greater that the maximum rated
LTC4267-1 start-up current to ensure there is enough current to charge CPVCC to the PVCC turn-on threshold. RSTART
should also be selected large enough to yield a worst-case
maximum charging current less than the minimum-rated
PVCC supply current, so that in operation, most of the
PVCC current is delivered through the third winding. This
results in the highest possible efﬁciency.
Capacitor CPVCC should then be made large enough to avoid
the relaxation oscillation behavior described previously.
This is difﬁcult to determine theoretically as it depends on
the particulars of the secondary circuit and load behavior.
Empirical testing is recommended.
The third transformer winding should be designed so
that its output voltage, after accounting for the forward
diode voltage drop, exceeds the maximum PVCC turn-off
threshold. Also, the third winding’s nominal output voltage
should be at least 0.5V below the minimum rated PVCC
clamp voltage to avoid running up against the LTC4267-1
shunt regulator, needlessly wasting power.
PVCC Shunt Regulator
In applications including a third transformer winding,
the internal PVCC shunt regulator serves to protect the
LTC4267-1 switching regulator from overvoltage transients
as the third winding is powering up.
If a third transformer winding is undesirable or unavailable, the shunt regulator allows the LTC4267-1 switching
regulator to be powered through a single dropping resistor
from VPORTP as shown in Figure 12. This simplicity comes
at the expense of reduced efﬁciency due to static power
dissipation in the RSTART dropping resistor.
The shunt regulator can sink up to 5mA through the PVCC
pin to PGND. The values of RSTART and CPVCC must be
selected for the application to withstand the worst-case
load conditions and drop on PVCC, ensuring that the PVCC
turn-off threshold is not reached. CPVCC should be sized
sufﬁciently to handle the switching current needed to drive
NGATE while maintaining minimum switching voltage.
External Preregulator
The circuit in Figure 13 shows a third way to power the
LTC4267-1 switching regulator circuit. An external series
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Figure 12. Powering the LTC4267-1 Switching
Regulator via the Shunt Regulator
virtually all the supply current required by the LTC4267-1
switching regulator. CPVCC should be sized sufﬁciently to
handle the switching current needed to drive NGATE while
maintaining minimum switching voltage.
The external preregulator has improved efﬁciency over
the simple resistor-shunt regulator method mentioned
previously. RB can be selected so that it provides a small
current necessary to maintain the zener diode voltage and
the maximum possible base current Q1 will encounter. The
actual current needed to power the LTC4267-1 switching
regulator goes through Q1 and PVCC sources current on
an “as-needed” basis. The static current is then limited
only to the current through RB and D1.
Compensating the Main Loop
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In an isolated topology, the compensation point is typically
chosen by the components conﬁgured around the external
error ampliﬁer. Shown in Figure 14, a series RC network
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Regulator with an External Preregulator
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preregulator consists of a series pass transistor Q1, zener
diode D1, and a bias resistor RB. The preregulator holds
PVCC at 7.6V nominal, well above the maximum rated PVCC
turn-off threshold of 6.8V. Resistor RSTART momentarily
charges the PVCC node up to the PVCC turn-on threshold,
enabling the switching regulator. The voltage on CPVCC
begins to decline as the switching regulator draws its
normal supply current, which exceeds the delivery of
RSTART. After some time, the output voltage approaches
the desired value. By this time, the pass transistor Q1
catches the declining voltage on the PVCC pin, and provides
Figure 14. Main Loop Compensation for an Isolated Design
is connected from the compare voltage of the error ampliﬁer to the error ampliﬁer output. In PD designs where
transient load response is not critical, replace RZ with a
short. The product of R2 and CC should be sufﬁciently large
to ensure stability. When fast settling transient response
is critical, introduce a zero set by RZCC. The PD designer
must ensure that the faster settling response of the output
voltage does not compromise loop stability.
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In a nonisolated design, the LTC4267-1 incorporates an
internal error ampliﬁer where the ITH/RUN pin serves as
a compensation point. In a similar manner, a series RC
network can be connected from ITH/RUN to PGND as
shown in Figure 15. CC and RZ are chosen for optimum
load and line transient response.
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Selecting the Switching Transistor
With the N-channel power MOSFET driving the primary of
the transformer, the inductance will cause the drain of the
MOSFET to traverse twice the voltage across VPORTP and
PGND. The LTC4267-1 operates with a maximum supply
of – 57V; thus the MOSFET must be rated to handle 114V
or more with sufﬁcient design margin. Typical transistors have 150V ratings while some manufacturers have
developed 120V rated MOSFETs speciﬁcally for Powerover-Ethernet applications.
The NGATE pin of the LTC4267-1 drives the gate of the
N-channel MOSFET. NGATE will traverse a rail-to-rail voltage from PGND to PVCC. The designer must ensure the
MOSFET provides a low “ON” resistance when switched
to PVCC as well as ensure the gate of the MOSFET can
handle the PVCC supply voltage.
For high efﬁciency applications, select an N-channel
MOSFET with low total gate charge. The lower total gate
charge improves the efﬁciency of the NGATE drive circuit
and minimizes the switching current needed to charge
and discharge the gate.
Auxiliary Power Source
In some applications, it may be desirable to power the
PD from an auxiliary power source such as a wall transformer. The auxiliary power can be injected into the PD
at several locations and various trade-offs exist. Power
can be injected at the 3.3V or 5V output of the isolated
power supply with the use of a diode ORing circuit. This
method accesses the internal circuits of the PD after the
isolation barrier and therefore meets the 802.3af isolation safety requirements for the wall transformer jack on
the PD. Power can also be injected into the PD interface
portion of the LTC4267-1. In this case, it is necessary to
ensure the user cannot access the terminals of the wall
transformer jack on the PD since this would compromise
the 802.3af isolation safety requirements.
Figure 16 demonstrates three methods of diode ORing
external power into a PD. Option 1 inserts power before
the LTC4267-1 interface controller while options 2 and
3 bypass the LTC4267-1 interface controller section and
power the switching regulator directly.
If power is inserted before the LTC4267-1 interface controller, it is necessary for the wall transformer to exceed
the LTC4267-1 UVLO turn-on requirement and include a
transient voltage suppressor (TVS) to limit the maximum
voltage to 57V. This option provides input current limit
for the transformer, provides a valid power good signal,
and simpliﬁes power priority issues. As long as the wall
transformer applies power to the PD before the PSE, it
will take priority and the PSE will not power up the PD
because the wall power will corrupt the 25k signature. If
the PSE is already powering the PD, the wall transformer
power will be in parallel with the PSE. In this case, priority will be given to the higher supply voltage. If the wall
transformer voltage is higher, the PSE should remove the
line voltage since no current will be drawn from the PSE.
On the other hand, if the wall transformer voltage is lower,
the PSE will continue to supply power to the PD and the
wall transformer will not be used. Proper operation should
occur in either scenario.
42671f
25
LTC4267-1
U
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APPLICATIO S I FOR ATIO
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Figure 16. Auxiliary Power Source for PD
42671f
26
LTC4267-1
U
W
U U
APPLICATIO S I FOR ATIO
If auxiliary power is applied directly to the LTC4267-1
switching regulator (bypassing the LTC4267-1 PD interface), a different set of tradeoffs arise. In the conﬁguration
shown in option 2, the wall transformer does not need
to exceed the LTC4267-1 turn-on UVLO requirement;
however, it is necessary to include diode D9 to prevent
the transformer from applying power to the LTC4267-1
interface controller. The transformer voltage requirement
will be governed by the needs of the onboard switching
regulator. However, power priority issues require more
intervention. If the wall transformer voltage is below
the PSE voltage, then priority will be given to the PSE
power. The LTC4267-1 interface controller will draw power
from the PSE while the transformer will sit unused. This
conﬁguration is not a problem in a PoE system. On the
other hand, if the wall transformer voltage is higher than
the PSE voltage, the LTC4267-1 switching regulator will
draw power from the transformer. In this situation, it is
necessary to address the issue of power cycling that may
occur if a PSE is present. The PSE will detect the PD and
apply power. If the switcher is being powered by the wall
transformer, then the PD will not meet the minimum load
requirement and the PSE will subsequently remove power.
The PSE will again detect the PD and power cycling will
start. With a transformer voltage above the PSE voltage,
it is necessary to either disable the signature, as shown
in option 2, or install a minimum load on the output of the
LTC4267-1 interface to prevent power cycling.
The third option also applies power directly to the
LTC4267-1 switching regulator, bypassing the LTC4267-1
interface controller and omitting diode D9. With the
diode omitted, the transformer voltage is applied to the
LTC4267-1 interface controller in addition to the switching
regulator. For this reason, it is necessary to ensure that the
transformer maintain the voltage between 38V and 57V
to keep the LTC4267-1 interface controller in its normal
operating range. The third option has the advantage of
automatically disabling the 25k signature resistor when
the external voltage exceeds the PSE voltage.
Power-Up Sequencing the LTC4267-1
The LTC4267-1 consists of two functional cells, the PD
interface and the switching regulator, and the power up
sequencing of these two cells must be carefully considered.
The PD designer should ensure that the switching regulator
does not begin operation until the interface has completed
charging up the load capacitor. This will ensure that the
switcher load current does not compete with the load
capacitor charging current provided by the PD interface
current limit circuit. Overlooking this consideration may
result in slow power supply ramp up, power-up oscillation,
and possibly thermal shutdown.
The LTC4267-1 includes a power good signal in the PD
interface that can be used to indicate to the switching
regulator that the load capacitor is fully charged and ready
to handle the switcher load. Figure 7 shows two examples
of ways the ⎯P⎯W⎯R⎯G⎯D signal can be used to control the
switching regulator. The ﬁrst example employs an N-channel MOSFET to drive the ITH/RUN port below the shutdown
threshold (typically 0.28V). The second example drives
PVCC below the PVCC turn-off threshold. Employing the
second example has the added advantage of adding delay
to the switching regulator start-up beyond the time the
power good signal becomes active. The second example
ensures additional timing margin at start-up without the
need for added delay components. In applications where it
is not desirable to utilize the power good signal, sufﬁcient
timing margin can be achieved with RSTART and CPVCC.
RSTART and CPVCC should be set to a delay of two to three
times longer than the duration needed to charge up C1.
Layout Considerations for the LTC4267-1
The most critical layout considerations for the LTC4267-1
are the placement of the supporting external components
associated with the switching regulator. Efﬁciency, stability,
and load transient response can deteriorate without good
layout practices around critical components.
42671f
27
LTC4267-1
U
W
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APPLICATIO S I FOR ATIO
For the LTC4267-1 switching regulator, the current loop
through C1, T1 primary, Q1, and RSENSE must be given
careful layout attention. (Refer to Figure 11.) Because of
the high switching current circulating in this loop, these
components should be placed in close proximity to each
other. In addition, wide copper traces or copper planes
should be used between these components. If vias are
necessary to complete the connectivity of this loop,
placing multiple vias lined perpendicular to the ﬂow of
current is essential for minimizing parasitic resistance and
reducing current density. Since the switching frequency
and the power levels are substantial, shielding and high
frequency layout techniques should be employed. A low
current, low impedance alternate connection should be
employed between the PGND pins of the LTC4267-1 and
the PGND side of RSENSE, away from the high current loop.
This Kelvin sensing will ensure an accurate representation
of the sense voltage is measured by the LTC4267-1.
The placement of the feedback resistors R1 and R2 as
well as the compensation capacitor CC is very important
in the accuracy of the output voltage, the stability of the
main control loop, and the load transient response. In
an isolated design application, R1, R2, and CC should be
placed as close as possible to the error ampliﬁer’s input
with minimum trace lengths and minimum capacitance.
In a nonisolated application, R1, and R2 should be placed
as close as possible to the VFB pin of the LTC4267-1
and CC should be placed close to the ITH/RUN pin of the
LTC4267-1.
In essence, a tight overall layout of the high current loop
and careful attention to current density will ensure successful operation of the LTC4267-1 in a PD.
The PD interface section of the LTC4267-1 is relatively immune to layout problems. Excessive parasitic capacitance
on the RCLASS pin should be avoided. The SIGDISA pin is
adjacent to the VPORTP pin and any coupling, whether resistive or capacitive may inadvertently disable the signature
resistance. To ensure consistent behavior, the SIGDISA
pin should be electrically connected and not left ﬂoating.
Voltages in a PD can be as large as –57V, so high voltage
layout techniques should be employed.
Electro Static Discharge and Surge Protection
The LTC4267-1 is speciﬁed to operate with an absolute
maximum voltage of –100V and is designed to tolerate
brief overvoltage events. However, the pins that interface
to the outside world (primarily VPORTN and VPORTP) can
routinely see peak voltages in excess of 10kV. To protect
the LTC4267-1, it is highly recommended that a transient
voltage suppressor be installed between the diode bridge
and the LTC4267-1 (D3 in Figure 2).
42671f
28
LTC4267-1
U
TYPICAL APPLICATIO S
Class 3 PD with 5V Nonisolated Power Supply
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TYPICAL APPLICATIO S
42671f
U
LTC4267-1
U
PACKAGE DESCRIPTIO
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.045 ±.005
16 15 14 13 12 11 10 9
.254 MIN
.009
(0.229)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ± .0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
RECOMMENDED SOLDER PAD LAYOUT
1
.015 ± .004
× 45°
(0.38 ± 0.10)
.007 – .0098
(0.178 – 0.249)
2 3
4
5 6
7
.0532 – .0688
(1.35 – 1.75)
8
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN16 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
42671f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC4267-1
TYPICAL APPLICATION
High-Efﬁciency Class 3 PD with 3.3V Isolated Power Supply
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RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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Burst Mode is a registered trademark of Linear Technology Corporation. No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
42671f
32
Linear Technology Corporation
LT 0207 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
●
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007

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