MIC2168A 1MHz PWM Synchronous Buck Control IC General Description Features The MIC2168A is a high-efficiency, simple to use 1MHz PWM synchronous buck control IC housed in a small MSOP-10 package. The MIC2168A allows compact DC/DC solutions with a minimal external component count and cost. The MIC2168A operates from a 3V to 14.5V input, without the need of any additional bias voltage. The output voltage can be precisely regulated down to 0.8V. The adaptive all N-Channel MOSFET drive scheme allows efficiencies over 95% across a wide load range. The MIC2168A senses current across the high-side NChannel MOSFET, eliminating the need for an expensive and lossy current-sense resistor. Current limit accuracy is maintained by a positive temperature coefficient that tracks the increasing RDS(ON) of the external MOSFET. Further cost and space are saved by the internal in-rush-current limiting digital soft-start. The MIC2168A is available in a 10-pin MSOP package, with a wide junction operating range of –40°C to +125°C. All support documentation can be found on Micrel’s web site at www.micrel.com. • • • • • • • • • • • • • • 3V to 14.5V input voltage range Adjustable output voltage down to 0.8V Up to 95% efficiency 1MHz PWM operation Adjustable current-limit senses high-side N-Channel MOSFET current No external current sense resistor Adaptive gate drive increases efficiency Ultra-fast response with hysteretic transient recovery mode Overvoltage protection protects the load in fault conditions Dual mode current limit speeds up recovery time Hiccup mode short-circuit protection Internal soft-start Dual function COMP and EN pin allows low-power shutdown Small size MSOP 10-lead package Applications • • • • • • • Point-of-load DC/DC conversion Set-top boxes Graphic cards LCD power supplies Telecom power supplies Networking power supplies Cable modems and routers Typical Application VIN = 5V SD103BWS 100µF 0.1µF VDD VIN 100nF 100pF 4kΩ BST CS HSD MIC2168A VSW COMP/EN GND LSD 85 80 1kΩ 2Ω 1000nF MIC2168 Efficiency 100 95 90 4.7µF IRF7821 1.2µH 75 3.3V 10kΩ IRF7821 3.24kΩ V IN = 5V V OUT = 3.3V 60 150µF x 2 FB 70 65 55 50 0 2 4 6 I LOAD (A) 8 10 MIC2168A Adjustable Output 1MHz Converter Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com April 2005 1 M9999-062205 MIC2168A Micrel, Inc. Ordering Information Part Number Standard Pb-Free MIC2168ABMM MIC2168AYMM Frequency Junction Temp. Range Package 1MHz -40°C to +125°C 10-Lead MSOP Pin Configuration VIN 1 10 BST VDD 2 9 HSD CS 3 8 VSW COMP/EN 4 7 LSD FB 5 6 GND 10-Pin MSOP (MM) Pin Description Pin Number Pin Name Pin Function 1 VIN Supply Voltage (Input): 3V to 14.5V. 2 VDD 5V Internal Linear Regulator (Output): VDD is the external MOSFET gate drive supply voltage and an internal supply bus for the IC. When VIN is <5V, this regulator operates in dropout mode. 3 CS Current Sense / Enable (Input): Current-limit comparator noninverting input. The current limit is sensed across the MOSFET during the ON time. The current can be set by the resistor in series with the CS pin. 4 COMP/EN 5 FB 6 GND Ground (Return). 7 LSD Low-Side Drive (Output): High-current driver output for external synchronous MOSFET. 8 VSW Switch (Return): High-side MOSFET driver return. 9 HSD 10 BST High-Side Drive (Output): High-current output-driver for the high-side MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold-rated MOSFETs should be used. At VIN > 5V, 5V threshold MOSFETs should be used. M9999-062205 Compensation (Input): Dual function pin. Pin for external compensation. If this pin is pulled below 0.2V, with the reference fully up the device shuts down (50µA typical current draw). Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V. Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The gate-drive voltage is higher than the source voltage by VIN minus a diode drop. 2 April 2005 MIC2168A Micrel, Inc. Absolute Maximum Ratings(1) Operating Ratings(2) Supply Voltage (VIN) ................................................... 15.5V Booststrapped Voltage (VBST) .................................VIN +5V Junction Temperature (TJ) ...................–40°C ≤ TJ ≤+125°C Storage Temperature (TS) ........................ –65°C to +150°C Supply Voltage (VIN) ..................................... +3V to +14.5V Output Voltage Range .......................... 0.8V to VIN × DMAX Package Thermal Resistance θJA 10-lead MSOP ............................................. 180°C/W Electrical Characteristics(3) TJ = 25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C Parameter Condition Min Typ Max Units Feedback Voltage Reference (± 1%) 0.792 0.8 0.808 V Feedback Voltage Reference (± 2% over temp) 0.784 0.8 0.816 V 30 100 nA Feedback Bias Current Output Voltage Line Regulation 0.03 %/V Output Voltage Load Regulation 0.5 % 0.6 % Output Voltage Total Regulation Oscillator Section 3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT = 2.5V)(4) Oscillator Frequency 900 Maximum Duty Cycle 90 Minimum On-Time(4) Input and VDD Supply 1000 1100 kHz % 30 60 ns PWM Mode Supply Current VCS = VIN –0.25V; VFB = 0.7V (output switching but excluding external MOSFET gate current.) 1.6 3 mA Shutdown Quiescent Current VCOMP/EN = 0V 50 150 µA 0.25 0.4 VCOMP Shutdown Threshold VCOMP Shutdown Blanking Period CCOMP = 100nF Digital Supply Voltage (VDD) VIN ≥ 6V 0.1 4 4.7 5 V ms 5.3 V Notes: 1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max), the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die temperature, and the regulator will go into thermal shutdown. 2. Devices are ESD sensitive, handling precautions required. 3. Specification for packaged product only. 4. Guaranteed by design. April 2005 3 M9999-062205 MIC2168A Micrel, Inc. Electrical Characteristics(5) Parameter Condition Min Typ Max Units Error Amplifier DC Gain 70 dB Transconductance 1 ms 8.5 µA Soft-Start Soft-Start Current After timeout of internal timer. See “Soft-Start” section. Current Sense CS Over Current Trip Point VCS = VIN –0.25V 160 Temperature Coefficient 200 240 µA +1800 ppm/°C +3 % Lower Threshold, VFB_UVT (relative to VFB) (relative to VFB) –3 % Rise/Fall Time Into 3000pF at VIN > 5V 30 Output Fault Correction Thresholds Upper Threshold, VFB_OVT Gate Drivers Output Driver Impedance 6 Ω 6 Ω Source, VIN = 3V 10 Ω Sink, VIN = 5V Driver Non-Overlap Time ns Source, VIN = 5V Sink, VIN = 3V 10 Note 6 10 20 Ω ns Notes: 5. Specification for packaged product only. 6. Guaranteed by design. M9999-062205 4 April 2005 MIC2168A Micrel, Inc. Typical Characteristics VIN = 5V PWM Mode Supply Current vs. Temperature 2.9 2.7 2.5 2.3 2.1 1.9 1.7 1.5 1.3 1.1 0.9 0.7 0.5 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (°C) 2.0 PWM Mode Supply Current vs. Supply Voltage 0.815 0.810 1.5 0.805 0.800 0.795 1.0 0.790 0.785 0.5 0 V FB vs. Temperature 0.820 5 10 SUPPLY VOLTAGE (V) 15 0.780 0 5 10 15 V IN (V) V DD Line Regulation 6 0.815 V FB Line Regulation 0.820 V DD Load Regulation 5.01 4.99 5 0.810 4.97 4 0.805 0.800 4.95 4.93 3 0.795 4.91 2 0.790 4.89 1 0.785 0.780 -60 -30 0 30 60 90 120 150 TEMPERATURE (°C) V DD Line Regulation vs. Temperature 5.0 0 0 5 10 15 1050 0.5 1000 0 2.0 950 -0.5 900 -1.0 850 0.5 0.0 -60 -30 0 30 60 90 120 150 TEMPERATURE (°C) Current Limit Foldback 4 800 -60 -30 240 30 1.0 3.0 2.5 1.0 5 10 15 20 25 LOAD CURRENT (mA) Oscillator Frequency vs. Supply Voltage 1.5 1100 1.5 0 Oscillator Frequency vs. Temperature 1150 3.5 4.85 V IN (V) 1200 4.5 4.0 4.87 0 30 60 90 120 150 TEMPERATURE (°C) -1.5 0 5 10 SUPPLY VOLTAGE (V) 15 Overcurrent Trip Point vs. Temperature 220 3 200 180 2 160 1 0 April 2005 0 Top MOSFET = Si4800 140 R CS = 1kΩ 120 2 4 6 I LOAD (A) 8 10 100 -60 -30 0 30 60 90 120 150 TEMPERATURE (°C) 5 M9999-062205 MIC2168A Micrel, Inc. Functional Diagram CIN RCS VIN CS VDD 5V LDO D1 Current Limit Comparator VDD 5V High-Side Driver 5V Bandgap Reference BG Valid SW Clamp & Startup Current Ramp Clock CBST 4W RSW Driver Logic 5V Soft-Start & Digital Delay Counter Q1 BOOST Current Limit Reference 0.8V HSD L1 VOUT COUT 5V Low-Side Driver LSD Q2 PWM Comparator Enable Error Loop 0.8V VREF +3% VREF 3% Error Amp FB Hys Comparator R3 R2 MIC2168A COMP C2 GND C1 R1 MIC2168A Block Diagram Functional Description version of VOUT to be slightly less than the reference voltage causing the output voltage of the error amplifier to go high. This will cause the PWM comparator to increase tON time of the top side MOSFET, causing the output voltage to go up and bringing VOUT back in regulation. Soft-Start The COMP/EN pin on the MIC2168A is used for the following three functions: 1. Disables the part by grounding this pin 2. External compensation to stabilize the voltage control loop 3. Soft-start For better understanding of the soft-start feature, let’s assume VIN = 12V, and the MIC2168A is allowed to power-up by un-grounding the COMP/EN pin. The COMP pin has an internal 8.5µA current source that charges the external compensation capacitor. As soon as this voltage rises to 180mV (t = Cap_COMP × 0.18V/8.5µA), the MIC2168A allows the internal VDD linear regulator to power up and as soon as it crosses the undervoltage lockout of 2.6V, the chip’s internal oscillator starts switching. At this point in time, the COMP pin current source increases to 40µA and an internal 11-bit counter starts counting which takes approximately 2ms to complete. During counting, the COMP voltage is clamped at 0.65V. After this counting cycle the COMP current source The MIC2168A is a voltage mode, synchronous step-down switching regulator controller designed for high output power without the use of an external sense resistor. It includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time, a PWM generator, a reference voltage, two MOSFET drivers, and short-circuit current limiting circuitry to form a complete 1MHz switching regulator. Theory of Operation The MIC2168A is a voltage mode step-down regulator. The figure above illustrates the block diagram for the voltage control loop. The output voltage variation due to load or line changes will be sensed by the inverting input of the transconductance error amplifier via the feedback resistors R3, and R2 and compared to a reference voltage at the non-inverting input. This will cause a small change in the DC voltage level at the output of the error amplifier which is the input to the PWM comparator. The other input to the comparator is a 0 to 1V triangular waveform. The comparator generates a rectangular waveform whose width tON is equal to the time from the start of the clock cycle t0 until t1, the time the triangle crosses the output waveform of the error amplifier. To illustrate the control loop, let us assume the output voltage drops due to sudden load turn-on, this would cause the inverting input of the error amplifier which is divided down M9999-062205 6 April 2005 MIC2168A Micrel, Inc. is reduced to 8.5µA and the COMP pin voltage rises from 0.65V to 0.95V, the bottom edge of the saw-tooth oscillator. This is the beginning of 0% duty cycle and it increases slowly causing the output voltage to rise slowly. The MIC2168A has two hysteretic comparators that are enabled when VOUT is within ±3% of steady state. When the output voltage reaches 97% of programmed output voltage, then the gm error amplifier is enabled along with the hysteretic comparator. This point onwards, the voltage control loop (gm error amplifier) is fully in control and will regulate the output voltage. Soft-start time can be calculated approximately by adding the following four time frames: t1 = Cap_COMP × 0.18V/8.5µA t2 = 12 bit counter, approx 2ms t3 = Cap_COMP × 0.3V/8.5µA The current limiting resistor RCS is calculated by the following equation: RCS = RDS(ON) Q1 × LL 200µA I L = I LOAD = VOUT = VIN Q1 MOSFET N 0 VOUT L1 Inductor RCS CS LSD Q2 MOSFET N C1 COUT 200µA Figure 1. The MIC2168A Current Limiting Circuit April 2005 (VIN - VOUT ) VIN × F SWITCHING × L FSWITCHING = 1MHz 200µA is the internal sink current to program the MIC2168A current limit. The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended to add a 50% margin to the load current (ILOAD) in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect RCS resistor directly to the drain of the top MOSFET Q1, and the RSW resistor to the source of Q1 to accurately sense the MOSFETs RDS(ON). A 0.1µF capacitor in parallel with RCS should be connected to filter some of the switching noise. Internal VDD Supply The MIC2168A controller internally generates VDD for self biasing and to provide power to the gate drives. This VDD supply is generated through a low-dropout regulator and generates 5V from VIN supply greater than 5V. For supply voltage less than 5V, the VDD linear regulator is approximately 200mV in dropout. Therefore, it is recommended to short the VDD supply to the input supply through a 10Ω resistor for input supplies between 2.9V to 5V. MOSFET Gate Drive The MIC2168A high-side drive circuit is designed to switch an N-Channel MOSFET. The block diagram in Figure 2 shows a bootstrap circuit, consisting of D2 and CBST, supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the VSW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the MOSFET turns on, the voltage on the VSW pin increases to approximately VIN. Diode D2 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. When the low-side switch is turned back on, CBST is recharged through D2. The drive voltage is derived from the internal 5V VDD bias supply. The nominal low-side gate drive voltage is 5V and the nominal high-side gate drive voltage is approximately 4.5V due the voltage drop across D2. An approximate 20ns delay between the high- and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. MOSFET Selection The MIC2168A controller works from input voltages of 3V to 13.2V and has an internal 5V regulator to provide power to turn the external N-Channel power MOSFETs for high- and low-side switches. For applications where VIN < 5V, the internal Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 + t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms Current Limit The MIC2168A uses the RDS(ON) of the top power MOSFET to measure output current. Since it uses the drain to source resistance of the power MOSFET MOSFET, it is not very accurate. This scheme is adequate to protect the power supply and external components during a fault condition by cutting back the time the top MOSFET is on if the feedback voltage is greater than 0.67V. In case of a hard short when feedback voltage is less than 0.67V, the MIC2168A discharges the COMP capacitor to 0.65V, resets the digital counter and automatically shuts off the top gate drive, and the gm error amplifier and the –3% hysteretic comparators are completely disabled and the soft-start cycles restarts. This mode of operation is called the “hiccup mode” and its purpose is to protect the down stream load in case of a hard short. The circuit in Figure 1 illustrates the MIC2168A current limiting circuit. HSD 1 2(Inductor Ripple Current) where: Inductor Ripple Current = V Cap_COMP t4 - OUT x 0.5 x 8.5µA V IN C2 CIN Equation (1) 7 M9999-062205 MIC2168A Micrel, Inc. VDD regulator operates in dropout mode, and it is necessary that the power MOSFETs used are low threshold and are in full conduction mode for VGS of 2.5V. For applications when VIN > 5V; logic-level MOSFETs, whose operation is specified at VGS = 4.5V must be used. It is important to note the on-resistance of a MOSFET increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25°C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current-sense (CS) resistor. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2168A gate drive circuit. At 1MHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2168A. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the highside MOSFET is: IG[high - side](avg) = Q G × fS of the conduction losses during the on-time (PCONDUCTION) and the switching losses that occur during the period of time when the MOSFETs turn on and off (PAC). PSW = P CONDUCTION + P AC where: PCONDUCTION ( VO D = duty cycle V IN Making the assumption the turn-on and turn-off transition times are equal; the transition times can be approximated by: C × V GS + C OSS × VIN tT = ISS IG where: CISS and COSS are measured at VDS = 0 IG = gate-drive current (1A for the MIC2168A) The total high-side MOSFET switching loss is: P AC = (V (VIN +VD ) × IPK × tT × fS where: tT = switching transition time (typically 20ns to 50ns) VD = freewheeling diode drop, typically 0.5V fS it the switching frequency, nominally 1MHz The low-side MOSFET switching losses are negligible and can be ignored for these calculations. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below. ) A convenient figure of merit for switching MOSFETs is the on resistance times the total gate charge RDS(ON) × QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2168A. Parameters that are important to MOSFET switch selection are: • Voltage rating • On-resistance • Total gate charge The voltage ratings for the top and bottom MOSFET are essentially equal to the input voltage. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitics. The power dissipated in the switching transistor is the sum M9999-062205 × R SW RSW = on-resistance of the MOSFET switch Since the current from the gate drive comes from the input voltage, the power dissipated in the MIC2168A due to gate drive is: = VIN IG[high -side](avg) - side](avg) + IG[low -side](avg) - side](avg) 2 P AC = P AC(off) + P AC(on) where: IG[high-side](avg) = average high-side MOSFET gate current. QG = total gate charge for the high-side MOSFET taken from manufacturer’s data sheet for VGS = 5V. The low-side MOSFET is turned on and off at VDS = 0 because the freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge. For the low-side MOSFET: IG[low -side](avg) side](avg) = C ISS × V GS × fS PGATEDRIVE = ISW(rms) L= VOUT × (V (V IN (max max ) − VOUT ) VIN (max max ) × fS × 0.2 × IOUT (max ) where: fS = switching frequency, 1MHz 0.2 = ratio of AC ripple current to DC output current VIN(max) = maximum input voltage The peak-to-peak inductor current (AC ripple current) is: 8 April 2005 MIC2168A IPP = Micrel, Inc. The output capacitor ESR also affects the overall voltage feedback loop from stability point of view. See “Feedback Loop Compensation” section for more information. The maximum value of ESR is calculated: ∆V OUT R ESR ≤ IPP VOUT × (V (V IN (max max ) − VOUT ) VIN (max max ) × fS × L The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor ripple current. IPK = IOUT (max ) + 0.5 × IPP The RMS inductor current is used to calculate the I2 × R losses in the inductor. IINDUCTOR(rms) 1 IP = IOUT (max max ) × 1 + 3 IOUT (max max ) where: VOUT = peak-to-peak output voltage ripple IPP = peak-to-peak inductor ripple current The total output ripple is a combination of the ESR output capacitance. The total ripple is calculated below: 2 Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2168A requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below: PINDUCTOR Cu = IINDUCTOR(rms) 2 × R WINDING 2 ∆V OUT = ( ) 2 COUT = output capacitance value fS = switching frequency The voltage rating of capacitor should be twice the voltage for a tantalum and 20% greater for an aluminum electrolytic. The output capacitor RMS current is calculated below: I IC = PP OUT(rms) 12 The power dissipated in the output capacitor is: P DISS(C × R ESR(C ) )= I C OUT OUT(rms) 2 OUT Input Capacitor Selection The input capacitor should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. Tantalum input capacitor voltage rating should be at least 2 times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: ∆V VIN = IINDUCTOR(peak) × R ESR(C ) ) where: THOT = temperature of the wire under operating load T20°C = ambient temperature RWINDING(20°C) is room temperature winding resistance (usually specified by the manufacturer) Output Capacitor Selection The output capacitor values are usually determined capacitors ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors selecting the output capacitor. Recommended capacitors tantalum, low-ESR aluminum electrolytics, and POSCAPS. The output capacitor’s ESR is usually the main cause of output ripple. April 2005 ( where: D = duty cycle The resistance of the copper wire, RWINDING, increases with temperature. The value of the winding resistance used should be at the operating temperature. R WINDING(hot) = R WINDING(20 °C) × 1 + 0.0042 0.0042 × (T HOT − T 20 °C ) I × (1 (1 − D D) PP + IPP × R ESR C OUT × fS IN The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor ripple current is low: 9 M9999-062205 MIC2168A Micrel, Inc. VF = forward voltage at the peak diode current The external Schottky diode, D1, is not necessary for circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending on the circuit components and operating conditions, an external Schottky diode will give a 1/2% to 1% improvement in efficiency. Feedback Loop Compensation The MIC2168A controller comes with an internal transconductance error amplifier used for compensating the voltage feedback loop by placing a capacitor (C1) in series with a resistor (R1) and another capacitor C2 in parallel from the COMP pin to ground. See “Functional Block Diagram.” Power Stage The power stage of a voltage mode controller has an inductor, L1, with its winding resistance (DCR) connected to the output capacitor, COUT, with its electrical series resistance (ESR) as shown in Figure 3. The transfer function G(s), for such a system is: IC IN (rms) ≈ IOUT (max max ) × D × (1 − D) The power dissipated in the input capacitor is: PDISS(C ) = IC (rms) 2 × R ESR(C ) IN IN IN Voltage Setting Components The MIC2168A requires two resistors to set the output voltage as shown in Figure 2. R1 Error Amp FB 7 R2 VREF 0.8V MIC2168 [adj.] Figure 2. Voltage-Divider Configuration Where: VREF for the MIC2168A is typically 0.8V The output voltage is determined by the equation: R1 V O = VREF × 1 + R2 A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: V ×R R1 R2 = REF V O − VREF L External Schottky Diode An external freewheeling diode is used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 15ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current. VO ESR COUT Figure 3. The Output LC Filter in a Voltage Mode Buck Converter (1 + ESR × s × C) G(S) - 2 DCR × s × C + s × L × C + 1 + ESR × s × C The reverse voltage requirement of the diode is: VDIODE(rrm) = VIN Plotting this transfer function with the following assumed values (L=2 µH, DCR=0.009Ω, COUT=1000µF, ESR=0.050Ω) gives lot of insight as to why one needs to compensate the loop by adding resistor and capacitors on the COMP pin. Figures 4 and 5 show the gain curve and phase curve for the above transfer function. The power dissipated by the Schottky diode is: PDIODE = ID(avg) × VF where: M9999-062205 DCR 10 April 2005 MIC2168A 30 Micrel, Inc. 30 7.5 GAIN 0 0 50 P 15 100 37.5 150 60 60 100 100 180 3 1.10 5 4 1 .10 f 6 1 .10 1 .10 1000000 100 100 P 50 100 150 3 1.10 100 100 4 1 .10 f 5 1 .10 6 1 .10 1000000 Figure 5. Phase Curve for G(s) It can be seen from the transfer function G(s) and the gain curve that the output inductor and capacitor create a two pole system with a break frequency at: fC = 1 2 × π √ L × COUT Therefore, fLC = 3.6kHz By looking at the phase curve, it can be seen that the output capacitor ESR (0.050Ω) cancels one of the two poles (LCOUT) system by introducing a zero at: fZERO = Error Amplifier (z) - gm × 1+ R1 × S × C1 C1 × C2 × S s × (C1 + C2)1 + R1 × C1 + C2 The above equation can be simplified by assuming C2<<C1, 1+ R1 × S × C1 Error Amplifier (z) - gm × s × (C1)(1+ R1 × C2 × S) 1 2 × π × ESR × C OUT Therefore, FZERO = 6.36kHz. From the point of view of compensating the voltage loop, it is recommended to use higher ESR output capacitors since they provide a 90° phase gain in the power path. For comparison purposes, Figure 6, shows the same phase curve with an ESR value of 0.002Ω. April 2005 6 1 .10 1000000 5 1 .10 It can be seen from Figure 5 that at 50kHz, the phase is approximately –90° versus Figure 6 where the number is –150°. This means that the transconductance error amplifier has to provide a phase boost of about 45° to achieve a closed loop phase margin of 45° at a crossover frequency of 50kHz for Figure 4, versus 105° for Figure 6. The simple RC and C2 compensation scheme allows a maximum error amplifier phase boost of about 90°. Therefore, it is easier to stabilize the MIC2168A voltage control loop by using high ESR value output capacitors. gm Error Amplifier It is undesirable to have high error amplifier gain at high frequencies because high frequency noise spikes would be picked up and transmitted at large amplitude to the output, thus, gain should be permitted to fall off at high frequencies. At low frequency, it is desired to have high open-loop gain to attenuate the power line ripple. Thus, the error amplifier gain should be allowed to increase rapidly at low frequencies. The transfer function with R1, C1, and C2 for the internal gm error amplifier can be approximated by the following equation: 0 180 4 1 .10 f Figure 6. The Phase Curve with ESR = 0.002Ω Figure 4. The Gain Curve for G(s) 0 3 1.10 From the above transfer function, one can see that R1 and C1 introduce a zero and R1 and C2 a pole at the following frequencies: Fzero= 1/2 π × R1 × C1 Fpole = 1/2 π × C2 × R1 Fpole@origin = 1/2 π × C1 Figures 7 and 8 show the gain and phase curves for the above transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF, and gm = .005Ω–1. It can be seen that at 50kHz, the error amplifier exhibits approximately 45° of phase margin. 11 M9999-062205 MIC2168A Micrel, Inc. 100 71.607 OPEN LOOP GAIN MARGIN 60 60 40 E 20 .001 3 1 .10 1000 4 1 .10 5 6 1 .10 f 1 .10 0 42.933 50 100 100 7 1 .10 10000000 3 1.10 OPEN LOOP PHASE MARGIN 220 240 E 3 1.10 4 f 1 .10 5 1 .10 1 .10 1000000 300 350 360 260 100 6 5 1 .10 250 269.097 200 215.856 270 4 1 .10 f Figure 9. Open-Loop Gain Margin Figure 7. Error Amplifier Gain Curve 10 10 50 10 10 6 1 .10 1000000 100 3 1.10 4 f 1 .10 5 1 .10 6 1 .10 1000000 Figure 10. Open-Loop Phase Margin Figure 8. Error Amplifier Phase Curve Total Open-Loop Response The open-loop response for the MIC2168A controller is easily obtained by adding the power path and the error amplifier gains together, since they already are in Log scale. It is desirable to have the gain curve intersect zero dB at tens of kilohertz, this is commonly called crossover frequency; the phase margin at crossover frequency should be at least 45°. Phase margins of 30° or less cause the power supply to have substantial ringing when subjected to transients, and have little tolerance for component or environmental variations. Figures 9 and 10 show the open-loop gain and phase margin. It can be seen from Figure 9 that the gain curve intersects the 0dB at approximately 50kHz, and from Figure 10 that at 50kHz, the phase shows approximately 50° of margin. M9999-062205 12 April 2005 MIC2168A Micrel, Inc. Design Example Layout and Checklist: 1. Connect the current limiting (CS) resistor directly to the drain of top MOSFET Q1. 2. Connect the VSW pin directly to the source of top MOSFET Q1 thru a 4Ω to 10Ω resistor. The purpose of this resistor is to filter the switch node. 3. The feedback resistors R1 and R2 should be placed close to the FB pin. The top side of R1 should connect directly to the output node. Run this trace away from the switch node (junction of Q1, Q2, and L1). The bottom side of R1 should connect to the GND pin on the MIC2168A. 4. The compensation resistor and capacitors should be placed right next to the COMP/EN pin and the other side should connect directly to the GND pin on the MIC2168A rather than going to the plane. 5. The input bulk capacitors should be placed close to the drain of the top MOSFET. 6. The 1µF ceramic capacitor should be placed right on the VIN pin of the MIC2168A. 7. The 4.7µF to 10µF ceramic capacitor should be placed right on the VDD pin. 8. The source of the bottom MOSFET should connect directly to the input capacitor GND with a thick trace. The output capacitor and the input capacitor should connect directly to the GND plane. 9. Place a 0.1µF ceramic capacitor in parallel with the CS resistor to filter any switching noise. April 2005 13 M9999-062205 MIC2168A Micrel, Inc. Package Information Rev.. 00 10-Pin MSOP (MM) MICREL INC. TEL 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA + 1 (408) 944-0800 FAX + 1 (408) 474-1000 WEB http://www.micrel.com This information furnished by Micrel in this data sheet is believed to be accurate and reliable. However no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2005 Micrel Incorporated M9999-062205 14 April 2005