AN-60-010 - Mini Circuits

(e.g., ERA SERIES)
The Mini-Circuits family of microwave monolithic integrated circuit (MMIC)
Darlington amplifiers offers the RF designer multi-stage performance in packages that
look like a discrete transistor. Included in this family are the model prefixes to which
the biasing considerations in this application note apply: ERA, Gali, LEE, MAR, MAV,
RAM, and VAM. These amplifiers’ advantages of wide bandwidth, impedance match,
and a choice of gain and output power levels result from their being monolithic circuits,
most of which contain InGaP HBT (indium-gallium-phosphide heterojunction bipolar
transistors). In addition, most of these amplifiers incorporate a patented circuit (US
Patent No. 6943629) that provides protection against damage due to power supply turnon transients.
The internal circuit configuration is a Darlington pair, embedded in a resistor network
as shown in the schematic diagram, Figure 1. Applying DC power to operate this kind
of amplifier is simpler than biasing a transistor. Like a discrete bipolar transistor, this
circuit is current-controlled rather than voltage-controlled. This means that for a range
of current around a recommended value, the device voltage varies much less than in
proportion to current. A constant-current DC source would be ideal for providing a
stable operating point. By contrast, with most of these models the use of a constantvoltage DC source would cause the current to vary widely with small changes in supply
voltage, temperature change, and device-to-device variations. Stable operating point
needs an external series resistor between the amplifier and a DC voltage supply to
approximate a constant-current source. This application note gives the user step-by-step
guidance in choosing external bias circuit components to obtain optimum performance.
A Darlington amplifier is a 2-port device: RF input, and combined RF output and bias
input. It is housed in a 4-lead package including 2 ground leads; connecting both of
them to external ground will minimize common path impedance for best RF
performance. Internal resistors in Figure 1
determine the DC operating point of the
transistors and provide feedback to set RF
gain, bandwidth, and input and output
impedances to optimum values.
Figure 1 Schematic Diagram
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Bias Circuit
A practical biasing configuration is shown in Figure 2. Bias current is delivered from a
voltage supply Vcc through the resistor Rbias and the RF choke (inductor), shown as
RFC in the figure. The resistor reduces the effect of device voltage (Vd) variation on the
bias current by approximating a current source.
Blocking capacitors are needed at the input and output ports. They should be of a type
having low ESR (effective series resistance), and should have reactance low enough not
to affect insertion loss or VSWR adversely at low frequency. The blocking capacitors
must be free of parasitic resonance up to the highest operating frequency. Use of a
bypass capacitor at the VCC end of Rbias is advised to prevent stray coupling from or to
other signal processing components via the DC supply line.
Figure 2
Typical biasing Configuration for Darlington Amplifiers
In this circuit, DC blocking capacitors are added at the input port (pin
number 1 on the packaged amplifier) and at the output port (pin 3).
Bias current is given by the equation: Ibias = (Vcc ! Vd) ) Rbias.
The individual data sheet for each of the amplifiers lists the values of the bias resistor
needed with several values of supply voltage. These values take into account the
variation of device voltage, both lot-to-lot and with temperature (-45° to 85EC). Also,
they are chosen from the readily available “1%” resistor values. The greater the
difference between the supply and device voltage, the easier it is to maintain constant
operating conditions; this will be discussed. A further consideration affecting
component choice is the DC power dissipated by the bias resistor, which increases with
increasing supply voltage.
The bias current values in the data sheets are the recommended values. Greater current
raises junction temperature, reducing MTTF (mean time to failure). The effect of bias
current on MTTF is discussed in more detail later.
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Advantage of an RF Choke
The advisability of using an RF choke in series with Rbias is evident from the following
analysis. Figure 3 shows an equivalent circuit of the output of a Darlington amplifier as
a current source in parallel with an internal 50-ohm source resistance, loaded by both
Rbias and an external 50-ohm load. The current in the 50-ohm load is:
I50Ω = Isource H Rbias ) (2 Rbias + 50)
and the loss in power gain relative to not having the output loaded by Rbias is:
Gloss = 20 log [(2 Rbias + 50) ) 2 Rbias] dB
Figure 3
Effect of Bias Resistor on the Output, Without an RF Choke
Suppose, for example, that Model ERA-4SM+ is used with a 12-volt supply without a
choke. From the above expression, the effect of the 115-ohm bias resistor (from the
data sheet) is found to be a 1.7 dB reduction in the gain of the amplifier.
An RF choke should be chosen such that its reactance is at least 500 ohms (10 times
the load impedance) at the lowest operating frequency. It must also be free of parasitic
(series) resonance up to the highest operating frequency.
Super Wide-band RF Choke
The circuit designer might consider using a commercially available inductor as the RF
choke in the bias circuit of Figure 2. The low end of the useful frequency range is
controlled by the value of the inductance; the higher the value, the lower the frequency.
The high end of the frequency range is determined by the series resonant frequency of
the inductor; it tends to decrease as the value of the inductance increases. The
frequency band of the overall amplifier circuit is often limited by the inductor rather
than the MMIC amplifier itself, which have performance up to 8 GHz. Besides,
inductors are not clearly specified for RF choke application, and design change by the
inductor manufacturer will have an unknown effect on the circuit. This complicates the
circuit designer’s job.
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Mini-Circuits solves the problem by offering super wide-band RF chokes, which enable
circuit designers to utilize easily the full capability of the MMIC Darlington amplifiers.
An RF choke is a 2-terminal device. In a 0.31 by 0.22-inch surface mount package,
models ADCH-80+ and ADCH-80A+ cover 50 to 10000 MHz with different pin-outs
to accommodate different PC board layouts. In a smaller 0.15 by 0.15-inch package,
TCCH-80+ covers 50 to 8200 MHz.
The equivalent inductance of the Super Wide-band RF choke ADCH-80+ is one
microhenry at 100 mA. For comparison, a typical commercially available onemicrohenry inductor has a series resonant frequency as low as 90 MHz, which is much
lower than this RF choke. Figure 4 plots the insertion loss and Figure 5 the VSWR at
various currents up to 100 mA, for the RF choke placed in shunt across a 50-ohm
transmission line. Note that the insertion loss and VSWR change very little with change
in current in the specified frequency band.
Figure 4 Insertion Loss of the RF Choke ADCH-80+
Figure 5 VSWR of the RF Choke ADCH-80+
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Higher Bias Resistor Values Reduce Variation
Increasing the supply voltage allows a higher value of Rbias to be used, in accordance
with the pertinent data sheet, and that reduces the variation in bias current. The benefit
is that RF performance, especially the 1-dB compression point, is made more constant.
The following quantifies this effect.
The device voltage Vd is a function of both Ibias and temperature T. Device voltage
increases with bias current, and the variation can be expressed as a rate of change:
Δ Vd / Δ Ibias,
in mV per mA. Device voltage in most models decreases with increasing temperature,
Δ Vd / ΔT
in millivolts per degree C. Over the useful range of bias current and temperature, these
rates of change can be assumed constant. Typical values of these variation coefficients
for the ERA series are listed in Table 1 as examples. This information is for
applications guidance only; for the latest specifications refer to published data sheets
where you can find the values of the coefficients for all Mini-Circuits MMIC
Darlington amplifiers.
Table 1
Model No.
Device Voltage Variation with Current and Temperature
Nominal Bias
Current, mA
Typ. Device Voltage
Variation with Current,
ΔVd/ΔIbias (mV/mA)
Typ. Device Voltage Variation
with Temperature, ΔVd/ΔT
- 2.0
The combined effect of the two device-voltage coefficients can be expressed as
variation of bias current with temperature, derived as follows:
Ibias = [Vcc ! Vd(Ibias, T)] ) Rbias,
where Vd(Ibias, T) = V0 + (Δ Vd / Δ Ibias) @ Ibias + (Δ Vd / ΔT) @ (T ! T0)
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The AΔ@ ratios are the respective device-voltage vs. current and device-voltage vs.
temperature coefficients. V0 is the Vd-axis intercept of the linear extension of the Vd vs.
Ibias curve, at room temperature T0. Substituting for Vd from (5) into (4) and solving for
Ibias = [Vcc!V0 ! (Δ Vd / ΔT) @ (T ! T0)]
) [Rbias + (Δ Vd / Δ Ibias)]
Differentiating (6) with respect to T gives the desired result:
Δ Ibias / ΔT = !(Δ Vd / ΔT) ) [Rbias + (Δ Vd / Δ Ibias)]
To find the change in bias current that will typically occur with a given change in
temperature ΔT, substitute values for the device-voltage coefficients from Table 1 or
the data sheet for the model, and the value of bias resistor being considered, into the
right-hand side of equation (7). Multiplying the result by ΔT yields the change in bias
current Δ Ibias over the temperature range.
To illustrate how the choice of supply voltage affects the variation of bias current with
temperature, let us compare two values of Vcc for Model ERA-1SM+.
For the first example, Vcc = 5 volts, find Rbias by inverting equation (1):
Rbias = (Vcc ! Vd) ) Ibias = (5 – 3.4) = 40 ohms.
Substituting in equation (7):
Δ Ibias / ΔT = !(!2.0) ) (40 + 9.4) = .040 mA/EC.
Over an operating temperature range of !45 to 85 degrees C, the total variation in
current for this example will be 5.2 mA, which is 13% of the recommended value of
current. The consequence is about 1.4 dB variation in output power at 1-dB
For the second example, let Vcc = 12 volts and Rbias = 215 ohms. A similar calculation
yields 1.16 mA total variation in current, which causes only 0.4 dB variation in output
Additional Sources of Variation
The variation of device voltage as discussed above pertains to an individual device. The
specifications include the limits of device voltage that the various models must meet.
Actual expected unit-to-unit variation is much less than indicated by those limits,
because of the high degree of process control used during manufacture. Therefore, a
user can be confident that results obtained when prototyping products incorporating
Mini-Circuits MMIC Darlington amplifiers that performance will be highly repeatable.
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Besides the variation of bias current due to the characteristics of the amplifier itself,
there are two additional causes that the user might need to consider:
• Available nominal values, tolerance and temperature coefficient of the bias resistor.
• Voltage setting error and regulation of the power supply.
Resistors readily available are generally the “1% values”. The nominal values, having
increments of 2%, could thus differ from the data-sheet values by as much as ±1%. If a
resistor having temperature coefficient of 200 ppm/EC is used, it could vary "1.3%
relative to the room-temperature value over the !45 to 85EC range. The resistance, and
correspondingly the bias current, might therefore be as much as 2.3% different from the
desired value in data sheet due to these contributions.
Good DC power supplies typically have about 10 mV combined line and load
regulation, and that tends not to be an important factor. However, the tolerance of a
fixed-voltage supply, or setting accuracy of a field adjustable one might be ±1%. As an
example, consider the ERA-51SM+ being used with a 9-volt supply. Since the nominal
device voltage is 4.5 V (half the supply voltage), a 1%, error in supply voltage would
change the current by 0.5%.
The combination of the resistor and power supply variations in the above example
would cause 2.3% + 0.5% = 2.8% error in current, which is 1.8 mA for a nominal value
of 65 mA. Besides, the temperature effect on current in the ERA-51SM+ device itself,
using equation (7), is:
Δ Ibias / ΔT = !(!3.2) ) (69 + 6.7) = 0.042 mA/EC.
Relative to the room temperature value, current would change 0.042 mA/EC × 65°C =
2.7 mA with temperature. The overall combined variation, including the bias resistor
and power supply effects, is "4.5 mA for this example. A user working within an
MTTF or current consumption budget who needs to ensure that 65 mA is not exceeded
may therefore choose to design for a nominal value of 60 mA in this example.
Minimizing Power Dissipation
In addition to bias current stability, stability of power dissipation of the MMIC
Darlington amplifier is favored by using a high Vcc value. This is because of the
negative temperature coefficient of device voltage Vd. In particular, if Vcc is at least 2
times Vd, then PD, the power dissipation decreases with increasing temperature, as
shown by the following analysis:
PD = Vd @ Ibias = Vd (Vcc ! Vd) ) Rbias
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Taking the derivative of PD with respect to Vd and setting it equal to zero, we find that
the maximum value of PD occurs when Vd = Vcc ) 2. This is illustrated in Figure 6.
Figure 6 Variation of Power Dissipation with Temperature, for Device
Voltage Above, and Below, Half the Supply Voltage Value
Let us see what happens to power dissipation in the case of ERA-4SM+ between 25
and 85 degrees C, for two values of supply voltage: one less than, and another greater
than, twice the value of Vd. We will use 65 mA and 4.6 V as the nominal (room
temperature) values of Ibias and Vd.
To find the change in PD, the power dissipation of the amplifier, we must use the
equations for Vd and Ibias with which we started in order to account for the
interdependence of these quantities as well as temperature:
Vd(Ibias, T) = V0 + (Δ Vd / Δ Ibias) @ Ibias + (Δ Vd / ΔT) @ (T ! T0)
Ibias = [Vcc ! V0 ! (Δ Vd / ΔT) @ (T ! T0)]
) [Rbias + (Δ Vd / Δ Ibias)]
as before.
Substituting for Ibias from equation (6) into the Vd equation (5), and solving for Vd:
Vd = [Rbias @ V0 + (Δ Vd / Δ Ibias) @ Vcc + Rbias @ (Δ Vd / ΔT) @ (T ! T0)]
[Rbias + (Δ Vd / Δ Ibias)]
PD can now obtained as the product of right-hand sides of equations (6) and (9). The
constant V0 is found by subtracting [Ibias H (Δ Vd /Δ Ibias)] from Vd, using the
room-temperature values of Ibias and Vd.
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For Vcc = 5.0 V, Rbias = 7.7 ohms. Using the value !2.9mV/EC for the Vd temperature
coefficient from Table 1 and the above equations, PD increases from 0.293 watt at
25EC to 0.331 watt at 85EC. Considering the thermal resistance of ERA-4SM+
(junction-to-case) of 196EC/W, this increase in power dissipation results in 7EC higher
junction temperature than if dissipation were constant. The consequence in reliability is
a factor of 2 reduction in MTTF.
Now, try Vcc = 12 V, for which Rbias = 115 ohms. For this case, PD decreases from
0.293watt at 25EC to 0.288 watt at 85EC.
Temperature Compensated Bias Network
An alternative method of biasing that allows use of lower supply voltage while
maintaining bias current stability and reducing power dissipation in the bias resistor is
to use a temperature compensating bias network in place of the single resistor Rbias. The
network consists of a linear positive-temperature-coefficient chip thermistor in parallel
with a regular chip resistor, and should be designed so that its resistance increases with
temperature just enough to make up for the decrease in device voltage, causing the bias
current to remain constant.
Commercially available chip thermistors have a very high TCR (temperature coefficient
of resistance), typically +4500 ppm/°C for the range 51 - 510 ohms. The temperature
coefficient needed for Rbias is much less than this, and thus achievable by using the 2components (resistor and thermistor) in parallel.
We now derive the values of the network components. Let R be the resistance of the
regular resistor, and Rt the 25°C resistance of the thermistor.
Let kb be the fractional increase in Rbias needed at the maximum operating temperature
relative to 25°C: (hot resistance ! 25°C resistance) ) 25°C resistance.
Let kt be the fractional increase in resistance of the thermistor,
(4500 H10!6) H (maximum operating temperature ! 25°C).
Because the resistor and thermistor are in parallel they must satisfy:
Rbias = [email protected] ) (R + Rt ) at 25°C,
and (1 + kb) Rbias = [email protected] (1 + kt) ) [R + Rt (1 + kt)]
at the maximum operating temperature.
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Solving (10) and (11) yields:
R = Rbias @ kt (1 + kb) ) ( kt !kb),
Rt = Rbias @ kt (1 + kb) ) [kb (1 + kt)]
Let us compare simple resistor biasing with the temperature compensating network for
biasing an ERA-6SM+ with a 7.0 V supply. Let Ibias = 70 mA and Vd = 4.9 V (at
25°C), and use the Rbias value 30.1 ohms from the ERA-6SM+ data sheet. If we use an
ordinary resistor for Rbias without a thermistor, the current will increase to 74 mA.
Now, we compute the network component values needed to make the current 70 mA at
85°C as well as at 25°C. At 85°C, Vd = 4.9 –(0.0031 V/ °C ) H (60°C) = 4.71V, and
Rbias should become 32.7 ohms. Thus, kb = (32.7 ! 30.1) ) 30.1 = 0.0864. Also, kt =
(4500 H10!6) H (85° ! 25°) = 0.27. Applying equations (12) and (13),
R = 30.1 H 0.27 (1 + 0.0864) ) (0.27− 0.0864) = 48.1 ohms
Rt = 30.1
H 0.27 (1 + 0.0864) ) [0.0864 (1 + 0.27)] = 80.5 ohms
If the thermistors are available only in A5% [email protected], sufficiently close compensation is
obtained by using Rt = 82 ohms and R = 47.5 ohms (for 30.1-ohm parallel equivalent).
For different supply voltages and device operating points different resistor and
thermistor values are needed, but the same concept and method can be used. The
benefit is the ability to keep the device current constant over temperature, thereby
avoiding increase in power dissipated in the amplifier and reduction in MTTF.
We have not mentioned the resistance-temperature coefficient of the ordinary resistor in
the bias network. The reason is that thick film chip resistors typically have a coefficient
of "100 ppm/EC. This is about 2% of the TCR of the thermistor, and does not influence
the results significantly. A word of caution is due regarding the thermistor, however. Its
temperature characteristic is controlled only at 25°C and at 75°C. The user should test
actual circuit operation at other temperatures of interest.
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