DS6207AB 02

®
RT6207A/B
5A, 18V, 650kHz, ACOTTM Synchronous Step-Down Converter
General Description
Features
The RT6207A/B is a high-performance 650kHz, 5A stepdown regulator with internal power switches and
synchronous rectifiers. It features quick transient response
using its Advanced Constant On-Time (ACOTTM) control
architecture that provides stable operation with small
ceramic output capacitors and without complicated
external compensation, among other benefits. The input
voltage range is from 4.5V to 18V and the output is
adjustable from 0.7V to 8V. The proprietary ACOTTM control
improves upon other fast response constant on-time
architectures, achieving nearly constant switching
frequency over line, load, and output voltage ranges. Since
there is no internal clock, response to transients is nearly
instantaneous and inductor current can ramp quickly to
maintain output regulation without large bulk output
capacitance. The RT6207A/B is stable with and optimized
for ceramic output capacitors. With internal 60mΩ switches
and 22mΩ synchronous rectifiers, the RT6207A/B displays
excellent efficiency and good behavior across a range of
applications, especially for low output voltages and low
duty cycles. Cycle-by-cycle current limit provides
protection against shorted outputs, input under-voltage
lock-out, externally-adjustable soft-start, output under- and
over-voltage protection, and thermal shutdown provide safe
and smooth operation in all operating conditions. The
RT6207A/B is available in the UQFN-13JL 2x3 (FC)
package, with exposed thermal pad.
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Fast Transient Response
Advanced Constant On-Time (ACOTTM) Control
4.5V to 18V Input Voltage Range
Adjustable Output Voltage from 0.7V to 8V
5A Output Current
60mΩ
Ω Internal High-Side N-MOSFET and 22mΩ
Ω
Internal Low-Side N-MOSFET
Steady 650kHz Switching Frequency
Up to 95% Efficiency
Optimized for All Ceramic Capacitors
Externally-Adjustable, Pre-Biased Compatible SoftStart
Cycle-by-Cycle Current Limit
Input Under-Voltage Lockout
Output Over- and Under-Voltage Protection
Power Good Output
Externally-Adjustable, Pre-Biased Compatible SoftStart
Thermal Shutdown
Applications
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Industrial and Commercial Low Power Systems
Computer Peripherals
LCD Monitors and TVs
Green Electronics/Appliances
Point of Load Regulation for High-Performance DSPs,
FPGAs, and ASICs
Simplified Application Circuit
RT6207A/B
SW
VIN
VIN
VPVCC
Enable
CSS
BOOT
December 2015
CFF
R1
COUT
PGOOD
EN
FB
SS
PVCC
GND
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
DS6207A/B-02
VOUT
CBOOT
CIN
RPG
L
VPVCC
CPVCC
R2
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RT6207A/B
Ordering Information
Pin Configurations
BOOT
12
11
1
10
VIN
EN
2
9
VIN
FB
3
8
PGOOD
NC
4
7
NC
A : PSM
B : PWM
Note :
5
6
PVCC
GND
UVP Option
H : Hiccup Mode
L : Latched-Off Mode
UQFN-13JL 2x3 (FC)
Richtek products are :

13
SS
Lead Plating System
G : Green (Halogen Free and Pb Free)
SW
Package Type
QUF : UQFN-13JL 2x3 (U-Type) (FC)
GND
(TOP VIEW)
RT6207A/B
RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.

Suitable for use in SnPb or Pb-free soldering processes.
Marking Information
RT6207AHGQUF
RT6207BHGQUF
0R : Product Code
W : Date Code
0RW
0P : Product Code
0PW
RT6207ALGQUF
RT6207BLGQUF
0Q : Product Code
W : Date Code
0QW
W : Date Code
0N : Product Code
0NW
W : Date Code
Functional Pin Description
Pin No.
1, 13
Pin Name
Pin Function
GND
Ground.
2
EN
Enable Control Input.
3
FB
Converter Feedback Input. Connect to output voltage feedback resistor divider.
4, 7
NC
No Internal Connection.
5
SS
Soft-Start Control. A external capacitor should be connected to GND.
6
PVCC
5V Power Supply Output. A capacitor (typical 1F) should be connected to GND.
8
PGOOD
Open Drain Power Good Output.
VIN
Power Input and Connected to High-Side MOSFET Drain.
11
BOOT
Bootstrap Supply for High-Side Gate Driver. This capacitor is needed to drive the
power switch's gate above the supply voltage. It is connected between SW and
BOOT pins to form a floating supply across the power switch driver. A 0.1F
capacitor is recommended for use.
12
SW
Switch Output.
9, 10
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DS6207A/B-02
December 2015
RT6207A/B
Function Block Diagram
BOOT
PVCC
VIN
PVCC
Reg
Min.
Off
VIBIAS
PVCC
VIN
VREF
UGATE
Control
OC
Driver
SW
LGATE
UV & OV
PVCC
SW
6µA
Ripple
Gen.
SS
FB
VIN
SW
GND
GND SW
+
Comparator
On-Time
Comparator
0.9 VREF
FB
PGOOD
+
-
EN
EN
Detailed Description
The RT6207A/B is a high-performance 650kHz 5A stepdown regulators with internal power switches and
synchronous rectifiers. It features an Advanced Constant
On-Time (ACOTTM) control architecture that provides
stable operation with ceramic output capacitors without
complicated external compensation, among other benefits.
The ACOTTM control mode also provide fast transient
response, especially for low output voltages and low duty
cycles.
The input voltage range is from 4.5V to 18V and the output
is adjustable from 0.7V to 8V. The proprietary ACOTTM
control scheme improves upon other constant on-time
architectures, achieving nearly constant switching
frequency over line, load, and output voltage ranges. The
RT6207A/B are optimized for ceramic output capacitors.
Since there is no internal clock, response to transients is
nearly instantaneous and inductor current can ramp quickly
to maintain output regulation without large bulk output
capacitance.
Constant On-Time (COT) Control
The heart of any COT architecture is the on-time one shot.
Each on-time is a pre-determined “fixed” period that is
triggered by a feedback comparator. This robust
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
DS6207A/B-02
December 2015
arrangement has high noise immunity and is ideal for low
duty cycle applications. After the on-time one-shot period,
there is a minimum off-time period before any further
regulation decisions can be considered. This arrangement
avoids the need to make any decisions during the noisy
time periods just after switching events, when the
switching node (SW) rises or falls. Because there is no
fixed clock, the high-side switch can turn on almost
immediately after load transients and further switching
pulses can ramp the inductor current higher to meet load
requirements with minimal delays.
Traditional current mode or voltage mode control schemes
typically must monitor the feedback voltage, current
signals (also for current limit), and internal ramps and
compensation signals, to determine when to turn off the
high-side switch and turn on the synchronous rectifier.
Weighing these small signals in a switching environment
is difficult to do just after switching large currents, making
those architectures problematic at low duty cycles and in
less than ideal board layouts.
Because no switching decisions are made during noisy
time periods, COT architectures are preferable in low duty
cycle and noisy applications. However, traditional COT
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RT6207A/B
control schemes suffer from some disadvantages that
preclude their use in many cases. Many applications require
a known switching frequency range to avoid interference
with other sensitive circuitry. True constant on-time control,
where the on-time is actually fixed, exhibits variable
switching frequency. In a step-down converter, the duty
factor is proportional to the output voltage and inversely
proportional to the input voltage. Therefore, if the on-time
is fixed, the off-time (and therefore the frequency) must
change in response to changes in input or output voltage.
Modern pseudo-fixed frequency COT architectures greatly
improve COT by making the one-shot on-time proportional
to VOUT and inversely proportional to VIN. In this way, an
on-time is chosen as approximately what it would be for
an ideal fixed-frequency PWM in similar input/output
voltage conditions. The result is a big improvement but
the switching frequency still varies considerably over line
and load due to losses in the switches and inductor and
other parasitic effects.
Another problem with many COT architectures is their
dependence on adequate ESR in the output capacitor,
making it difficult to use highly-desirable, small, low-cost,
but low-ESR ceramic capacitors. Most COT architectures
use AC current information from the output capacitor,
generated by the inductor current passing through the
ESR, to function in a way like a current mode control
system. With ceramic capacitors the inductor current
information is too small to keep the control loop stable,
like a current mode system with no current information.
ACOTTM Control Architecture
Making the on-time proportional to VOUT and inversely
proportional to VIN is not sufficient to achieve good
constant-frequency behavior for several reasons. First,
voltage drops across the MOSFET switches and inductor
cause the effective input voltage to be less than the
measured input voltage and the effective output voltage to
be greater than the measured output voltage. As the load
changes, the switch voltage drops change causing a
switching frequency variation with load current. Also, at
light loads if the inductor current goes negative, the switch
dead-time between the synchronous rectifier turn-off and
the high-side switch turn-on allows the switching node to
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4
rise to the input voltage. This increases the effective on
time and causes the switching frequency to drop
noticeably.
One way to reduce these effects is to measure the actual
switching frequency and compare it to the desired range.
This has the added benefit eliminating the need to sense
the actual output voltage, potentially saving one pin
connection. ACOTTM uses this method, measuring the
actual switching frequency and modifying the on-time with
a feedback loop to keep the average switching frequency
in the desired range.
To achieve good stability with low-ESR ceramic capacitors,
ACOTTM uses a virtual inductor current ramp generated
inside the IC. This internal ramp signal replaces the ESR
ramp normally provided by the output capacitor's ESR.
The ramp signal and other internal compensations are
optimized for low-ESR ceramic output capacitors.
ACOTTM One-Shot Operation
The RT6207A/B control algorithm is simple to understand.
The feedback voltage, with the virtual inductor current ramp
added, is compared to the reference voltage. When the
combined signal is less than the reference and the ontime one-shot is triggered, as long as the minimum offtime one-shot is clear and the measured inductor current
(through the synchronous rectifier) is below the current
limit. The on-time one-shot turns on the high-side switch
and the inductor current ramps up linearly. After the on
time, the high-side switch is turned off and the synchronous
rectifier is turned on and the inductor current ramps down
linearly. At the same time, the minimum off-time one-shot
is triggered to prevent another immediate on-time during
the noisy switching time and allow the feedback voltage
and current sense signals to settle. The minimum off-time
is kept short (230ns typical) so that rapidly-repeated ontimes can raise the inductor current quickly when needed.
Discontinuous Operating Mode (RT6207A Only)
After soft-start, the RT6207A operates in fixed frequency
mode to minimize interference and noise problems. The
RT6207A uses variable-frequency discontinuous switching
at light loads to improve efficiency. During discontinuous
switching, the on-time is immediately increased to add
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DS6207A/B-02
December 2015
RT6207A/B
“hysteresis” to discourage the IC from switching back to
continuous switching unless the load increases
substantially.
The IC returns to continuous switching as soon as an ontime is generated before the inductor current reaches zero.
The on-time is reduced back to the length needed for
650kHz switching and encouraging the circuit to remain
in continuous conduction, preventing repetitive mode
transitions between continuous switching and
discontinuous switching.
Current Limit
The RT6207A/B current limit is a cycle-by-cycle “valley”
type, measuring the inductor current through the
synchronous rectifier during the off-time while the inductor
current ramps down. The current is determined by
measuring the voltage between source and drain of the
synchronous rectifier. If the inductor current exceeds the
current limit, the on-time one-shot is inhibited (Mask high
side signal) until the inductor current ramps down below
the current limit. Thus, only when the inductor current is
well below the current limit is another on time permitted.
This arrangement prevents the average output current from
greatly exceeding the guaranteed current limit value, as
typically occurs with other valley-type current limits. If
the output current exceeds the available inductor current
(controlled by the current limit mechanism), the output
voltage will drop. If it drops below the output under-voltage
protection level, the IC will stop switching (see next
section).
Output Over-Voltage Protection and Under-Voltage
Protection
If the output voltage VOUT rises above the regulation level
and lower 1.2 times regulation level, the high-side switch
naturally remains off and the synchronous rectifier turns
on. For RT6207B, if the output voltage remains high the
synchronous rectifier remains on until the inductor current
reaches the low side current limit. If the output voltage
still remains high, then IC's switches remain that the
synchronous rectifier turns on and high-side MOS keeps
off to operate at typical 500kHz switching protection, again
if inductor current reaches low side current, the
synchronous rectifier will turn off until next protection
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DS6207A/B-02
December 2015
clock. If the output voltage exceeds the OVP trip threshold
(1.25 times regulation level) for longer than 10μs (typical),
then IC's output Over-Voltage Protection (OVP) is
triggered. RT6207BL chip enters latch mode.
For RT6207A If the output voltage Vo rises above the
regulation level and lower 1.2 times regulation level, the
high-side switch naturally remains off and the synchronous
rectifier turns on until the inductor current reaches zero
current. If the output voltage remains high, then IC's
switches remain off. If the output voltage exceeds the OVP
trip threshold (1.25 times regulation level) for longer than
10μs (typical), the IC's OVP is triggered. RT6207AL chip
enters latch mode.
The RT6207A/B include output Under-Voltage Protection
(UVP). If the output voltage drops below the UVP trip
threshold for longer than 250μs (typical) then IC's UVP is
triggered. Chip into latch or hiccup mode. (see next
section).
Hiccup Mode
The RT6207AH/BH, use hiccup mode for UVP. When the
protection function is triggered, the IC will shut down for a
period of time and then attempt to recover automatically.
Hiccup mode allows the circuit to operate safely with low
input current and power dissipation, and then resume
normal operation as soon as UVP is removed. During
hiccup mode, the shutdown time is determined by the
capacitor at SS. A 2μA current source discharges VSS
from its starting voltage (normally VPVCC). The IC remains
shut down until VSS reaches 0.2V, about 10ms for a 3.9nF
capacitor. At that point the IC begins to charge the SS
capacitor at 6μA, and a normal start-up occurs. If the fault
remains, UVP protection will be enabled when VSS reaches
2.2V (typical). The IC will then shut down and discharge
the SS capacitor from the 2.2V level, taking about 4ms for
a 3.9nF SS capacitor.
Latch-Off Mode
The RT6207AL/BL, use latch-off mode OVP and UVP.
When the protection function is triggered the IC will shut
down in Latch-Off Mode. The IC stops switching, leaving
both switches open, and is latched off. To restart operation,
toggle EN or power the IC off and then on again.
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RT6207A/B
Shut-down, Start-Up and Enable (EN)
The enable input (EN) has a logic-low level of 0.4V. When
VEN is below this level the IC enters shutdown mode and
supply current drops to less than 10μA. When VEN exceeds
its logic-high level of 2V the IC is fully operational.
Between these 2 levels there are 2 thresholds (1.2V typical
and 1.4V typical). When VEN exceeds the lower threshold
the internal bias regulators begin to function and supply
current increases above the shutdown current level.
Switching operation begins when VEN exceeds the upper
threshold.
Input Under-Voltage Lock-Out
In addition to the enable function, the RT6207A/B feature
an Under-Voltage Lock-Out (UVLO) function that monitors
the internal linear regulator output (VIN). To prevent
operation without fully-enhanced internal MOSFET
switches, this function inhibits switching when VIN drops
below the UVLO-falling threshold. The IC resumes
switching when VIN exceeds the UVLO-rising threshold.
Soft-Start (SS)
The RT6207A/B soft-start uses an external pin (SS) to
clamp the output voltage and allow it to slowly rise. After
VEN is high and VIN exceeds its UVLO threshold, the IC
begins to source 6μA from the SS pin. An external capacitor
at SS is used to adjust the soft-start timing. Following
below equation to get the minimum capacitance range in
order to avoid UV occur.
T=
COUT  VOUT  0.6  1.2
I
 LIM  Load Current   0.8
CSS 
T  6μA
VREF
Do not leave SS unconnected. During start-up, while the
SS capacitor charges, the RT6207A/B operates in
discontinuous switching mode with very small pulses. This
prevents negative inductor currents and keeps the circuit
from sinking current. Therefore, the output voltage may
be pre-biased to some positive level before start-up. Once
the VSS ramp charges enough to raise the internal
reference above the feedback voltage, switching will begin
and the output voltage will smoothly rise from the pre-
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6
biased level to its regulated level. After VSS rises above
about 2.2V output over- and under-voltage protections are
enabled and the RT6207A/B begins continuous-switching
operation.
Internal Regulator (PVCC)
An internal linear regulator (PVCC) produces a 5V supply
from VIN. The 5V power supplies the internal control
circuit, such as internal gate drivers, PWM logic, reference,
analog circuitry, and other blocks. 1μF ceramic capacitor
for decoupling and stability is required.
PGOOD Comparator
PGOOD is an open-drain output controlled by a comparator
connected to the feedback signal. If FB exceeds 90% of
the internal reference or OVP event is cleared, , PGOOD
will be high impedance. Otherwise, the PGOOD output is
connected to GND.
External Bootstrap Capacitor (CBOOT)
Connect a 0.1μF low ESR ceramic capacitor between
BOOT and SW. This bootstrap capacitor provides the gate
driver supply voltage for the high-side N-channel MOSFET
switch.
Some of case, such like duty ratio is higher than 65%
application or input voltage is lower than 5.5V which are
recommended to add an external bootstrap diode between
an external 5V and BOOT pin for efficiency improvement
The bootstrap diode can be a low cost one such as IN4148
or BAT54. The external 5V can be a 5V fixed input from
system or a 5V output of the RT6207A/B. Note that the
external boot voltage must be lower than 5.5V
Over-Temperature Protection
The RT6207A/B includes an Over-Temperature Protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. The OTP will shut down switching
operation when the junction temperature exceeds 150°C.
Once the junction temperature cools down by
approximately 20°C the IC will resume normal operation
with a complete soft-start. For continuous operation,
provide adequate cooling so that the junction temperature
does not exceed 150°C.
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DS6207A/B-02
December 2015
RT6207A/B
Absolute Maximum Ratings
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(Note 1)
Supply Voltage, VIN -----------------------------------------------------------------------------------------------Switch Voltage, SW -----------------------------------------------------------------------------------------------Switch Voltage, <10ns --------------------------------------------------------------------------------------------BOOT Voltage -------------------------------------------------------------------------------------------------------EN to GND ------------------------------------------------------------------------------------------------------------Other Pins ------------------------------------------------------------------------------------------------------------Power Dissipation, PD @ TA = 25°C
UQFN-13JL 2x3 (FC) ----------------------------------------------------------------------------------------------Package Thermal Resistance (Note 2)
UQFN-13JL 2x3 (FC), θJA -----------------------------------------------------------------------------------------UQFN-13JL 2x3 (FC), θJC ----------------------------------------------------------------------------------------Junction Temperature Range -------------------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------Storage Temperature Range -------------------------------------------------------------------------------------ESD Susceptibility (Note 3)
HBM (Human Body Model) ----------------------------------------------------------------------------------------
Recommended Operating Conditions
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−0.3V to 21V
−0.3V to (VIN + 0.3V)
−3V to (VIN + 0.3V)
−0.3V to 27.3V
−0.3V to 6V
−0.3V to 6V
1.54W
64.8°C/W
10°C/W
150°C
260°C
−65°C to 150°C
2kV
(Note 4)
Supply Voltage, VIN ------------------------------------------------------------------------------------------------ 4.5V to 18V
Junction Temperature Range -------------------------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range -------------------------------------------------------------------------------------- −40°C to 85°C
Electrical Characteristics
(VIN = 12V, TA = 25°C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
4.5
--
18
V
4
4.2
4.4
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
Under-Voltage Lockout
Threshold
VUVLO
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
Shutdown Current
ISHDN
Quiescent Current
IQ
V
--
0.5
--
VEN = 0V
--
1.5
10
A
VEN = 2V, VFB = 0.7V
--
0.8
1.2
mA
1.1
1.2
1.3
V
--
200
--
mV
Enable Voltage
Enable Voltage Threshold
VEN Rising
Enable Voltage Hysteresis
Feedback Voltage
Feedback Voltage Threshold
VFB
4.5V  VIN  18V
0.693
0.7
0.707
V
Feedback Input Current
IFB
VFB = 0.71V
0.1
--
0.1
A
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DS6207A/B-02
December 2015
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RT6207A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
6V  VIN  18V, 0 < IPVCC  5mA
--
5
--
V
PVCC Line Regulation
6V  VIN  18V, IPVCC = 5mA
--
--
20
PVCC Load Regulation
0  IPVCC  20mA
--
--
100
IPVCC
VIN = 6V, VPVCC = 4V
--
150
--
High-Side On-Resistance
RDS(ON)_H
VBOOT  VSW = 5V
--
60
--
Low-Side On-Resistance
RDS(ON)_L
--
22
--
ILIM
5.6
6.7
7.9
TSD
--
150
--
--
20
--
--
153
--
PVCC Output
PVCC Output Voltage
PVCC Output Current
VPVCC
mV
mA
Internal MOSFET
m
Current Limit
Low-Side Switch Valley
Current Limit
A
Thermal Shutdown
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis TSD
C
On-Time Timer Control
On-Time
tON
VOUT = 1.2V
Minimum On-Time
tON(MIN)
--
60
--
Minimum Off-Time
tOFF(MIN)
--
230
--
VSS = 0V
5
6
7
VFB Rising (Good)
85
90
95
VFB Rising (Fault)
--
120
--
ns
Soft-Start
Internal Charge Current
A
Power Good
PGOOD Good Rising
Threshold
%VFB
PGOOD Good Falling
Threshold
VFB Falling (Good)
--
112
--
VFB Falling (Fault)
--
80
--
PGOOD Sink Current
PGOOD = 0.1V
10
20
--
mA
115
120
125
%VFB
--
10
--
s
UVP Detect
55
60
65
Hysteresis
--
10
--
--
250
--
s
--
tSS
x 1.7
--
--
Output Under Voltage and Over Voltage Protections
OVP Trip Threshold
OVP Detect
OVP Propagation Delay
UVP Trip Threshold
UVP Propagation Delay
UVP Enable Delay
Relative to Soft-Start Time
%VFB
Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may
affect device reliability.
Note 2. θJA is measured at TA = 25°C on a highly thermal conductive 4-Layer test board. θJC is measured at the top of the
package.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
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is a registered trademark of Richtek Technology Corporation.
DS6207A/B-02
December 2015
RT6207A/B
Typical Application Circuit
9, 10
VIN
CIN
10µF x 2
VPVCC
RPG
Enable
L
RT6207A/B
12
SW
VIN
BOOT
VOUT
CBOOT
11
0.1µF
8 PGOOD
2 EN
FB 3
5 SS
PVCC 6
CSS
CFF
VPVCC
CPVCC
GND
1, 13
R1
COUT
22µF x 3
R2
1µF
Table 1. Suggested Component Values (VIN = 12V)
VOUT (V)
R1 (k)
R2 (k)
L (H)
COUT (F)
CFF (pF)
1
10.2
24
1
66
--
1.2
17
24
1.5
66
--
1.8
37.4
24
2.2
66
--
2.5
61.9
24
2.2
66
22
5
147
24
3.3
66
22
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December 2015
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9
RT6207A/B
Typical Operating Characteristics
Efficiency vs. Output Current
100
90
90
80
80
VIN = 4.5V
VIN = 12V
VIN = 18V
70
60
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
100
50
40
30
20
VIN = 4.5V
VIN = 12V
VIN = 18V
70
60
50
40
30
20
10
10
RT6207A, VOUT = 1.2V
0
0
1
2
3
4
RT6207B, VOUT = 1.2V
0
5
0
1
2
Output Current (A)
Efficiency vs. Output Current
100
90
90
VOUT = 5V
VOUT = 3.3V
VOUT = 1.2V
60
50
40
30
20
70
60
50
40
30
20
10
10
RT6207A, VIN = 12V
0
RT6207B, VIN = 12V
0
0
1
2
3
4
5
0
1
2
Output Current (A)
1.23
1.22
1.22
1.21
Output Voltage (V)
1.23
RT6207A
1.20
RT6207B
1.19
4
5
Output Voltage vs. Input Voltage
1.24
1.21
3
Output Current (A)
Output Voltage vs. Output Current
Output Voltage (V)
5
VOUT = 5V
VOUT = 3.3V
VOUT = 1.2V
80
Efficiency (%)
Efficiency (%)
70
4
Efficiency vs. Output Current
100
80
3
Output Current (A)
1.18
1.17
1.20
1.19
IOUT = 1A
IOUT = 3A
IOUT = 5A
1.18
1.17
1.16
VIN = 12V, VOUT = 1.2V
1.16
0
1
2
3
4
Output Current (A)
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10
VOUT = 1.2V
1.15
5
4
6
8
10
12
14
16
18
Input Voltage (V)
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RT6207A/B
Switching Frequency vs. Output Current
Switching Frequency vs. Temperature
850
700
RT6207A
600
RT6207B
800
Switching Frequency (kHz)1
Switching Frequency (kHz)1
800
500
400
300
200
100
750
700
650
600
550
500
IOUT = 1A
0
450
0
1
2
3
4
5
-50
-25
0
Output Current (A)
75
100
125
Feedback Voltage vs. Temperature
0.73
0.73
0.72
0.72
Feedback Voltage (V)
Feedback Voltage (V)
50
Temperature (°C)
Feedback Voltage vs. Input Voltage
0.71
0.70
0.69
0.68
0.67
0.71
0.70
0.69
0.68
0.67
0.66
0.66
4
6
8
10
12
14
16
18
-50
-25
0
Input Voltage (V)
25
50
75
100
125
Temperature (°C)
Inductor Valley Current Limit vs. Input Voltage
Inductor Valley Current Limit vs. Temperature
7.0
7.0
Inductor Valley Current Limit (A)1
Inductor Valley Current Limit (A)1
25
6.5
6.0
5.5
5.0
4.5
4.0
6.5
6.0
5.5
5.0
4.5
4.0
4
6
8
10
12
14
16
Input Voltage (V)
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DS6207A/B-02
December 2015
18
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT6207A/B
3.5
3.0
2.5
2.0
1.5
1.0
VEN = 0V
0.5
4
6
8
10
12
14
16
Shutdown Quiescent Current vs. Temperature
Shutdown Quiescent Current (μA)1
Shutdown Quiescent Current (μA)1
Shutdown Quiescent Current vs. Input Voltage
18
15
12
9
6
3
VEN = 0V
0
18
-50
-25
0
Input Voltage (V)
800
800
700
600
500
400
8
10
12
14
16
600
500
400
300
18
-50
-25
0
Input UVLO vs. Temperature
50
75
100
125
Enable Threshold vs. Temperature
1.5
4.4
1.4
Enable Threshold (V)
Input UVLO (V)
25
Temperature (°C)
4.6
4.2
Rising
3.8
3.6
125
700
Input Voltage (V)
4.0
100
VEN = 2V, VFB = 0.7V
VEN = 2V, VFB = 0.7V
6
75
Quiescent Current vs. Temperature
900
Quiescent Current (μA)
Quiescent Current (μA)
Quiescent Current vs. Input Voltage
4
50
Temperature (°C)
900
300
25
Falling
3.4
1.3
Rising
1.2
1.1
1.0
Falling
0.9
3.2
0.8
-50
-25
0
25
50
75
100
Temperature (°C)
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12
125
-50
-25
0
25
50
75
100
125
Temperature (°C)
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December 2015
RT6207A/B
Load Transient Response
VOUT
(100mV/Div)
IOUT
(2A/Div)
Load Transient Response
VOUT
(100mV/Div)
RT6207A, VIN = 12V, VOUT = 1.2V, IOUT = 0 to 5A
IOUT
(2A/Div)
Time (100μs/Div)
Time (100μs/Div)
Voltage Ripple
Voltage Ripple
VSW
(10V/Div)
VSW
(10V/Div)
VOUT
(10mV/Div)
VOUT
(10mV/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 5A
VIN = 12V, VOUT = 1.2V, IOUT = 1A
Time (1μs/Div)
Time (1μs/Div)
Power On from Input Voltage
Power Off from Input Voltage
VIN
(20V/Div)
VIN
(20V/Div)
VSW
(10V/Div)
VSW
(10V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
IOUT
(5A/Div)
IOUT
(5A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 5A
Time (25ms/Div)
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
DS6207A/B-02
RT6207B, VIN = 12V, VOUT = 1.2V, IOUT = 0 to 5A
December 2015
VIN = 12V, VOUT = 1.2V, IOUT = 5A
Time (25ms/Div)
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RT6207A/B
Power On from Enable Voltage
Power Off from Enable Voltage
VEN
(5V/Div)
VEN
(5V/Div)
VPGOOD
(5V/Div)
VPGOOD
(5V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
IOUT
(5A/Div)
IOUT
(5A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 5A
Time (25ms/Div)
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VIN = 12V, VOUT = 1.2V, IOUT = 5A
Time (2.5ms/Div)
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RT6207A/B
Application information
Inductor Selection
Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor value
is generally flexible and is ultimately chosen to obtain the
best mix of cost, physical size, and circuit efficiency.
Lower inductor values benefit from reduced size and cost
and they can improve the circuit's transient response, but
they increase the inductor ripple current and output voltage
ripple and reduce the efficiency due to the resulting higher
peak currents. Conversely, higher inductor values increase
efficiency, but the inductor will either be physically larger
or have higher resistance since more turns of wire are
required and transient response will be slower since more
time is required to change current (up or down) in the
inductor. A good compromise between size, efficiency,
and transient response is to use a ripple current (ΔIL) about
20% to 50% of the desired full output load current.
Calculate the approximate inductor value by selecting the
input and output voltages, the switching frequency (fSW),
the maximum output current (IOUT(MAX)) and estimating a
ΔIL as some percentage of that current.
L=
VOUT   VIN  VOUT 
VIN  fSW  IL
Once an inductor value is chosen, the ripple current (ΔIL)
is calculated to determine the required peak inductor
current.
VOUT   VIN  VOUT 
I
IL =
and IL(PEAK) = IOUT(MAX)  L
VIN  fSW  L
2
To guarantee the required output current, the inductor
needs a saturation current rating and a thermal rating that
exceeds IL(PEAK). These are minimum requirements. To
maintain control of inductor current in overload and short
circuit conditions, some applications may desire current
ratings up to the current limit value. However, the IC's
output under-voltage shutdown feature make this
unnecessary for most applications.
IL(PEAK)should not exceed the minimum value of IC's upper
current limit level or the IC may not be able to meet the
desired output current. If needed, reduce the inductor ripple
current (ΔIL) to increase the average inductor current (and
the output current) while ensuring that IL(PEAK) does not
exceed the upper current limit level.
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
DS6207A/B-02
December 2015
For best efficiency, choose an inductor with a low DC
resistance that meets the cost and size requirements.
For low inductor core losses some type of ferrite core is
usually best and a shielded core type, although possibly
larger or more expensive, will probably give fewer EMI
and other noise problems.
Considering the Typical Operating Circuit for 1.2V output
at 5A and an input voltage of 12V, using an inductor ripple
of 1A (20%), the calculated inductance value is :
L=
1.2  12  1.2 
= 1.66μH
12  650kHz  1A
The ripple current was selected at 1A and, as long as we
use the calculated 1.8μH inductance, that should be the
actual ripple current amount. The ripple current and required
peak current as below :
IL =
1.2  12  1.2 
= 0.923A
12  650kHz  1.8μH
and IL(PEAK) = 5A  0.923A = 5.4615A
2
For the 1.8μH value, the inductor's saturation and thermal
rating should exceed at least 5.4615A. For more
conservative, the rating for inductor saturation current must
be equal to or greater than switch current limit of the device
rather than the inductor peak current.
Input Capacitor Selection
The input filter capacitors are needed to smooth out the
switched current drawn from the input power source and
to reduce voltage ripple on the input. The actual
capacitance value is less important than the RMS current
rating (and voltage rating, of course). The RMS input ripple
current (IRMS) is a function of the input voltage, output
voltage, and load current :
V
IRMS = IOUT(MAX)  OUT
VIN
VIN
1
VOUT
Ceramic capacitors are most often used because of their
low cost, small size, high RMS current ratings, and robust
surge current capabilities. However, take care when these
capacitors are used at the input of circuits supplied by a
wall adapter or other supply connected through long, thin
wires. Current surges through the inductive wires can
induce ringing at the RT6207A/B input which could
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RT6207A/B
potentially cause large, damaging voltage spikes at VIN.
If this phenomenon is observed, some bulk input
capacitance may be required. Ceramic capacitors (to meet
the RMS current requirement) can be placed in parallel
with other types such as tantalum, electrolytic, or polymer
(to reduce ringing and overshoot).
Output Transient Undershoot and Overshoot
Choose capacitors rated at higher temperatures than
required. Several ceramic capacitors may be paralleled to
meet the RMS current, size, and height requirements of
the application. The typical operating circuit uses two 10μF
and one 0.1μF low ESR ceramic capacitors on the input.
However, the combination of small ceramic output
capacitors (with little capacitance), low output voltages
(with little stored charge in the output capacitors), and
low duty cycle applications (which require high inductance
to get reasonable ripple currents with high input voltages)
increases the size of voltage variations in response to
very quick load changes. Typically, load changes occur
slowly with respect to the IC's 650kHz switching frequency.
Output Capacitor Selection
The RT6207A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level and
transient response requirements for sag (undershoot on
positive load steps) and soar (overshoot on negative load
steps).
Output Ripple
Output ripple at the switching frequency is caused by the
inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are similar
in amplitude and both should be considered if ripple is
critical.
VRIPPLE = VRIPPLE(ESR)  VRIPPLE(C)
VRIPPLE(ESR) = IL  RESR
VRIPPLE(C) =
IL
8  COUT  fSW
For the Typical Operating Circuit for 1.2V output and an
inductor ripple of 0.923A, with 3 x 22μF output capacitance
each with about 5mΩ ESR including PCB trace resistance,
the output voltage ripple components are :
VRIPPLE(ESR) = 0.923A  5m = 4.615mV
VRIPPLE(C) =
0.923A
= 2.689mV
8  66μF  650kHz
VRIPPLE = 4.615mV+2.689mV = 7.304mV
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In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load steps
up and down abruptly. The ACOT transient response is
very quick and output transients are usually small.
But some modern digital loads can exhibit nearly
instantaneous load changes and the following section
shows how to calculate the worst-case voltage swings in
response to very fast load steps.
The output voltage transient undershoot and overshoot each
have two components : the voltage steps caused by the
output capacitor's ESR, and the voltage sag and soar due
to the finite output capacitance and the inductor current
slew rate. Use the following formulas to check if the ESR
is low enough (typically not a problem with ceramic
capacitors) and the output capacitance is large enough to
prevent excessive sag and soar on very fast load step
edges, with the chosen inductor value.
The amplitude of the ESR step up or down is a function of
the load step and the ESR of the output capacitor :
VESR _STEP = ΔIOUT x RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the maximum
duty cycle. The maximum duty cycle during a fast transient
is a function of the on-time and the minimum off-time since
the ACOTTM control scheme will ramp the current using
on-times spaced apart with minimum off-times, which is
as fast as allowed. Calculate the approximate on-time
(neglecting parasites) and maximum duty cycle for a given
input and output voltage as :
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RT6207A/B
tON =
VOUT
tON
and DMAX =
VIN  fSW
tON  tOFF(MIN)
The actual on-time will be slightly longer as the IC
compensates for voltage drops in the circuit, but we can
neglect both of these since the on-time increase
compensates for the voltage losses. Calculate the output
voltage sag as :
L  (IOUT )2
VSAG =
2  COUT   VIN(MIN)  DMAX  VOUT 
The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
and the output voltage :
VSOAR =
L  (IOUT )2
2  COUT  VOUT
For the Typical Operating Circuit for 1.2V output, the circuit
has an inductor 1.8μH and 3 x 22μF output capacitance
with 5mΩ ESR each. The ESR step is 5A x 1.67mΩ =
8.35mV which is small, as expected. The output voltage
sag and soar in response to full 0A-5A-0A instantaneous
transients are :
tON =
1.2V
= 153ns
12V  650kHz
and DMAX =
153ns
= 0.399
153ns  230ns
where 230ns is the minimum off time
VSAG =
1.8μH  (5A)2
= 95mV
2  66μF  12V  0.399  1.2V 
VSOAR =
1.8μH  (5A)2
= 284mV
2  66μF  1.2V
The sag is about 7.92% of the output voltage and the soar
is a full 23.7% of the output voltage. The ESR step is
negligible here but it does partially add to the soar, so
keep that in mind whenever using higher-ESR output
capacitors.
The soar is typically much worse than the sag in high
input, low-output step-down converters because the high
input voltage demands a large inductor value which stores
lots of energy that is all transferred into the output if the
load stops drawing current. Also, for a given inductor, the
soar for a low output voltage is a greater voltage change
and an even greater percentage of the output voltage.
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DS6207A/B-02
December 2015
Any sag is always short-lived, since the circuit quickly
sources current to regain regulation in only a few switching
cycles. With the RT6207B, any overshoot transient is
typically also short-lived since the converter will sink
current, reversing the inductor current sharply until the
output reaches regulation again. The RT6207A
discontinuous operation at light loads prevents sinking
current so, for that IC, the output voltage will soar until
load current or leakage brings the voltage down to normal.
Most applications never experience instantaneous full load
steps and the RT6207A/B high switching frequency and
fast transient response can easily control voltage regulation
at all times. Also, since the sag and soar both are
proportional to the square of the load change, if load steps
were reduced to 1A (from the 5A examples preceding) the
voltage changes would be reduced by a factor of almost
ten. For these reasons sag and soar are seldom an issue
except in very low-voltage CPU core or DDR memory
supply applications, particularly for devices with high clock
frequencies and quick changes into and out of sleep
modes. In such applications, simply increasing the amount
of ceramic output capacitor (sag and soar are directly
proportional to capacitance) or adding extra bulk
capacitance can easily eliminate any excessive voltage
transients.
In any application with large quick transients, always
calculate soar to make sure that over-voltage protection
will not be triggered. Under-voltage is not likely since the
threshold is very low (60%), that function has a long delay
(250μs), and the IC will quickly return the output to
regulation. Over-voltage protection has a minimum
threshold of 120% and short delay of 10μs and can actually
be triggered by incorrect component choices, particularly
for the RT6207A which does not sink current.
Feed-forward Capacitor (Cff)
The RT6207A/B are optimized for ceramic output
capacitors and for low duty cycle applications. However
for high-output voltages, with high feedback attenuation,
the circuit's response becomes over-damped and transient
response can be slowed. In high-output voltage circuits
(VOUT > 3.3V) transient response is improved by adding a
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RT6207A/B
small “feed-forward” capacitor (Cff) across the upper FB
divider resistor (Figure 1), to increase the circuit's Q and
reduce damping to speed up the transient response without
affecting the steady-state stability of the circuit. Choose
a suitable capacitor value that following below step.

Get the BW the quickest method to do transient
response form no load to full load. Confirm the damping
frequency. The damping frequency is BW.
VOUT
Cff
EN
VIN
FB
RT6207A/B
For automatic start-up the low-voltage EN pin can be
connected to VIN through a 100kΩ resistor. Its large
hysteresis band makes EN useful for simple delay and
timing circuits. EN can be externally pulled to VIN by
adding a resistor-capacitor delay (REN and CEN in Figure
2). Calculate the delay time using EN's internal threshold
where switching operation begins.
An external MOSFET can be added to implement digital
control of EN when no system voltage above 2V is available
(Figure 3). In this case, a 100kΩ pull-up resistor, REN, is
connected between VIN and the EN pin. MOSFET Q1 will
be under logic control to pull down the EN pin. To prevent
enabling circuit when VIN is smaller than the VOUT target
value or some other desired voltage level, a resistive voltage
divider can be placed between the input voltage and ground
and connected to EN to create an additional input under
voltage lockout threshold (Figure 4).
BW
R1
Enable Operation (EN)
R2
REN
EN
RT6207A/B
CEN
GND
GND
Figure 1. Cff Capacitor Setting

Cff can be calculated base on below equation :
1
Cff 
2  3.1412  R1 BW  0.8
Figure 2. External Timing Control
VIN
REN
100k
EN
Q1
Enable
RT6207A/B
Soft-Start (SS)
The RT6207A/B soft-start uses an external capacitor at
SS to adjust the soft-start timing according to the following
equation :
CSS  nF   0.7V
t  ms  
ISS μA 
Following below equation to get the minimum capacitance
range in order to avoid UV occur.
COUT  VOUT  0.6  1.2
T
(ILIM  Load Current)  0.8
T  6μA
CSS 
VREF
GND
Figure 3. Digital Enable Control Circuit
VIN
REN1
REN2
EN
RT6207A/B
GND
Figure 4. Resistor Divider for Lockout Threshold Setting
Do not leave SS unconnected.
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RT6207A/B
Output Voltage Setting
External BOOT Capacitor Series Resistance
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected to
FB. The output voltage is set according to the following
equation :
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low power
loss and good efficiency, but also slow enough to reduce
EMI. Switch turn-on is when most EMI occurs since VSW
rises rapidly. During switch turn-off, SW is discharged
relatively slowly by the inductor current during the dead
time between high-side and low-side switch on-times. In
some cases it is desirable to reduce EMI further, at the
expense of some additional power dissipation. The switch
turn-on can be slowed by placing a small (<47Ω)
resistance between BOOT and the external bootstrap
capacitor. This will slow the high-side switch turn-on and
VSW's rise. To remove the resistor from the capacitor
charging path (avoiding poor enhancement due to
undercharging the BOOT capacitor), use the external diode
shown in Figure 6 to charge the BOOT capacitor and place
the resistance between BOOT and the capacitor/diode
connection.
VOUT = 0.7V x (1 + R1 / R2)
VOUT
R1
FB
RT6207A/B
R2
GND
Figure 5. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin. Choose
R2 between 10kΩ and 100kΩ to minimize power
consumption without excessive noise pick-up and
calculate R1 as follows :
R2  (VOUT  VREF )
VREF
For output voltage accuracy, use divider resistors with 1%
or better tolerance.
R1 
External BOOT Bootstrap Diode
When the input voltage is lower than 5.5V it is
recommended to add an external bootstrap diode between
VIN (or VINR) and the BOOT pin to improve enhancement
of the internal MOSFET switch and improve efficiency.
The bootstrap diode can be a low cost one such as 1N4148
or BAT54.
5V
0.1µF
SW
Figure 6. External Bootstrap Diode
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DS6207A/B-02
December 2015
Decouple PVCC to GND with a 1μF ceramic capacitor.
High grade dielectric (X7R, or X5R) ceramic capacitors
are recommended for their stable temperature and bias
voltage characteristics.
Thermal Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature. The
maximum power dissipation can be calculated by the
following formula :
PD(MAX) = (TJ(MAX) − TA) / θJA
BOOT
RT6207A/B
PVCC Capacitor Selection
where TJ(MAX) is the maximum junction temperature, TA is
the ambient temperature, and θJA is the junction to ambient
thermal resistance.
For recommended operating condition specifications, the
maximum junction temperature is 125°C. The junction to
ambient thermal resistance, θJA, is layout dependent. For
UQFN-13JL 2x3 (FC) package, the thermal resistance,
θJA, is 64.8°C/W on a standard four-layer thermal test
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RT6207A/B
board. The maximum power dissipation at TA = 25°C can
be calculated by the following formula :
Layout Consideration
PD(MAX) = (125°C − 25°C) / (64.8°C/W) = 1.54W for
UQFN-13JL 2x3 (FC) package
The maximum power dissipation depends on the operating
ambient temperature for fixed T J(MAX) and thermal
resistance, θJA. The derating curve in Figure 7 allows the
designer to see the effect of rising ambient temperature
on the maximum power dissipation.
Maximum Power Dissipation (W)1
2.4

Follow the PCB layout guidelines for optimal
performance of the device.

Keep the traces of the main current paths as short and
wide as possible.

Put the input capacitor as close as possible to VIN and
VIN pins.

SW node is with high frequency voltage swing and
should be kept at small area. Keep analog components
away from the SW node to prevent stray capacitive noise
pickup.

Connect feedback network behind the output capacitors.
Keep the loop area small. Place the feedback
components near the device.

Connect all analog grounds to common node and then
connect the common node to the power ground behind
the output capacitors.

An example of PCB layout guide is shown in Figure 8
for reference.
Four-Layer PCB
2.0
1.6
1.2
0.8
0.4
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
Figure 7. Derating Curve of Maximum Power Dissipation
Place the input and output capacitors
as close to the IC as possible.
SW should be connected to inductor by
wide and short trace, and keep sensitive
components away from this trace.
VOUT
CBOOT
L
R2
SW
BOOT
12
11
CIN
GND
1
10
VIN
EN
2
9
VIN
FB
3
8
PGOOD
NC
4
7
NC
CSS
5
6
PVCC
Place the feedback as
close to the IC as possible.
13
SS
R1
GND
COUT
GND
GND
CPVCC
Figure 8. PCB Layout Guide
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
www.richtek.com
20
is a registered trademark of Richtek Technology Corporation.
DS6207A/B-02
December 2015
RT6207A/B
Outline Dimension
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min.
Max.
Min.
Max.
A
0.500
0.600
0.020
0.024
A1
0.000
0.050
0.000
0.002
A3
0.100
0.152
0.004
0.006
b
0.200
0.300
0.008
0.012
b1
0.370
0.470
0.015
0.019
D
1.900
2.100
0.075
0.083
E
2.900
3.100
0.114
0.122
K
0.750
0.030
e
0.500
0.020
e1
0.585
0.023
e2
0.500
0.020
L
0.400
0.500
0.016
0.020
L1
0.950
1.050
0.037
0.041
L2
0.325
0.425
0.013
0.017
L3
1.325
1.425
0.052
0.056
U-Type 13JL QFN 2x3 (FC) Package
Copyright © 2015 Richtek Technology Corporation. All rights reserved.
DS6207A/B-02
December 2015
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
21
RT6207A/B
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot
assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be
accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
www.richtek.com
22
DS6207A/B-02
December 2015