DATASHEET

ISL6266, ISL6266A
Data Sheet
Two-phase Core Controllers
(Montevina, IMVP-6+)
The ISL6266 and ISL6266A are two-phase buck converter
regulators implementing Intel® IMVP-6 protocol with
embedded gate drivers. Both converters use interleaved
channels to double the output voltage ripple frequency and
thereby reduce output voltage ripple amplitude with fewer
components, lower component cost, reduced power
dissipation, and smaller real estate area.
The ISL6266A utilizes the patented R3 Technology™,
Intersil’s Robust Ripple Regulator modulator. Compared with
traditional multiphase buck regulators, the R3 Technology™
has the fastest transient response. This is due to the R3
modulator commanding variable switching frequency during
load transient events.
August 25, 2015
FN6398.4
Features
• Precision Two/One-phase CORE Voltage Regulator
- 0.5% System Accuracy Over-Temperature
- Enhanced Load Line Accuracy
• Internal Gate Driver with 2A Driving Capability
• Dynamic Phase Adding/Dropping
• Microprocessor Voltage Identification Input
- 7-Bit VID Input
- 0.300V to 1.500V in 12.5mV Steps
- Support VID Change On-the-Fly
• Multiple Current Sensing Schemes Supported
- Lossless Inductor DCR Current Sensing
- Precision Resistive Current Sensing
Intel Mobile Voltage Positioning (IMVP) is a smart voltage
regulation technology, which effectively reduces power
dissipation in Intel Pentium processors. To boost battery life,
the ISL6266A supports DPRSLRVR (deeper sleep),
DPRSTP# and PSI# functions, which maximizes efficiency
by enabling different modes of operation. In active mode
(heavy load), the regulator commands the two phase
continuous conduction mode (CCM) operation. When PSI#
is asserted in active mode (medium load), the ISL6266A
operates in one-phase CCM. When the CPU enters deeper
sleep mode, the ISL6266A enables diode emulation to
maximize efficiency.
• CPU Power Monitor
For better system power management, the ISL6266A
provides a CPU power monitor output. The analog output at
the power monitor pin can be fed into an A/D converter to
report instantaneous or average CPU power.
Ordering Information
A 7-bit digital-to-analog converter (DAC) allows dynamic
adjustment of the core output voltage from 0.300V to 1.500V.
Over-temperature, the ISL6266A achieves a 0.5% system
accuracy of core output voltage.
ISL6266HRZ
(No longer
available or
supported)
ISL6266 HRZ
-10 to +100 48 Ld 7x7 QFN L48.7x7
ISL6266HRZ-T*
(No longer
available or
supported)
ISL6266 HRZ
-10 to +100 48 Ld 7x7 QFN L48.7x7
ISL6266AIRZ
ISL6266A IRZ -40 to +100 48 Ld 7x7 QFN L48.7x7
A unity-gain differential amplifier is provided for remote CPU
die sensing. This allows the voltage on the CPU die to be
accurately measured and regulated per Intel IMVP-6+
specifications. Current sensing can be realized using either
lossless inductor DCR sensing or discrete resistor sensing.
A single NTC thermistor network thermally compensates the
gain and the time constant of the DCR variations.
The ISL6266 also includes all the functions for IMVP-6+
core power delivery. In addition, it has been optimized for
use with coupled-inductor solutions. More information on the
differences between ISL6266 and ISL6266A can be found in
the “Electrical Specifications” on page 3 and the “ISL6266
Features” on page 21.
1
• Thermal Monitor
• User Programmable Switching Frequency
• Differential Remote CPU Die Voltage Sensing
• Static and Dynamic Current Sharing
• Support All Ceramic Output with Coupled Inductor
(ISL6266)
• Overvoltage, Undervoltage and Overcurrent Protection
• Pb-Free (RoHS Compliant)
PART NUMBER
(Note)
PART
MARKING
TEMP.
RANGE
(°C)
PACKAGE
(Pb-free)
PKG.
DWG. #
ISL6266AIRZ-T* ISL6266A IRZ -40 to +100 48 Ld 7x7 QFN L48.7x7
*Please refer to TB347 for details on reel specifications.
NOTE: These Intersil Pb-free plastic packaged products employ special
Pb-free material sets, molding compounds/die attach materials, and
100% matte tin plate plus anneal (e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering
operations). Intersil Pb-free products are MSL classified at Pb-free peak
reflow temperatures that meet or exceed the Pb-free requirements of
IPC/JEDEC J STD-020.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas LLC.
Copyright Intersil Americas LLC. 2007-2010, 2015. All Rights Reserved. R3 Technology™ is a trademark of Intersil Americas LLC.
All other trademarks mentioned are the property of their respective owners.
ISL6266, ISL6266A
Pinout
3V3
CLK_EN#
DPRSTP#
DPRSLPVR
VR_ON
VID6
VID5
VID4
VID3
VID2
VID1
VID0
ISL6266, ISL6266A
(48 LD 7x7 QFN)
TOP VIEW
48
47
46
45
44
43
42
41
40
39
38
37
PGOOD
1
36 BOOT1
PSI#
2
35 UGATE1
PMON
3
34 PHASE1
RBIAS
4
33 PGND1
VR_TT#
5
32 LGATE1
NTC
6
SOFT
7
OCSET
8
29 PGND2
VW
9
28 PHASE2
COMP 10
27 UGATE2
31 PVCC
GND PAD
(BOTTOM)
30 LGATE2
FB 11
26 BOOT2
2
13
14
15
16
17
18
19
20
21
22
23
24
VSEN
RTN
DROOP
DFB
VO
VSUM
VIN
GND
VDD
ISEN2
ISEN1
25 NC
VDIFF
FB2 12
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Absolute Maximum Ratings
Thermal Information
Supply Voltage (VDD) . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V
Battery Voltage (VIN) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +28V
Boot Voltage (BOOT) . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
Boot to Phase Voltage (BOOT to PHASE). . . . . . -0.3V to +7V (DC)
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +9V (<10ns)
Phase Voltage (PHASE) . . . . . . . . . -7V (<20ns Pulse Width, 10µJ)
UGATE Voltage (UGATE) . . . . . . . . . . PHASE -0.3V (DC) to BOOT
. . . . . . . . . . . . . .PHASE-5V (<20ns Pulse Width, 10µJ) to BOOT
LGATE Voltage (LGATE) . . . . . . . . . . . -0.3V (DC) to (VDD + 0.3V)
. . . . . . . . . . . . . .-2.5V (<20ns Pulse Width, 5µJ) to (VDD + 0.3V)
All Other Pins . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VDD + 0.3V)
Open Drain Outputs, PGOOD, VR_TT# . . . . . . . . . . . -0.3V to +7V
Thermal Resistance (Typical)
JA°C/W
JC°C/W
QFN Package (Notes 1, 2). . . . . . . . . .
29
4.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . -65°C to +150°C
Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5%
Battery Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V to 25V
Ambient Temperature . . . . . . . . . . . . . . . . . . . . . . -40°C to +100°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . -40°C to +125°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For JC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
VDD = 5V, TA = -40°C to +100°C, unless otherwise specified.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 4)
TYP
MAX
(Note 4) UNITS
INPUT POWER SUPPLY
+5V Supply Current
IVDD
VR_ON = 3.3V
5.1
5.7
mA
VR_ON = 0V
1
µA
+3.3V Supply Current
I3V3
No load on CLK_EN#
1
µA
Battery Supply Current at VIN pin
IVIN
VR_ON = 0V, VIN = 25V
1
µA
POR (Power-On Reset) Threshold
PORr
VDD Rising
4.5
V
PORf
VDD Falling
4.0
No load, closed loop, active mode,
TA = 0°C to +100°C, VID = 0.75V to 1.5V
-0.5
0.5
%
VID = 0.5V to 0.7375V
-8
8
mV
VID = 0.3V to 0.4875V
-15
15
mV
No load, closed loop, active mode,
VID = 0.75V to 1.5V
-0.8
0.8
%
VID = 0.5V to 0.7375V
-10
10
mV
VID = 0.3V to 0.4875V
-18
18
mV
RRBIAS = 147k
1.45
1.47
1.49
V
1.188
1.2
1.212
V
4.35
4.15
V
SYSTEM AND REFERENCES
System Accuracy ( ISL6266AHRZ)
System Accuracy (ISL6266AIRZ)
%Error
(VCC_CORE)
%Error
(Vcc_core)
RBIAS Voltage
RRBIAS
Boot Voltage
VBOOT
Output Voltage Range
VID Off State
3
VCC_CORE
(max)
VID = [0000000]
1.5
V
VCC_CORE
(min)
VID = [1100000]
0.3
V
VID = [1111111]
0
V
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Electrical Specifications
VDD = 5V, TA = -40°C to +100°C, unless otherwise specified. (Continued)
PARAMETER
MIN
(Note 4)
TYP
ISL6266, 2 channel operation
410
440
470
kHz
ISL6266A, 2 channel operation
280
300
320
kHz
100
600
kHz
-0.25
0.25
mV
SYMBOL
TEST CONDITIONS
MAX
(Note 4) UNITS
CHANNEL FREQUENCY
Nominal Channel Frequency
fSW
Adjustment Range
AMPLIFIERS
Droop Amplifier Offset
Error Amp DC Gain
AV0
Error Amp Gain-Bandwidth Product
Error Amp Slew Rate
FB Input Current
(Note 3)
90
dB
GBW
CL = 20pF (Note 3)
18
MHz
SR
CL = 20pF (Note 3)
5
V/µs
10
IIN(FB)
150
nA
2
mV
ISEN
Imbalance Voltage
Input Bias Current
20
nA
SOFT-START CURRENT
Soft-Start Current
ISS
Soft Geyserville Current
IGV
|SOFT - REF| > 100mV
Soft Deeper Sleep Entry Current
IC4
Soft Deeper Sleep Exit Current
Soft Deeper Sleep Exit Current
-47
-42
-37
µA
±180
±205
±230
µA
DPRSLPVR = 3.3V
-47
-42
-37
µA
IC4EA
DPRSLPVR = 3.3V
37
42
47
µA
IC4EB
DPRSLPVR = 0V
180
205
230
µA
1.5

GATE DRIVER DRIVING CAPABILITY
UGATE Source Resistance
RSRC(UGATE)
500mA Source Current (Note 3)
1
UGATE Source Current
ISRC(UGATE)
VUGATE_PHASE = 2.5V (Note 3)
2
UGATE Sink Resistance
RSNK(UGATE)
500mA Sink Current (Note 3)
1
UGATE Sink Current
ISNK(UGATE)
VUGATE_PHASE = 2.5V (Note 3)
2
LGATE Source Resistance
RSRC(LGATE)
500mA Source Current (Note 3)
1
LGATE Source Current
ISRC(LGATE)
VLGATE = 2.5V (Note 3)
2
LGATE Sink Resistance
RSNK(LGATE)
500mA Sink Current (Note 3)
LGATE Sink Current
ISNK(LGATE)
VLGATE = 2.5V (Note 3)
UGATE to PHASE Resistance
0.5
Rp(UGATE)
A
1.5

A
1.5

A
0.9

4
A
1
k
GATE DRIVER SWITCHING TIMING (refer to “ISL6266, ISL6266A Gate Driver Timing Diagram” on page 6)
UGATE Rise Time
tRU
PVCC = 5V, 3nF Load (Note 3)
8.0
ns
LGATE Rise Time
tRL
PVCC = 5V, 3nF Load (Note 3)
8.0
ns
UGATE Fall Time
tFU
PVCC = 5V, 3nF Load (Note 3)
8.0
ns
LGATE Fall Time
tFL
PVCC = 5V, 3nF Load
4.0
ns
UGATE Turn-on Propagation Delay
4
tPDHU
ISL6266AHRZ
TA = -10°C to +100°C
PVCC = 5V, Outputs Unloaded
20
30
44
ns
tPDHU
ISL6266AIRZ
PVCC = 5V, Outputs Unloaded
18
30
44
ns
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Electrical Specifications
VDD = 5V, TA = -40°C to +100°C, unless otherwise specified. (Continued)
PARAMETER
SYMBOL
LGATE Turn-on Propagation Delay
tPDHL
ISL6266AHRZ
tPDHL
ISL6266AIRZ
MIN
(Note 4)
TYP
TA = -10°C to +100°C
PVCC = 5V, Outputs Unloaded
7
15
30
ns
PVCC = 5V, Outputs Unloaded
5
15
30
ns
0.43
0.58
0.72
V
1
µA
0.4
V
1
µA
TEST CONDITIONS
MAX
(Note 4) UNITS
BOOTSTRAP DIODE
Forward Voltage
VDDP = 5V, Forward Bias Current = 2mA
Leakage
VR = 16V
POWER GOOD and PROTECTION MONITOR
PGOOD Low Voltage
VOL
IPGOOD = 4mA
PGOOD Leakage Current
IOH
PGOOD = 3.3V
-1
PGOOD Delay
tpgd
CLK_EN# Low to PGOOD High
6.3
7.6
8.9
ms
Overvoltage Threshold
OVH
VO rising above setpoint >1ms
155
195
235
mV
OVHS
VO rising above setpoint >0.5µs
1.675
1.7
1.725
V
10
10.2
µA
3.5
mV
Severe Overvoltage Threshold
0.26
OCSET Reference Current
I(RBIAS) = 10µA
9.8
OC Threshold Offset
DROOP rising above OCSET >120µs
-3.5
Current Imbalance Threshold
Difference between ISEN1 and ISEN2 >1ms
Undervoltage Threshold
(VDIFF-SOFT)
UVf
VO falling below setpoint for >1ms
9
-360
-300
mV
-240
mV
1
V
LOGIC INPUTS
VR_ON, DPRSLPVR Input Low
VIL(3.3V)
VR_ON, DPRSLPVR Input High
VIH(3.3V)
Leakage Current of VR_ON
IIL(3.3V)
Logic input is low
IIH(3.3V)
Logic input is high at 3.3V
2.3
IIL_DPRSLP(3.3V) DPRSLPVR input is low
Leakage Current of DPRSLPVR
-1
VIL(1V)
DAC(VID0-VID6), PSI# and
DPRSTP# Input High
VIH(1V)
Leakage Current of DAC
(VID0-VID6), PSI# and DPRSTP#
IIL(1V)
Logic input is low
IIH(1V)
Logic input is high at 1V
0
0
-1
IIH_DPRSLP(3.3V) DPRSLPVR input is high at 3.3V
DAC(VID0-VID6), PSI# and
DPRSTP# Input Low
V
µA
1
µA
0
0.45
µA
1
µA
0.3
V
0.7
-1
V
0
µA
0.45
1
µA
53
60
67
µA
1.18
1.2
1.22
V
6.5
9

THERMAL MONITOR
NTC Source Current
NTC = 1.3V
Over-Temperature Threshold
V(NTC) falling
VR_TT# Low Output Resistance
RTT
I = 20mA
POWER MONITOR
PMON Output Voltage Range
Vpmon
PMON Maximum Voltage
Vpmonmax
5
VSEN = 1.2V, Droop - VO = 80mV
1.638
1.680
1.722
V
VSEN = 1V, Droop - VO = 20mV
0.308
0.350
0.392
V
2.8
3.0
V
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Electrical Specifications
VDD = 5V, TA = -40°C to +100°C, unless otherwise specified. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
(Note 4)
TYP
MAX
(Note 4) UNITS
PMON Sourcing Current
Isc_pmon
VSEN = 1V, Droop - VO = 50mV
2
mA
PMON Sinking Current
Isk_pmon
VSEN = 1V, Droop - VO = 50mV
2
mA
Maximum Current Sinking Capability
Refer to Figure 29
PMON/
250
PMON Impedance
When PMON is within its sourcing/sinking
current range (Note 3)
PMON/
180
PMON/
100
A
7

3.1
V
CLK_EN# OUTPUT LEVELS
CLK_EN# High Output Voltage
VOH
3V3 = 3.3V, I = -4mA
CLK_EN# Low Output Voltage
VOL
ICLK_EN# = 4mA
2.9
0.26
0.4
V
NOTES:
3. Limits established by characterization and are not production tested.
4. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
ISL6266, ISL6266A Gate Driver Timing Diagram
PWM
tPDHU
tFU
tRU
1V
UGATE
1V
LGATE
tRL
tFL
tPDHL
6
FN6398.4
August 25, 2015
ISL6266, ISL6266A
3V3
CLK_EN#
DPRSTP#
DPRSLPVR
VR_ON
VID6
VID5
VID4
VID3
VID2
VID1
VID0
Functional Pin Description
48
47
46
45
44
43
42
41
40
39
38
37
PGOOD
1
36 BOOT1
PSI#
2
35 UGATE1
PMON
3
34 PHASE1
RBIAS
4
33 PGND1
VR_TT#
5
32 LGATE1
NTC
6
SOFT
7
OCSET
8
29 PGND2
VW
9
28 PHASE2
COMP 10
27 UGATE2
31 PVCC
GND PAD
(BOTTOM)
30 LGATE2
FB 11
26 BOOT2
7
13
14
15
16
17
18
19
20
21
22
23
24
VSEN
RTN
DROOP
DFB
VO
VSUM
VIN
GND
VDD
ISEN2
ISEN1
25 NC
VDIFF
FB2 12
FN6398.4
August 25, 2015
ISL6266, ISL6266A
PGOOD - Power good open-drain output. Connect
externally with 680 to VCCP or 1.9k to 3.3V.
N/C - Not connected. Grounding this pin to signal ground in
the practical layout.
PSI# - Current indicator input. When asserted low, indicates
a reduced load-current condition and initiates single-phase
operation.
BOOT2 - This pin is the upper gate driver supply voltage for
Phase 2. An internal boot strap diode is connected to the
PVCC pin.
PMON - Analog output. PMON is proportional to the product
of Vsen and droop voltage.
UGATE2 - Upper MOSFET gate signal for Phase 2.
RBIAS - 147k resistor to VSS sets internal current
reference.
VR_TT# - Thermal overload output indicator with open-drain
output. Over-temperature pull-down resistance is 10.
NTC - Thermistor input to VRTT# circuit and a 60µA current
source is connected internally to this pin.
SOFT - A capacitor from this pin to GND sets the maximum
slew rate of the output voltage. SOFT is the non-inverting
input of the error amplifier.
OCSET - Overcurrent set input. A resistor from this pin to
VO sets DROOP voltage limit for OC trip. A 10µA current
source is connected internally to this pin.
VW - A resistor from this pin to COMP programs the
switching frequency (for example, 6.45k 400kHz).
COMP - This pin is the output of the error amplifier.
FB - This pin is the inverting input of error amplifier.
FB2 - There is a switch between FB2 pin and the FB pin.
The switch is closed in single-phase operation and is
opened in two phase operation. The components connecting
to FB2 are to adjust the compensation in single phase
operation to achieve optimum performance.
VDIFF - This pin is the output of the differential amplifier.
VSEN - Remote core voltage sense input.
RTN - Remote core voltage sense return.
DROOP - Output of the droop amplifier. The voltage level on
this pin is the sum of VO and the droop voltage.
DFB - Inverting input to droop amplifier.
VO - An input to the IC that reports the local output voltage.
VSUM - This pin is connected to the summation junction of
channel current sensing.
PHASE2 - The phase node of Phase 2. Connect this pin to
the source of the Channel 2 upper MOSFET.
PGND2 - The return path of the lower gate driver for
Phase 2.
LGATE2 - Lower-side MOSFET gate signal for Phase 2.
PVCC - 5V power supply for gate drivers.
LGATE1 - Lower-side MOSFET gate signal for Phase 1.
PGND1 - The return path of the lower gate driver for
Phase 1.
PHASE1 - The phase node of phase 1. Connect this pin to
the source of the Channel 1 upper MOSFET.
UGATE1 - Upper MOSFET gate signal for Phase 1.
BOOT1 - This pin is the upper-gate-driver supply voltage for
Phase 1. An internal boot strap diode is connected to the
PVCC pin.
VID0, VID1, VID2, VID3, VID4, VID5, VID6 - VID input with
VID0 is the least significant bit (LSB) and VID6 is the most
significant bit (MSB).
VR_ON - Digital enable input. A logic high signal on this pin
enables the regulator.
DPRSLPVR - Deeper sleep enable signal. A logic high
signal on this pin indicates the micro-processor is in
deeper-sleep mode and also indicates a slow C4 entry or
exit rate with 41µA discharging or charging the SOFT
capacitor.
DPRSTP# - Deeper sleep slow wake up signal. A logic low
signal on this pin indicates the micro-processor is in
deeper-sleep mode.
CLK_EN# - Digital output for system clock. Goes active
10µs after VCORE is within 10% of Boot voltage.
3V3 - 3.3V supply voltage for CLK_EN#.
VIN - Battery supply voltage. It is used for input voltage feed
forward to improve input line transient performance.
VSS - Signal ground. Connect to local controller ground.
VDD - 5V control power supply.
ISEN2 - Individual current sharing sensing for Channel 2.
ISEN1 - Individual current sharing sensing for Channel 1.
8
FN6398.4
August 25, 2015
ISL6266, ISL6266A
PGND2
LGATE2
PHASE2
UGATE2
BOOT2
PGND1
LGATE1
PHASE1
UGATE1
BOOT1
VR_TT#
NTC
Functional Block Diagram
6µA
54µA
PVCC
PVCC
+
PVCC
PVCC
VDD
VIN
PVCC
1.2V
VIN
PVCC
1.24V
DRIVER
LOGIC
DRIVER
LOGIC
ULTRASONIC
TIMER
FLT
FLT
ISEN2
CURRENT
BALANCE
ISEN1
VSOFT
I_BALF
VIN
VIN
MODULATOR
MODULATOR
OC
CH1
CH2
Vw
PGOOD
MONITOR
AND LOGIC
FAULT AND
PGOOD
LOGIC
Vw
PHASE
SEQUENCER
PHASE
CONTROL
LOGIC
PGOOD
VO
E/A
VIN
FB
VDIFF
+
+
1
+
-
+
+
1
0.5
RTN
VSUM
OCSET
VO
DROOP
+
10µA
DPRSTP#
DPRSLPVR
PSI#
VR_ON
VID6
VID5
-
MULTIPLIER
MODE CHANGE
REQUEST
SINGLE
PHASE
MODE
CONTROL
VID4
VID3
VID2
PMON
OC
VO
DAC
VID1
SOFT
VSOFT
DACOUT
VID0
FB2
-
SINGLE
PHASE
SOFT
RBIAS
COMP
SINGLE
PHASE
VSEN
VO
-
DROOP
FLT
CH2
+
CH1
DFB
CLK_EN#
OC
VW
3V3
PGOOD
GND
VSOFT
FIGURE 1. SIMPLIFIED FUNCTIONAL BLOCK DIAGRAM OF ISL6266, ISL6266A
9
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Typical Performance Curves
1.16
100
90
1.14
70
VIN = 8.0V
60
VIN = 12.6V
VIN = 8.0V
1.12
VIN = 19.0V
VOUT (V)
EFFICIENCY (%)
80
50
40
VIN = 12.6V
1.10
VIN = 19.0V
1.08
30
20
1.06
10
0
0
5
10
15
20
25
30
35
40
45
1.04
50
0
10
20
IOUT (A)
FIGURE 2. ACTIVE MODE EFFICIENCY, 2-PHASE, CCM,
PSI# = HIGH, VID = 1.15V
40
50
FIGURE 3. ACTIVE MODE LOAD LINE, 2-PHASE, CCM,
PSI# = HIGH, VID = 1.15V
100
1.01
VIN = 8.0V
90
VIN = 12.6V
1.00
80
70
0.99
VIN = 8.0V
60
VIN = 12.6V
50
VOUT (V)
EFFICIENCY (%)
30
IOUT (A)
VIN = 19.0V
40
30
0.98
0.97
VIN = 19.0V
0.96
20
0.95
10
0
0
5
10
15
20
0.94
25
0
5
10
IOUT (A)
FIGURE 4. ACTIVE MODE EFFICIENCY, 1-PHASE, CCM,
PSI# = LOW, VID = 1.00V (ISL6266 ONLY)
25
0.765
90
0.764
80
0.763
70
60
50
40
VIN = 12.6V
VIN = 12.6V
0.762
VIN = 8.0V
VOUT (V)
EFFICIENCY (%)
20
FIGURE 5. ACTIVE MODE LOAD LINE, 1-PHASE, CCM,
PSI# = LOW, VID = 1.00V (ISL6266 ONLY)
100
VIN = 19.0V
30
0.761
0.760
0.759
VIN = 19.0V
20
0.758
10
0
15
IOUT (A)
0.1
1.0
IOUT (A)
FIGURE 6. DEEPER SLEEP MODE EFFICIENCY
10
10.0
0.757
VIN = 8.0V
0
1
2
3
IOUT (A)
FIGURE 7. DEEPER SLEEP MODE LOAD LINE
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Typical Performance Curves
(Continued)
VR_ON
VOUT
VOUT
VSOFT
VR_ON
VSOFT
CSOFT = 15nF
FIGURE 8. SOFT-START WAVEFORM SHOWING SLEW RATE
OF 2.5mV/µs AT VID = 1V, ILOAD = 0A
CSOFT = 15nF
FIGURE 9. SOFT-START WAVEFORM SHOWING SLEW RATE
OF 2.5mV/µs AT VID = 1.4375V, ILOAD = 0A
CLK_EN#
VIN
IMVP-6_PWRGD
IIN
VOUT @ 1.15V
VOUT
FIGURE 10. SOFT-START WAVEFORM SHOWING CLK_EN#
AND IMVP-6 PGOOD
VR_ON
FIGURE 11. 8V TO 20V INPUT LINE TRANSIENT RESPONSE,
CIN = 240µF
DPRSTP#
VOUT
VID6
DPRSLPVR
IIN
VOUT
FIGURE 12. NRUSH CURRENT AT START-UP, VIN = 14.6V,
VID = 1.4375V, ILOAD = 5A
11
FIGURE 13. SLOW C4 EXIT WITH DELAY OF DPRSLPVR,
FROM VID1000000 (0.7V) TO 0110000 (0.9V)
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Typical Performance Curves
(Continued)
VOUT
VOUT
FIGURE 14. LOAD STEP-UP RESPONSE AT THE CPU
SOCKET MPGA479, 35A LOAD STEP @
1000A/µs, 2-PHASE CCM
FIGURE 15. LOAD DUMP RESPONSE AT THE CPU SOCKET
MPGA479, 35A LOAD STEP @ 1000A/µs,
2-PHASE CCM
VID3
VID3
VOUT
VOUT
PHASE1
PHASE1
PHASE2
PHASE2
FIGURE 16. VID3 CHANGE OF 010X000 FROM 1V TO 1.1V
WITH DPRSLPVR = 0, DPRSTP# = 1, PSI# = 1
FIGURE 17. VID3 CHANGE OF 010X000 FROM 1.1V TO 1V
WITH DPRSLPVR = 0, DPRSTP# = 1, PSI# = 1
PSI#
PSI#
VOUT
VOUT
PHASE1
PHASE2
FIGURE 18. 2-CCM TO 1-CCM UPON PSI# ASSERTION WITH
DPRSLPVR = 0, DPRSTP# = 1
12
PHASE1
PHASE2
FIGURE 19. 1-CCM TO 2-CCM UPON PSI# DEASSERTION
WITH DPRSLPVR = 0, DPRSTP# = 1
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Typical Performance Curves
(Continued)
DPRSLPVR
DPRSLPVR/PSI
VOUT
VOUT
PHASE1
PHASE2
FIGURE 20. C4 ENTRY WITH VID CHANGE 0011X00 FROM
1.2V TO 1.15V, ILOAD = 2A, TRANSITION OF
2-CCM TO 1-DCM, PSI# TOGGLE FROM 1 TO 0
WITH DPRSLPVR FROM 0 TO 1
PHASE1
PHASE2
FIGURE 21. VID3 CHANGE OF 010X000 FROM 1V TO 1.1V
WITH DPRSLPVR = 0, DPRSTP# = 1, PSI# = 1
VOUT
DPRSLPVR
VOUT
IMVP-6_PWRGD
PHASE1
PHASE2
IOUT
FIGURE 22. C4 ENTRY WITH VID CHANGE OF 011X011 FROM
0.8625V TO 0.7625V, ILOAD = 3A, 1-CCM TO
1-DCM
FIGURE 23. OVERCURRENT PROTECTION
VID3
IMVP-6_PWRGD
VOUT
VOUT
PMON UNFILTERED
PHASE1
FIGURE 24. 1.7V OVERVOLTAGE PROTECTION SHOWS
OUTPUT VOLTAGE PULLED TO 0.9V AND PWM
TRI-STATE
13
PMON FILTERED
FIGURE 25. VID TRANSITION FROM 1V TO 1.10V ILOAD = 24A,
EXTERNAL FILTER 40k AND 100pF AT PMON
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Typical Performance Curves
(Continued)
VOUT
VOUT
PMON UNFILTERED
PMON UNFILTERED
PMON FILTERED
PMON FILTERED
FIGURE 26. VID = 1.15V, LOAD TRANSIENT OF 0A TO 36A
WITH INTEL VTT TOOL, 1kHz RATE, 50% DUTY
CYCLE, TR = 35
FIGURE 27. VID = 1.15V, LOAD APPLICATION FROM
0A TO 36A WITH INTEL VTT TOOL, 1kHz RATE,
50% DUTY CYCLE, TR = 35
VOUT
PMON UNFILTERED
PMON FILTERED
FIGURE 28. VID = 1.15V, LOAD RELEASE FROM 36A TO 0A WITH INTEL VTT TOOL, 1kHz RATE, 50% DUTY CYCLE, TR = 35
1.8
0.8
1.6
19V, 1.15V, 40A
0.6
1.2
1.0
19V, 1.15V, 30A
19V, 1.15V, 20A
PMON (V)
PMON (V)
1.4
0.8
19V, 1.15V, 10A
0.6
19V, 1.15V, 5A
0.5
0.2
0.1
1
2
3
4
5
CURRENT SOURCING (mA)
6
7
FIGURE 29. POWER MONITOR CURRENT SOURCING
CAPABILITY
14
180
0.3
0.2
0
VID = 1.15V, IOUT = 10A
0.4
0.4
0.0
VID = 1.15V, IOUT = 15A
0.7
7
0.0
0.0
VID = 1.15V, IOUT = 5A
VID = 1.15V, IOUT = 2.5A
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
CURRENT SINKING (mA)
FIGURE 30. POWER MONITOR CURRENT SINKING
CAPABILITY
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Simplified Coupled Inductor Application Circuit for DCR Current Sensing
+5V
R12
+3.3V
VIN
3V3
VDD PVCC VIN
VIN
RBIAS
NTC
R13
VR_TT#
C8
VID<0:6>
ISL6266
C7
VR_TT#
UGATE1
BOOT1
SOFT
C6
VIDs
PHASE1
R8
DPRSTP#
DPRSTP#
VSUM
LGATE1
DPRSLPVR
DPRSLPVR
PSI#
PGND1
PSI#
ISEN1
ISEN1
PMON
CLK_ENABLE#
CLK_EN#
VR_ON
VR_ON
IMVP-6_PWRGD
PGOOD
RL
VIN
CL VO'
C8
R10
VSEN
REMOTE
SENSE
RTN
R2
R3
VDIFF
R1
LO
BOOT2
PHASE2
C3
R7
C1
VO
UGATE2
CO
C5
R11
RL
LGATE2
FB2
FB
VO'
R9
PGND2
CL
VSUM
COMP
ISEN2
C2
RFSET
ISEN2
VSUM
VSUM
VW
OCSET
C9
GND
DFB
DROOP VO
R5
R6
R4
C4
RN
NTC
NETWORK
CCS
VO'
FIGURE 31. ISL6266 BASED TWO-PHASE COUPLED INDUCTOR DESIGN WITH DCR SENSING
15
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Simplified Application Circuit for DCR Current Sensing
+5V
VIN
+3.3V
R12
3V3
VDD PVCC VIN
VIN
RBIAS
NTC
R13
VR_TT#
C8
VID<0:6>
ISL6266A
C7
VR_TT#
UGATE1
BOOT1
SOFT
LO
C6
VIDs
PHASE1
R10
DPRSTP#
DPRSTP#
CL
RL
LGATE1
DPRSLPVR
ISEN1
DPRSLPVR
PSI#
VO'
R8
PGND2
PSI#
VO
VSUM
ISEN1
PMON
CO
CLK_ENABLE#
CLK_EN#
VR_ON
VR_ON
IMVP-6_PWRGD
PGOOD
VIN
C8
VSEN
REMOTE
SENSE
UGATE2
RTN
R2
VDIFF
R3
C1
PHASE2
C3
R7
R1
LO
BOOT2
C5
R11
RL
LGATE2
FB2
FB
R9
PGND2
ISEN2
CL
VO'
VSUM
COMP
ISEN2
C2
RFSET
VSUM
VSUM
VW
OCSET
C9
GND
DFB
DROOP VO
R5
R6
R4
C4
RN
NTC
NETWORK
CCS
VO'
FIGURE 32. ISL6266A BASED TWO-PHASE BUCK CONVERTER WITH INDUCTOR DCR CURRENT SENSING
16
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Simplified Application Circuit for Resistive Current Sensing
+5V
VIN
+3.3V
R11
3V3
VDD PVCC VIN
VIN
RBIAS
ISL6266A
NTC
R12
VR_TT#
C9
VID<0:6>
C7
VR_TT#
UGATE1
BOOT1
SOFT
L
RS
C6
VIDs
PHASE1
R10
DPRSTP#
DPRSTP#
CL
RL
LGATE1
DPRSLPVR
ISEN2
DPRSLPVR
PSI#
VO'
R8
PGND2
PSI#
VO
VSUM
ISEN1
PMON
CO
CLK_ENABLE#
CLK_EN#
VR_ON
VR_ON
IMVP-6_PWRGD
PGOOD
VIN
C8
VSEN
REMOTE
SENSE
UGATE2
RTN
R2
VDIFF
R3
C1
PHASE2
C3
R7
L
BOOT2
RS
C5
R11
RL
LGATE2
FB2
FB
R9
PGND2
R1
ISEN2
CL
VO'
VSUM
COMP
ISEN2
C2
RFSET
VSUM
VSUM
VW
OCSET
C9
GND
DFB
DROOP VO
R5
R6
R4
CHF
C4
VO'
FIGURE 33. ISL6266A BASED TWO-PHASE BUCK CONVERTER WITH RESISTIVE CURRENT SENSING
17
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Theory of Operation
VDD
The ISL6266A is a two-phase regulator implementing Intel®
IMVP-6 protocol and includes embedded gate drivers for
reduced system cost and board area. The regulator provides
optimum steady-state and transient performance for
microprocessor core applications up to 50A. System
efficiency is enhanced by idling one phase at low-current
and implementing automatic DCM-mode operation.
The heart of the ISL6266A is R3 Technology™, Intersil’s
Robust Ripple Regulator modulator. The R3 modulator
combines the best features of fixed frequency PWM and
hysteretic PWM while eliminating many of their
shortcomings. The ISL6266A modulator internally
synthesizes an analog of the inductor ripple current and
uses hysteretic comparators on those signals to establish
PWM pulse widths. Operating on these large-amplitude,
noise-free synthesized signals allows the ISL6266A to
achieve lower output ripple and lower phase jitter than either
conventional hysteretic or fixed frequency PWM controllers.
Unlike conventional hysteretic converters, the ISL6266A has
an error amplifier that allows the controller to maintain a
0.5% voltage regulation accuracy throughout the VID range
from 0.75V to 1.5V.
The hysteresis window voltage is relative to the error
amplifier output such that load current transients results in
increased switching frequency, which gives the R3 regulator
a faster response than conventional fixed frequency PWM
controllers. Transient load current is inherently shared
between active phases due to the use of a common
hysteretic window voltage. Individual average phase
voltages are monitored and controlled to equally share the
static current among the active phases.
10mV/µs
VR_ON
2.8mV/µs
100µs
VBOOT
SOFT AND VO
VID COMMANDED
VOLTAGE
90%
13 SWITCHING CYCLES
CLK_EN#
~7ms
IMVP-6 PGOOD
FIGURE 34. SOFT-START WAVEFORMS USING A 15nF SOFT
CAPACITOR
Static Operation
After the start sequence, the output voltage will be regulated
to the value set by the VID inputs shown in Table 1. The
entire VID table is presented in the intel IMVP-6
specification. The ISL6266A will control the no-load output
voltage to an accuracy of ±0.5% over the range of 0.75V to
1.5V.
TABLE 1. TRUNCATED VID TABLE FOR INTEL IMVP-6+
SPECIFICATION
VOUT
(V)
VID6
VID5
VID4
VID3
VID2
VID1
VID0
0
0
0
0
0
0
0
1.5000
0
0
0
0
0
0
1
1.4875
0
0
0
0
1
0
1
1.4375
0
0
1
0
0
0
1
1.2875
Start-Up Timing
0
0
1
1
1
0
0
1.15
With the controller's VDD voltage above the POR threshold,
the start-up sequence begins when VR_ON exceeds the
3.3V logic HIGH threshold. Approximately 100µs later, SOFT
and VOUT begin ramping to the boot voltage of 1.2V. At
start-up, the regulator always operates in a 2-phase CCM
mode regardless of control signal assertion levels. During
this interval, the SOFT capacitor is charged by 41µA current
source. If the SOFT capacitor is selected to be 20nF, the
SOFT ramp will be at 2mV/µs for a soft-start time of 600µs.
Once VOUT is within 10% of the boot voltage for 13 PWM
cycles (43µs for frequency = 300kHz), then CLK_EN# is
pulled LOW and the SOFT capacitor is charged/discharged
by approximately 200µA. Therefore, VOUT slews at
10mV/µs to the voltage set by the VID pins. Approximately
7ms later, PGOOD is asserted HIGH. Typical start-up timing
is shown in Figure 34.
0
1
1
0
1
0
1
0.8375
0
1
1
1
0
1
1
0.7625
1
1
0
0
0
0
0
0.3000
1
1
1
1
1
1
1
0.0000
18
A fully-differential amplifier implements core voltage sensing
for precise voltage control at the microprocessor die. The
inputs to the amplifier are the VSEN and RTN pins.
As the load current increases from zero, the output voltage
will droop from the VID table value by an amount
proportional to current to achieve the IMVP-6+ load line. The
ISL6266A provides options for current to be measured using
either resistors in series with the channel inductors as shown
in the application circuit of Figure 33, or using the intrinsic
series resistance of the inductors as shown in the application
circuit of Figure 32. In both cases, signals representing the
inductor currents are summed at VSUM, which is the
non-inverting input to the DROOP amplifier shown in the
block diagram of Figure 1. The voltage at the DROOP pin
FN6398.4
August 25, 2015
ISL6266, ISL6266A
minus the output voltage, VO´, is a high-bandwidth analog of
the total inductor current. This voltage is used as an input to
a differential amplifier to achieve the IMVP-6+ load line, and
also as the input to the overcurrent protection circuit.
When using inductor DCR current sensing, a single NTC
element is used to compensate the positive temperature
coefficient of the copper winding thus maintaining the
load-line accuracy.
In addition to monitoring the total current (used for DROOP
and overcurrent protection), the individual channel average
currents are also monitored and used for balancing the load
between channels. The IBAL circuit will adjust the channel
pulse-widths up or down relative to the other channel to
cause the voltages presented at the ISEN pins to be equal.
The ISL6266A controller can be configured for two-channel
operation, with the channels operating 180° apart. The
channel PWM frequency is determined by the value of
RFSET connected to pin VW as shown in Figures 32 and 33.
Input and output ripple frequencies will be the channel PWM
frequency multiplied by the number of active channels.
High Efficiency Operation Mode
The ISL6266A has several operating modes to optimize
efficiency. The controller's operational modes are designed
to work in conjunction with the Intel IMVP-6+ control signals
to maintain the optimal system configuration for all IMVP-6+
conditions. These operating modes are established by the
IMVP-6+ control signal inputs PSI#, DPRSLPVR, and
DPRSTP# as shown in Table 2. At high current levels, the
system will operate with both phases fully active, responding
rapidly to transients and delivering maximum power to the
load. At reduced load-current levels, one of the phases may
be idled. This configuration will minimize switching losses,
while still maintaining transient response capability. At the
lowest current levels, the controller automatically configures
the system to operate in single-phase automatic-DCM
mode, thus achieving the highest possible efficiency. In this
mode of operation, the lower MOSFET will be configured to
automatically detect and prevent discharge current flowing
from the output capacitor through the inductors, and the
switching frequency will be proportionately reduced, thus
greatly reducing both conduction and switching losses.
Smooth mode transitions are facilitated by the R3
Technology™, which correctly maintains the internally
synthesized ripple currents throughout mode transitions. The
controller is thus able to deliver the appropriate current to the
load throughout mode transitions. The controller contains
embedded mode-transition algorithms that maintain
voltage-regulation for all control signal input sequences and
durations.
While the ISL6266A will respond according to the logic
states shown in Table 2, it can deviate from the commanded
state during sleep state exit. If the core voltage is directed by
the CPU to make a VID change that causes excessive
output capacitor inrush current when going from 1-phase
DCM to 1-phase CCM, the controller will automatically add
Phase 2 until the VID transition is complete. This is
beneficial for designs that have very large COUT values.
The controller contains internal counters that prevent
spurious control signal glitches from resulting in unwanted
mode transitions. Control signals of less than two switching
periods do not result in phase-idling.
TABLE 2. CONTROL SIGNAL TRUTH TABLES FOR OPERATION MODES OF ISL6266 AND ISL6266A
DPRSLPVR
DPRSTP#
PSI#
0
0
0
0
0
0
ISL6266
ISL6266A
VID SLEW RATE
CPU MODE
1-phase CCM
1-phase diode emulation
fast
awake
1
2-phase CCM
2-phase CCM
fast
awake
1
0
1-phase CCM
1-phase diode emulation
fast
awake
0
1
1
2-phase CCM
2-phase CCM
fast
awake
1
0
0
1-phase diode emulation
1-phase diode emulation
slow (Note 5)
sleep
1
0
1
1-phase diode emulation
1-phase diode emulation
slow (Note 5)
sleep
1
1
0
1-phase CCM
1-phase diode emulation
slow
awake
1
1
1
2-phase CCM
2-phase CCM
slow
awake
NOTE:
5. The negative VID slew rate when DPRSTP# = 0 and DPRSLPVR = 1 is limited to no faster than the slow slew rate. However, slower slew rates
can be seen. To conserve power, the ISL6266A will tri-state UGATE and LGATE and let the load gradually pull the core voltage back into
regulation.
19
FN6398.4
August 25, 2015
ISL6266, ISL6266A
While transitioning to single-phase operation, the controller
smoothly transitions current from the idling-phase to the activephase, and detects the idling-phase zero-current condition.
During transitions into automatic-DCM or forced-CCM mode,
the timing is carefully adjusted to eliminate output voltage
excursions. When a phase is added, the current balance
between phases is quickly restored.
When commanded into 1-phase CCM operation according
to Table 2, both MOSFETs of Phase 2 will be off. The
controller will thus eliminate switching losses associated with
the unneeded channel.
VOUT AND VSOFT
Dynamic Operation
Figure 35 shows that the ISL6266A responds to changes in
VID command voltage by slewing to new voltages with a
dV/dt set by the SOFT capacitor and by the state of
DPRSLPVR. With CSOFT = 15nF and DPRSLPVR HIGH,
the output voltage will move at ±2.8mV/µs for large changes
in voltage. For DPRSLPVR LOW, the large signal dV/dt will
be ±10mV/µs. As the output voltage approaches the VID
command value, the dV/dt moderates to prevent overshoot.
10mV/µs
-2.5mV/µs
The ISL6266A can be configured to operate as a single
phase regulator using the same layout as a two phase
design to accommodate lower power CPUs. To accomplish
this, the designer must connect ISEN1 and ISEN2 to
VCC_PRM (reference AN1376 for signal names). Channel 2
components can be removed as well as current balance
circuitry. The ISL6266A will power-up and regulate in DCM
or CCM based on the state of PSI#, as outlined in Table 2.
The OCP threshold will also change based on the state of
PSI#, as outlined in “Protection” on page 20.
2.5mV/µs
DPRSLPVR
Keeping DPRSLPVR HIGH for voltage transitions into and
out of Deeper Sleep will result in low dV/dt output voltage
changes with resulting minimized audio noise. For fastest
recovery from Deeper Sleep to Active mode, holding
DPRSLPVR LOW results in maximum dV/dt. Therefore, the
ISL6266A is IMVP-6+ compliant for DPRSTP# and
DPRSLPVR logic.
VID#
FIGURE 35. DEEPER SLEEP TRANSITION SHOWING
DPRSLPVR'S EFFECT ON EXIT SLEW RATE
When commanded to single-phase DCM mode, both
MOSFETs associated with Phase 2 are off, and the
ISL6266A turns off the lower MOSFET of Channel 1
whenever the Channel 1 current decays to zero. As load is
further reduced, the Phase 1 channel switching frequency
decreases to maintain high efficiency. The operation of the
inactive for 1-phase DCM and 1-phase CCM described
previously refers to the ISL6266A only. See “ISL6266
Features” on page 21 for information on the ISL6266.
Intersil's R3 Technology™ has intrinsic voltage feedforward.
As a result, high-speed input voltage steps do not result in
significant output voltage perturbations. In response to load
current step increases, the ISL6266A will transiently raise
the switching frequency so that response time is decreased
and current is shared by two channels.
Protection
The ISL6266A provides overcurrent, overvoltage,
undervoltage protection and over-temperature protection, as
shown in Table 3.
TABLE 3. FAULT-PROTECTION SUMMARY OF ISL6266, ISL6266A
FAULT DURATION PRIOR
TO PROTECTION
PROTECTION ACTIONS
FAULT RESET
Overcurrent fault
120µs
PWM1, PWM2 three-state, PGOOD latched low
VR_ON toggle or VDD toggle
Way-Overcurrent fault
<2µs
PWM1, PWM2 three-state, PGOOD latched low
VR_ON toggle or VDD toggle
Overvoltage fault (1.7V)
Immediately
Low-side MOSFET on until VCORE <0.85V, then PWM VDD toggle
three-state, PGOOD latched low (0V to 1.7V always)
Overvoltage fault (+200mV)
1ms
PWM1, PWM2 three-state, PGOOD latched low
VR_ON toggle or VDD toggle
Undervoltage fault
(-300mV)
1ms
PWM1, PWM2 three-state, PGOOD latched low
VR_ON toggle or VDD toggle
Current imbalance fault
(7.5mV)
1ms
PWM1, PWM2 three-state, PGOOD latched low
VR_ON toggle or VDD toggle
Over-temperature fault
(NTC <1.18V)
Immediately
VR_TT# goes low
N/A
20
FN6398.4
August 25, 2015
ISL6266, ISL6266A
The ISL6266A has a thermal throttling feature. If the voltage
on the NTC pin goes below the 1.2V over-temperature
threshold, the VR_TT# pin is pulled low indicating the need
for thermal throttling to the system oversight processor. No
other action is taken within the ISL6266A in response to
NTC pin voltage.
Overcurrent protection is tied to the voltage droop, which is
determined by the resistors selected as described in
“Component Selection and Application” on page 22. After
the load-line is set, the OCSET resistor can be selected to
detect overcurrent at any level of droop voltage. An
overcurrent fault will occur when the load current exceeds
the overcurrent setpoint voltage while the regulator is in a
2-phase mode. While the regulator is in a 1-phase mode of
operation, the overcurrent setpoint is automatically reduced
to 50% of two-phase overcurrent level, and the fast-trip
way-overcurrent set point is reduced to 66%. For
overcurrents less than 2.5 times the OCSET level, the overload condition must exist for 120µs in order to trip the OC
fault latch. This is shown in Figure 25.
Power Monitor
For over-loads exceeding 2.5 times the set level, the PWM
outputs will immediately shut off and PGOOD goes low to
maximize protection due to hard shorts.
V PMON = V CCSENSE   V DROOP – V O   17.5
In addition, excessive phase imbalance (for example, due to
gate driver failure) will be detected in two-phase operation
and the controller will be shut-down 1ms after detection of
the excessive phase current imbalance. The phase
imbalance is detected by the voltage on the ISEN pins if the
difference is greater than 9mV.
Undervoltage protection is independent of the overcurrent
limit. If the output voltage is less than the VID set value by
300mV or more, a fault will latch after 1ms in that condition,
turning the PWM outputs off and pulling PGOOD to ground.
Note that most practical core regulators will have the
overcurrent set to trip before the -300mV undervoltage limit.
There are two levels of overvoltage protection and response.
1. For output voltage exceeding the set value by +200mV
for 1ms, a fault is declared. All of the above faults have
the same action taken: PGOOD is latched low and the
upper and lower power MOSFETs are turned off so that
inductor current will decay through the MOSFET(s) body
diode(s). This condition can be reset by bringing VR_ON
low or by bringing VDD below 4V. When these inputs are
returned to their high operating levels, a soft-start will
occur.
2. The second level of overvoltage protection behaves
differently (see Figure 26). If the output exceeds 1.7V, an
OV fault is immediately declared, PGOOD is latched low
and the low-side MOSFETs are turned on. The low-side
MOSFETs will remain on until the output voltage is pulled
down below about 0.85V, at which time all MOSFETs are
turned off. If the output again rises above 1.7V, the
protection process is repeated. This offers the maximum
amount of protection against a shorted high-side
MOSFET while preventing output ringing below ground.
The 1.7V OV is not reset with VR_ON, but requires that
VDD be lowered to reset. The 1.7V OV detector is active
at all times that the controller is enabled including after
one of the other faults occurs so that the processor is
protected against high-side MOSFET leakage while the
MOSFETs are commanded off.
21
The power monitor signal is an analog output. Its magnitude
is proportional to the product of VCCSENSE and the voltage
difference between Vdroop and VO, which is the
programmed voltage droop value, equal to load current
multiplied by the load line impedance (for example 2.1m).
The output voltage of the PMON pin in two-phase design is
given by Equation 1:
(EQ. 1)
Equation 1 can be expressed in terms of load current as
seen in Equation 2:
V PMON =  V CCSENSE  I CORE   2.1m  17.5
(EQ. 2)
The power consumed by the CPU can be calculated by
Equation 3:
P CPU = V PMON   17.5  0.0021    WATT 
(EQ. 3)
where 0.0021 is the typical load line slope. The power
monitor load regulation is approximately 7. Within its
sourcing/sinking current capability range, when the power
monitor loading changes to 1mA, the output of the power
monitor will change to 7mV. The 7 impedance is
associated with the layout and package resistance of PMON
inside the IC. In practical applications, compared to the load
resistance on the PMON pin, 7 output impedance
contributes no significant error.
ISL6266 Features
The ISL6266 incorporates all the features previously listed
for the ISL6266A. However, the sleep state logic is slightly
altered (see Table 2). In addition to those differences, the
ISL6266 has been optimized to work with coupled-inductor
solutions. Due to mutual magnetic fields between the
individual phase windings of the coupled-inductor, the
effective per-phase inductance equals the leakage
inductance of the transformer. This can be very low (e.g.
90nH), which allows for faster channel current slew rates
and, consequently, an all-ceramic output capacitor bank can
be utilized. Additionally, the current ripple is lower than would
be produced with two discrete inductors of equivalent value
to the coupled-inductor leakage. This improves
coupled-inductor efficiency over discrete inductor solutions
for the same transient response.
In single phase operation, the active channel inductor will
continue to build a mutual field in the inactive channel inductor.
This field must be dissipated every cycle to maintain inductor
FN6398.4
August 25, 2015
ISL6266, ISL6266A
volt-second balance. The ISL6266 continues to turn on the
lower MOSFET for the inactive channel to deplete the induced
field with minimum power loss.
Component Selection and Application
Soft-Start and Mode Change Slew Rates
The ISL6266A uses two slew rates for various modes of
operation. The first is a slow slew rate used to reduce in-rush
current during start-up. It is also used to reduce audible noise
when entering or exiting Deeper Sleep Mode. A faster slew rate
is used to exit out of Deeper Sleep and to enhance system
performance by achieving active mode regulation more quickly.
Note that the SOFT capacitor current is bidirectional. The
current is flowing into the SOFT capacitor when the output
voltage is commanded to rise and out of the SOFT capacitor
when the output voltage is commanded to fall.
Figure 36 illustrates how the two slew rates are determined
by commanding one of two current sources into or out of the
SOFT pin. The capacitor from the SOFT pin to ground holds
the voltage commanded by the two current sources. The
voltage is fed into the non-inverting input of the error
amplifier and sets the regulated system voltage. Depending
on the state of the system (Start-Up or Active mode) and the
state of the DPRSLPVR pin, one of the two currents shown
in Figure 36 will be used to charge or discharge this
capacitor, thereby controlling the slew rate of the newly
commanded voltage. These currents can be found under
“SOFT-START CURRENT” on page 4 of the “Electrical
Specifications” table.
ISL6266, ISL6266A
ISS
I2
ERROR
AMPLIFIER
+
SOFT
+
CSOFT
VREF
FIGURE 36. SOFT PIN CURRENT SOURCES FOR FAST AND
SLOW SLEW RATES
The first current, labeled ISS, is given in the “Electrical
Specifications” table on page 4 as 42µA. This current is used
during soft-start. The second current (I2) sums with ISS to
get the larger of the two currents, labeled IGV in the
“Electrical Specifications” table on page 4. This total current
is typically 205µA with a minimum of 180µA.
22
The IMVP-6+ specification dictates the critical timing
associated with regulating the output voltage. The symbol,
SLEWRATE, as given in the IMVP-6+ specification will
determine the choice of the SOFT capacitor (CSOFT) by
Equation 4.
I GV
C SOFT = -----------------------------------SLEWRATE
(EQ. 4)
Using a SLEWRATE of 10mV/µs and the typical IGV value
given in the “Electrical Specifications” table on page 4 of
205µA, CSOFT is as shown in Equation 5.
(EQ. 5)
C SOFT = 205A   10mV  1s 
A choice of 0.015µF would guarantee a SLEWRATE of
10mV/µs is met for the minimum IGV value given in the
“Electrical Specifications” table on page 4. This choice of
CSOFT will then control the start-up slewrate as well. One
should expect the output voltage to slew to the boot value of
1.2V at a rate given by Equation 6.
I SS
41A
dV
------- = ------------------= ----------------------- = 2.8mV  s
0.015F
C SOFT
dt
(EQ. 6)
Selecting RBIAS
To properly bias the ISL6266A, a reference current is
established by placing a 147k, 1% tolerance resistor from
the RBIAS pin to ground. This will provide a highly accurate
10µA current source from which the OCSET reference
current can be derived.
Care should be taken in layout that the resistor is placed
very close to the RBIAS pin and that a good quality signal
ground is connected to the opposite side of the RBIAS
resistor. Do not connect any other components to this pin as
this would negatively impact performance. Capacitance on
this pin would create instabilities and should be avoided.
Start-Up Operation - CLK_EN# and PGOOD
The ISL6266A provides a 3.3V logic output pin for
CLK_EN#. The 3V3 pin allows for a system 3.3V source to
be connected to separated circuitry inside the ISL6266A,
solely devoted to the CLK_EN# function. The output is a
3.3V CMOS signal with 4mA sourcing and sinking capability.
This implementation removes the need for an external
pull-up resistor on this pin, thereby removing a leakage path
from the 3.3V supply to ground when the logic state is low.
The lack of superfluous current leakage paths serves to
prolong battery life. For noise immunity, the 3.3V supply
should be decoupled to digital ground rather than to analog
ground.
As mentioned in “Theory of Operation” on page 18,
CLK_EN# is logic level high at start-up until approximately
43µs after the VCC_CORE is in regulation at the Boot level.
Afterwards, CLK_EN# transitions low, triggering an internal
timer for the IMVP6_PWRGD signal. When the timer
reaches 6.8ms, IMVP-6_PWRGD will toggle high.
FN6398.4
August 25, 2015
ISL6266, ISL6266A
ISEN1
ISEN2
ISEN2
ISEN1
10µA
OCSET
+
OC
ROCSET
VO'
VSUM
+
DROOP
INTERNAL TO
ISL6266
+
+
-
VSUM
DFB
Rdrp2
Vdcr1
DCR
RL1
Cn
IPHASE2
RPAR
RS
VSUM
VSEN
ISEN1
L2
RL2
RNTC
VO'
VDIFF
+
C L1
RO1
RSERIES
+
1 RTN
L1
RS
VSUM
DROOP
+
1 -
IPHASE1
Rdrp1
ISEN2
VO'
VO'
DCR
+
Vdcr2
VOUT
RO2
CBULK
CL2
VO'
82nF
10
Ropn1
0.018µF
0.018µF
VCC_SENSE
VSS_SENSE
ROPN2
ESR
TO VOUT
TO PROCESSOR
SOCKET KELVIN
CONNECTIONS
FIGURE 37. SIMPLIFIED SCHEMATIC FOR DROOP AND DIE SENSING WITH INDUCTOR DCR CURRENT SENSING
Static Mode of Operation - Processor Die Sensing
Die sensing is the ability of the controller to regulate the core
output voltage at a remotely sensed point. This allows the
voltage regulator to compensate for various resistive drops
in the power path and ensure that the voltage seen at the
CPU die is the correct level independent of load current.
The VSEN and RTN pins of the ISL6266A are connected to
Kelvin sense leads at the die of the processor through the
processor socket. These signal names are VCC_SENSE and
VSS_SENSE respectively. This allows the voltage regulator to
tightly control the processor voltage at the die, independent
of layout inconsistencies and voltage drops. This Kelvin
sense technique provides for extremely tight load line
regulation.
These traces should be treated as noise sensitive traces.
For optimum load line regulation performance, the traces
connecting these two pins to the Kelvin sense leads of the
processor must be laid out away from rapidly rising/falling
voltage nodes (switching nodes) and other noisy traces. To
achieve optimum performance, place common mode and
differential mode RC filters to analog ground on VSEN and
RTN as shown in Figure 37. The filter resistors should be
10 so that they do not interact with the 50k input
resistance of the differential amplifier. The filter resistor may
be inserted between VCC_SENSE and the VSEN pin.
Another option is to place to the filter resistor between
Vcc_sense and VSEN pin and between VSS_SENSE and
RTN pin. The need for RC filters really depends on the
actual board layout and noise environment.
23
Intersil recommends the use of the ROPN1 and ROPN2
connected to VOUT and ground as shown in Figure 37.
These resistors provide voltage feedback in the event that
the system is powered up without a processor installed.
These resistors typically range from 20 to 100.
Setting the Switching Frequency - FSET
The R3 modulator scheme is not a fixed frequency PWM
architecture. The switching frequency can increase during
the application of a load to improve transient performance.
It also varies slightly due to changes in input and output
voltage and output current, but this variation is normally less
than 10% in continuous conduction mode.
See Figure 32. The resistor connected between the VW and
COMP pins of the ISL6266A adjusts the switching window,
and therefore adjusts the switching frequency. The RFSET
resistor that sets up the switching frequency of the converter
operating in CCM can be determined using Equation 7,
where RFSET is in k and the switching frequency is in kHz.
F SW  kHz  – 1.1202
R FSET  k  =  -----------------------------
 2232 
(EQ. 7)
Equation 7 is only a rough estimate of actual frequency. It
should be used to choose an RFSET value in the vicinity of
the desired switching frequency. Empirical fine tuning may
be necessary to achieve the actual frequency target. In
addition, droop amplifier gain may slightly affect the
switching frequency. Equation 7 is derived using the droop
gain seen on the ISL6266AEVAL1Z REV A evaluation
board.
FN6398.4
August 25, 2015
ISL6266, ISL6266A
For 300kHz operation, RFSET is suggested to be 9.53kIn
discontinuous conduction mode (DCM), the ISL6266A runs
in period stretching mode. The switching frequency is
dependent on the load current level. In general, lighter loads
will produce lower switching frequencies. Therefore,
switching loss is greatly reduced for light load operation,
which conserves battery power in portable applications.
Voltage Regulator Thermal Throttling
Proper selection and placement of the NTC thermistor
allows for detection of a designated temperature rise by the
system.
Figure 38 shows the thermal throttling feature with
hysteresis. At low temperature, SW1 is on and SW2
connects to the 1.2V side. The total current going into NTC
pin is 60µA. The voltage on the NTC pin is higher than the
threshold voltage of 1.2V and the comparator output is low.
VR_TT# is pulled high by the external resistor.
6µA
VR_TT#
SW1
NTC
+
VNTC
-
Rs
1.24V
LOGIC_0
T2
SW2
INTERNAL TO
ISL6266
FIGURE 38. CIRCUITRY ASSOCIATED WITH THE THERMAL
THROTTLING FEATURE IN ISL6266
When the temperature increases, the NTC resistor value
decreases, thus reducing the voltage on the NTC pin. When
the voltage decreases to a level lower than 1.2V, the
comparator output changes polarity and turns SW1 off and
connects SW2 to 1.24V. This pulls VR_TT# low and sends
the signal to start thermal throttle. There is a 6µA current
reduction on the NTC pin and 20mV voltage increase on the
threshold voltage of the comparator in this state. The
VR_TT# signal will be used to change the CPU operation
and decrease the power consumption. Temperature will
decrease over time and the NTC thermistor voltage will go
up. When the NTC pin voltage achieves 1.24V, the
comparator output will resume its original state. This
temperature hysteresis feature of VR_TT# is illustrated in
24
T (°C)
Usually, the NTC thermistor's resistance can be
approximated by Equation 8.
R NTC  T  = R NTCTo  e
1
1
b   -------------------- – -----------------------
 T + 273 To + 273
(EQ. 8)
T is the temperature of the NTC thermistor and b is a
parameter constant depending on the thermistor material.
To is the reference temperature in which the approximation
is derived. The most common temperature for To is +25°C.
For example, there are commercial NTC thermistor products
with b = 2750k, b = 2600k, b = 4500k or b = 4250k.
From the operation principle of the VR_TT# circuit
explained, the NTC resistor satisfies Equations 9 through 13:
1.24V
R NTC  T 2  + R S = ---------------- = 22.96k
54A
1.20V
T1
FIGURE 39. TEMPERATURE HYSTERESIS OF VR_TT#
1.2V
R NTC  T 1  + R S = --------------- = 20k
60A
+
RNTC
VR_TT#
LOGIC_1
lntel® IMVP-6+ technology supports thermal throttling of the
processor to prevent catastrophic thermal damage to the
voltage regulator. The ISL6266A features a thermal monitor
that senses the voltage change across an externally placed
negative temperature coefficient (NTC) thermistor.
54µA
Figure 39. T1 represents the higher temperature point at
which the VR_TT# goes from low to high due to the system
temperature rise. T2 represents the lower temperature point
at which the VR_TT# goes high from low because the
system temperature decreases to acceptable levels.
(EQ. 9)
(EQ. 10)
From Equation 9 and Equation 10, Equation 11 can be
derived:
R NTC  T 2  – R NTC  T 1  = 2.96k
(EQ. 11)
Using Equation 8 into Equation 11, the required nominal
NTC resistor value can be obtained by Equation 12:
1
b   -----------------------
 T + 273
o
2.96k  e
R NTCTo = -----------------------------------------------------------------------------e
1
b   -----------------------
T 2 + 273
–e
1
b   -----------------------
T 1 + 273
(EQ. 12)
For those cases where the constant b is not accurate
enough to approximate the resistor value, the manufacturer
provides the resistor ratio information at different
temperatures. The nominal NTC resistor value may be
expressed in another way shown in Equation 13.
2.96k
R NTCTo = -----------------------------------------------------------------------
– 
R NTC  T  R NTC  T 1 
(EQ. 13)
2
FN6398.4
August 25, 2015
ISL6266, ISL6266A

The closest standard resistor to this result is 4.42kThe NTC
resistance at T2 is given by Equation 18.
where R NTC  T  is the normalized NTC resistance to its
nominal value. Most data sheets of the NTC thermistor give
the normalized resistor value based on its value at +25°C.
R NTC_T2 = 2.96k + R NTC_T1 = 18.16k
Once the NTC thermistor resistor is determined, the series
resistor can be derived by Equation 14:
1.2V
R S = --------------- – R NTC  T1  = 20k – R NTC_T
60A
1
Therefore, the NTC branch is designed to have a 470k
NTC and 4.42k resistor in series. The part number of the
NTC thermistor is ERTJ0EV474J in an 0402 package. The
NTC thermistor should be placed in the spot that provides
the best indication of the voltage regulator circuit
temperature.
(EQ. 14)
Once RNTCTo and Rs is designed, the actual NTC resistance
at T2 and the actual T2 temperature can be found in
Equations 15 and 16:
R NTC_T
2
= 2.96k + R NTC_T
Static Mode of Operation - Static Droop Using DCR
Sensing
(EQ. 15)
1
1
T 2_actual = ----------------------------------------------------------------------------------- – 273
R NTC_T

1 
--- ln  -------------------------2 + 1   273 + To 
b  R NTCTo 
(EQ. 18)
As previously mentioned, the ISL6266A has a differential
amplifier that provides precision voltage monitoring at the
processor die for both single-phase and two-phase
operation. This enables the ISL6266A to achieve an
accurate load line in accordance with the IMVP-6+
specification.
(EQ. 16)
For example, if using Equations 12, 13 and 14 to design a
thermal throttling circuit with the temperature hysteresis
+100°C to +105°C, since T1 = +105°C and T2 = +100°C,
and if we use a Panasonic NTC with b = 4700, Equation 12
gives the required NTC nominal resistance as
RNTC_To = 459k.
DESIGN EXAMPLE
The process of compensation for DCR resistance variation
to achieve the desired load line droop has several steps and
may be iterative.
In fact, the data sheet gives the resistor ratio value at
+100°C to +105°C, which is 0.03956 and 0.03322
respectively. The b value 4700k in the Panasonic data
sheet only covers to +85°C. Therefore, using Equation 13 is
more accurate for +100°C design, the required NTC nominal
resistance at +25°C is 467k. The closest NTC resistor
value from the manufacturer is 467k. The series resistance
is given by Equation 17 as follows:
A two-phase solution using DCR sensing is shown in Figure 37.
There are two resistors connecting to the terminals of inductor
of each phase. These are labeled RS and RO. These resistors
are used to obtain the DC voltage drop across each inductor.
The DC current flowing through each inductor will create a DC
voltage drop across the real winding resistance (DCR). This
voltage is proportional to the average inductor current by Ohm’s
Law. When this voltage is summed with the other channel’s DC
voltage, the total DC load current can be derived.
R S = 20k – R NTC_105C = 20k – 15.65k = 4.35k
(EQ. 17)
10µA
OCSET
+
OC
VSUM
+
DROOP
-
INTERNAL TO
ISL6266
VDIFF
DROOP
+
1 -
+
+
1 -
RTN VSEN
VO'
Cn
Rdrp1
+
DFB
Rdrp2
+
VSUM
RS
RS EQV = -------2
DCR
Vdcr EQV = I OUT  ------------2
+
-
VN
-
 R ntc + R series   R par
Rn = -------------------------------------------------------------- R ntc + R series  + R par
VO'
RO
RO EQV = --------2
FIGURE 40. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DCR SENSING
25
FN6398.4
August 25, 2015
ISL6266, ISL6266A
RO is typically 1 to 10. This resistor is used to tie the
outputs of all channels together and thus create a summed
average of the local CORE voltage output. RS is determined
through an understanding of both the DC and transient load
currents. This value will be covered in the next section.
However, it is important to keep in mind that the outputs of
each of these RS resistors are tied together to create the
VSUM voltage node. With both the outputs of RO and RS
tied together, the simplified model for the droop circuit can
be derived. This is presented in Figure 40.
The non-inverting droop amplifier circuit has the gain
Kdroopamp expressed as Equation 25:
Figure 40 shows the simplified model of the droop circuitry.
Essentially, one resistor can replace the RO resistors of each
phase and one RS resistor can replace the RS resistors of
each phase. The total DCR drop due to load current can be
replaced by a DC source, the value of which is given by
Equation 19:
For the G1target = 0.76:
Rntc = 10k with b = 4300,
Rseries = 2610, and
Rpar = 11k
I OUT  DCR
V DCR_EQU = --------------------------------2
(EQ. 19)
For the convenience of analysis, the NTC network
comprised of Rntc, Rseries and Rpar, given in Figure 37, is
labeled as a single resistor RN in Figure 40.
The first step in droop load line compensation is to adjust
RN, ROEQV and RSEQV such that sufficient droop voltage
exists even at light loads between the VSUM and VO' nodes.
As a rule of thumb, we start with the voltage drop across the
RN network, Vn, to be 0.5x to 0.8x VDCR_EQU. This ratio
provides for a fairly reasonable amount of light load signal
from which to arrive at droop.
The resultant NTC network resistor value is dependent on
the temperature and given by Equation 20.
 R series + R ntc   R par
R n  T  = -------------------------------------------------------------R series + R ntc + R par
(EQ. 20)
For simplicity, the gain of Vn to the VDCR_EQU is defined by
G1, also dependent on the temperature of the NTC
thermistor.
R drp2
k droopamp = 1 + ---------------R drp1
(EQ. 25)
G1target is the desired gain of Vn over IOUT •DCR/2.
Therefore, the temperature characteristics of gain of Vn is
described by Equation 26.
G 1t arg et
G 1  T  = ------------------------------------------------------ 1 + 0.00393*(T-25) 
(EQ. 26)
RSEQV = 1825 generates a desired G1, close to the
feature specified in Equation 26.
The actual G1 at +25°C is 0.769. A design file is available to
generate the proper values of Rntc, Rseries, Rpar, and
RSEQV for values of the NTC thermistor and G1 that differ
from the example provided here.
The individual resistors from each phase to the VSUM node,
labeled RS1 and RS2 in Figure 37, are then given by
Equation 27.
Rs = 2  RS EQV
(EQ. 27)
So, RS = 3650. Once we know the attenuation of the RS
and RN network, we can then determine the droop amplifier
gain required to achieve the load line. Setting
Rdrp1 = 1k_1%, then Rdrp2 can be found using Equation 28.
2  R droop
R drp2 =  ----------------------------------------------- – 1  R drp1
 DCR  G1  25C 

(EQ. 28)
Droop Impedance (Rdroop) = 0.0021 (V/A) as per the Intel
IMVP-6+ specification. Using DCR = 0.0008 typical for a
0.36µH inductor, Rdrp1 = 1k and the attenuation gain
(G1) = 0.77, Rdrp2 is then given by Equation 29:

Rn  T 
G 1  T  = ------------------------------------------R n  T  + RS EQV
(EQ. 21)
2  R droop
R drp2 =  --------------------------------------- – 1  1k  5.82k
 0.0008  0.769

DCR  T  = DCR 25C   1 + 0.00393*(T-25) 
(EQ. 22)
Note, we choose to ignore the RO resistors because they do
not add significant error.
Therefore, the output of the droop amplifier divided by the
total load current can be expressed as shown in
Equation 23, where Rdroop is the realized load line slope
and 0.00393 is the temperature coefficient of the copper.
DCR 25
R droop = G 1  T   -------------------   1 + 0.00393*(T-25)   k droopamp
2
(EQ. 23)
How to achieve the droop value independent of the inductor
temperature is expressed by Equation 24.
G 1  T    1 + 0.00393*(T-25)   G 1t arg et
26
(EQ. 24)
(EQ. 29)
These designed values in Rn network are very sensitive to
the layout and coupling factor of the NTC to the inductor. As
only one NTC is required in this application, this NTC should
be placed as close to the Channel 1 inductor as possible and
PCB traces sensing the inductor voltage should route
directly to the inductor pads.
Due to layout parasitics, small adjustments may be
necessary to accurately achieve the full load droop voltage.
This can be easily accomplished by allowing the system to
achieve thermal equilibrium at full load, and then adjusting
Rdrp2 to obtain the appropriate load line slope.
FN6398.4
August 25, 2015
ISL6266, ISL6266A
To see whether the NTC has compensated the temperature
change of the DCR, the user can apply full load current and
wait for the thermal steady state and see how much the
output voltage will deviate from the initial voltage reading. A
good compensation can limit the drift to 2mV. If the output
voltage is decreasing with temperature increase, the ratio
between the NTC thermistor value and the rest of the
resistor divider network has to be increased. The user is
strongly encouraged to use the evaluation board values and
layout to minimize engineering time.
The 2.1mV/A load line should be adjusted by Rdrp2 based
on maximum current. The droop gain might vary slightly
between small steps (e.g. 10A). For example, if the max
current is 40A and the load line 2.1mthe user load the
converter to 40A and look for 84mV of droop. If the droop
voltage is less than 84mV (e.g. 80mV) the new value will be
calculated by Equation 30:
84mV
R drp2 new = ----------------  R drp1 + R drp2  – R drp1
80mV
(EQ. 30)
For the best accuracy, the effective resistance on the DFB
and VSUM pins should be identical so that the bias current
of the droop amplifier does not cause an offset voltage. In
the previous example, the resistance on the DFB pin is
Rdrp1 in parallel with Rdrp2, that is, 1k in parallel with
5.82k or 853. The resistance on the VSUM pin is Rn in
parallel with RSEQV or 5.87k in parallel with 1.825k,
which equals 1392. The mismatch in the effective
resistances is 1404 - 53 = 551. The mismatch cannot be
larger than 600. To reduce the mismatch, multiply both
Rdrp1 and Rdrp2 by the appropriate factor. The appropriate
factor in this example is 1404/853 = 1.65. In summary, the
predicted load line with the designed droop network
parameters based on the Intersil design tool is shown in
Figure 41.
LOAD LINE (mV/A)
2.25
2.20
2.15
2.10
2.05
0
20
40
60
80
100
INDUCTOR TEMPERATURE (°C)
FIGURE 41. LOAD LINE PERFORMANCE WITH NTC
THERMAL COMPENSATION
Dynamic Mode of Operation - Dynamic Droop
Using DCR Sensing
Droop is very important for load transient performance. If the
system is not compensated correctly, the output voltage
could sag excessively upon load application and potentially
27
create a system failure. The output voltage could also take a
long period of time to settle to its final value, which could be
problematic if a load dump were to occur during this time.
This situation would cause the output voltage to rise above
the no load setpoint of the converter and could potentially
damage the CPU.
The L/DCR time constant of the inductor must be matched to
the Rn*Cn time constant as shown in Equation 31.
R n  RS EQV
L
-  Cn
------------- = --------------------------------R n + RS EQV
DCR
(EQ. 31)
Solving for Cn we now have Equation 32.
L
------------DCR
C n = ----------------------------------R n  RS EQV
---------------------------------R n + RS EQV
(EQ. 32)
Note, RO was neglected. As long as the inductor time
constant matches the Cn, Rn and Rs time constants as given
previously, the transient performance will be optimum. As in
the static droop case, this process may require a slight
adjustment to correct for layout inconsistencies. For the
example of L = 0.36µH with 0.8m DCR, Cn is calculated in
Equation 33.
0.36H
-------------------0.0008
C n = ----------------------------------------------------------------------  330nF
parallel  5.823K, 1.825K 
(EQ. 33)
The value of this capacitor is selected to be 330nF. As the
inductors tend to have 20% to 30% tolerances, this capacitor
generally will be tuned on the board by examining the
transient voltage. If the output voltage transient has an initial
dip lower than the voltage required by the load line and
slowly increases back to steady state, the capacitor is too
small and vice versa. It is better to have the capacitor value
a little bigger to cover the tolerance of the inductor to prevent
the output voltage from going lower than the spec. This
capacitor needs to be a high grade capacitor like X7R with
low tolerance. There is another consideration in order to
achieve better time constant match mentioned previously.
The NPO/COG (class-I) capacitors have only 5% tolerance
and very good thermal characteristics. However, these
capacitors are only available in small capacitance values. In
order to use such capacitors, the resistors and thermistors
surrounding the droop voltage sensing and droop amplifier
has to be resized up to 10x larger to reduce the capacitance
by 10x. Careful attention must be paid in balancing the
impedance of droop amplifier in this case.
Dynamic Mode of Operation - Compensation
Parameters
Considering the voltage regulator as a black box with a
voltage source controlled by VID and a series impedance, in
order to achieve the 2.1mV/A load line, the impedance
needs to be 2.1m. The compensation design has to target
the output impedance of the converter to be 2.1m. There is
FN6398.4
August 25, 2015
ISL6266, ISL6266A
a mathematical calculation file available to the user. The
power stage parameters such as L and Cs are needed as
the input to calculate the compensation component values.
Attention must be paid to the input resistor to the FB pin. Too
high of a resistor will cause an error to the output voltage
regulation because of bias current flowing in the FB pin. It is
better to keep this resistor below 3k when using this file.
Static Mode of Operation - Current Balance Using
DCR or Discrete Resistor Current Sensing
Current Balance is achieved in the ISL6266A by measuring
the voltages present on the ISEN pins and adjusting the duty
cycle of each phase until they match. RL and CL around
each inductor, or around each discrete current resistor, are
used to create a rather large time constant such that the
ISEN voltages have minimal ripple voltage and represent the
DC current flowing through each channel's inductor. For
optimum performance, RL is chosen to be 10k and CL is
selected to be 0.22µF. When discrete resistor sensing is
used, a capacitor most likely needs to be placed in parallel
with RL to properly compensate the current balance circuit.
ISL6266A uses an RC filter to sense the average voltage on
phase node and forces the average voltage on the phase
node to be equal for current balance. Even though the
ISL6266A forces the ISEN voltages to be almost equal, the
inductor currents will not be exactly equal. Using DCR
current sensing as an example, two errors have to be added
to find the total current imbalance.
1. Mismatch of DCR: If the DCR has a 5% tolerance, the
resistors could mismatch by 10% worst case. If each
phase is carrying 20A, the phase currents mismatch by
20A*10% = 2A.
2. Mismatch of phase voltages/offset voltage of ISEN pins:
The phase voltages are within 2mV of each other by the
current balance circuit. The error current that results is
given by 2mV/DCR. If DCR = 1m then the error is 2A.
In the previous example, the two errors add to 4A. For the
two phase DC/DC, the currents would be 22A in one phase
and 18A in the other phase. In the previous analysis, the
current balance can be calculated with 2A/20A = 10%. This
is the worst case calculation. For example, the actual
tolerance of two 10% DCRs is 10%*(2) = 7%.
There are provisions to correct the current imbalance due to
layout or to purposely divert current to certain phase for
better thermal management. The Customer can put a
resistor in parallel with the current sensing capacitor on the
phase of interest in order to purposely increase the current in
that phase.
If the PC board trace resistance from the inductor to the
microprocessor are significantly different between two
phases, the current will not be balanced perfectly. Intersil
has a proprietary method to achieve the perfect current
sharing in cases of severely imbalanced layouts.
28
When choosing the current sense resistor, both the
tolerance of the resistance and the TCR are important. Also,
the current sense resistor’s combined tolerance at a wide
temperature range should be calculated.
Droop Using Discrete Resistor Sensing Static/Dynamic Mode of Operation
Figure 42 shows the equivalent circuit of a discrete current
sense approach. Figure 33 shows a more detailed
schematic of this approach. Droop is solved the same way
as the DCR sensing approach with a few slight
modifications.
First, because there is no NTC required for thermal
compensation, the Rn resistor network in the previous
section is not required. Second, because there is no time
constant matching required, the Cn component is not
matched to the L/DCR time constant. This component does
indeed provide noise immunity and therefore is populated
with a 39pF capacitor.
The RS values in the previous section, RS = 1.5k_1%, are
sufficient for this approach.
Now the input to the droop amplifier is essentially the
Vrsense voltage. This voltage is given by Equation 34.
R sense
Vrsense EQV = --------------------  I OUT
2
(EQ. 34)
The gain of the droop amplifier, Kdroopamp, must be adjusted
for the ratio of the Rsense to droop impedance, Rdroop by
using Equation 35.
R droop
K droopamp = -------------------------------- R sense  2 
(EQ. 35)
Solving for the Rdrp2 value, Rdroop = 0.0021(V/A) as per the
Intel IMVP-6+ specification, Rsense = 0.001 and Rdrp1 = 1k,
Equation 36 is obtained:
R drp2 =  K droopamp – 1   R drp1 = 3.2k
(EQ. 36)
Because these values are extremely sensitive to layout,
some tweaking may be required to adjust the full load droop.
This is fairly easy and can be accomplished by allowing the
system to achieve thermal equilibrium at full load, and then
adjusting Rdrp2 to obtain the desired droop value.
Fault Protection - Overcurrent Fault Setting
As previously described, the overcurrent protection of the
ISL6266A is related to the droop voltage. Previously the
droop voltage was calculated as ILoad*Rdroop, where Rdroop
is the load line slope specified as 0.0021 (V/A) in the Intel
IMVP-6+ specification. Knowing this relationship, the
overcurrent protection threshold can be programmed as an
equivalent droop voltage droop. Knowing the voltage droop
level allows the user to program the appropriate drop across
the ROC resistor. This voltage drop will be referred to as
FN6398.4
August 25, 2015
ISL6266, ISL6266A
10µA
OCSET
+Voc -Roc
+
OC
RS
VSUM
+
DROOP
-
INTERNAL TO
ISL6266A
VDIFF
+
DROOP
+
1 -
1
Rsense
Vrsense EQV = I OUT  ----------------------2
+
+
-
RTN VSEN
VO'
-
VN
Cn
-
Rdrp1
+
RS
= -------2
DFB
Rdrp2
+
VSUM
EQV
VO'
RO
EQV
RO
= --------2
FIGURE 42. EQUIVALENT MODEL FOR DROOP AND DIE SENSING USING DISCRETE RESISTOR SENSING
VOC. Once the droop voltage is greater than VOC, the PWM
drives will turn off and PGOOD will go low.
The selection of ROC is given in Equation 37. Assuming an
overcurrent trip level, IOC, of 55A, and knowing from the Intel
specification of the load line slope, Rdroop = 0.0021 (V/A),
ROC is calculated by Equation 37.
I OC  R droop
55  0.0021
R OC = ----------------------------------- = ------------------------------ = 11.5k
–6
10A
10  10
(EQ. 37)
Note, if the droop load line slope is not -0.0021 (V/A) in the
application, the overcurrent setpoint will differ from
predicted. In addition, due to the saturation limitations of the
DROOP amplifier, there is a maximum way-overcurrent
(WOC) set point for each VID code. The maximum OC set
point that will ensure WOC can be reached is expressed in
Equation 38:
1.75 – VID
I OC = -------------------------------------2.5  R DROOP
(EQ. 38)
The WOC limitation is only problematic at very high VID
settings (~1.350V and above).
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9001 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
29
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted. Please go to the web to make sure that
you have the latest revision.
DATE
REVISION
August 25, 2015
FN6398.4
CHANGE
Updated Ordering Information table on page 1.
Added Revision History and About Intersil sections.
Updated Package Outline Drawing L48.7X7 to the latest revision.
-Revision 4 to Revision 5 changes - Corrected Note 4 from: "Dimension b applies to.." to: "Dimension
applies to.." and enclosed Notes #'s 4, 5 and 6 in a triangle.
About Intersil
Intersil Corporation is a leading provider of innovative power management and precision analog solutions. The company's products
address some of the largest markets within the industrial and infrastructure, mobile computing and high-end consumer markets.
For the most updated datasheet, application notes, related documentation and related parts, please see the respective product
information page found at www.intersil.com.
You may report errors or suggestions for improving this datasheet by visiting www.intersil.com/ask.
Reliability reports are also available from our website at www.intersil.com/support.
30
FN6398.4
August 25, 2015
ISL6266, ISL6266A
Package Outline Drawing
L48.7x7
48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 5, 4/10
4X 5.5
7.00
A
44X 0.50
B
37
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
48
1
7.00
36
4. 30 ± 0 . 15
12
25
(4X)
0.15
13
24
0.10 M C A B
48X 0 . 40± 0 . 1
TOP VIEW
4 0.23 +0.07 / -0.05
BOTTOM VIEW
SEE DETAIL "X"
( 6 . 80 TYP )
(
0.10 C
BASE PLANE
0 . 90 ± 0 . 1
4 . 30 )
C
SEATING PLANE
0.08 C
SIDE VIEW
( 44X 0 . 5 )
C
0 . 2 REF
5
( 48X 0 . 23 )
( 48X 0 . 60 )
0 . 00 MIN.
0 . 05 MAX.
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
31
FN6398.4
August 25, 2015
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