AN1137

AN1137
Using the MCP1631 Family to Develop Low-Cost
Battery Chargers
Author:
Terry Cleveland
Microchip Technology Inc.
INTRODUCTION
As portable rechargable applications continue to grow,
there is an increase in demand for unique or custom
battery charger designs. In addition to the increase in
portable rechargable applications, battery chemistry
continues to improve and with that new charge
methods and profiles are emerging. This all leads to the
increase in demand for new or custom charge profile
designs. In this application note, a mixed signal multichemistry battery charger design technique will be
discussed that can accommodate the changing
portable power management world.
The reliability and safety concerns with charging batteries can also benefit from programmable mixed signal
designs. Charge rates and constant voltage levels can
be updated in the field with a change in firmware. This
allows the user to adapt to new smart battery packs and
select desired runtime versus cycle life. By charging
the battery to a lower constant voltage, the run time is
shortened but the number of charge cycles will
increase.
Another programmable battery charger feature is its
ability to charge multi-chemistry battery packs. By
detecting the number of cells and cell chemistry, a programmable charger can adapt to a new battery pack.
This enables customers to choose between portability,
runtime and cost when purchasing a portable system.
© 2007 Microchip Technology Inc.
COMMON CHARGE PROFILES
NiMH Charge Profile
Figure 1 shows a typical charge profile for NiMH
batteries. The charge cycle begins once a battery is
detected by regulating a small current or conditioning
current into the battery pack. If the cell voltage is above
0.9V per cell, it is safe to charge the pack with a fast
charge or high current (for NiMH or NiCd, this current
can range from 50% to over 100% of the batteries
capacity). When the battery reaches capacity, cell
manufactures recommend a top-off charge to complete
the charge cycle. It is typically not recommended to
trickle charge NiMH batteries, this can lead to
overheating and reduced battery life. Fast charge
termination for NiMH batteries can be tricky. As the
battery reaches capacity, it no longer can accept a
charge. The energy from the charger that was stored in
the battery, now turns into heat causing the battery
temperature to rise. There are two primary methods to
determine when the battery has reached full charge,
one is a sudden increase in temperature, the other
being a subtle drop in battery voltage or -dV/dt. With
NiMH batteries, the -dV/dt can be difficult to detect,
since the change can be very small, especially with
lower charge rate designs. The +dT/dt or temperature
rise is typically easier to detect. For a robust design,
both methods should be used so either can terminate
the fast charge portion of the charge cycle. Once the
fast charge is terminated, a timed top off charge is
recommended, a continuous constant charge is not
recommended for NiMH batteries.
DS01137A-page 1
AN1137
NiMH and NiCd Charge Profile
Stage 1
Pre-Charge
Stage 2
Fast Charge
Stage 3
End Fast Charge
Stage 4
Top Off Charge
-dV/dt
0.8V
VCELL
ICH = 1.0C
-dT/dt
VCELL (V)
ICH (A)
T(°C)
FIGURE 1:
ICH = 0.2C
Pack T (°C)
1 hour
ICH = 0.05C
NiMH / NiCd Charge Profile.
Li-Ion Charge Profile
The charge profile for Li-Ion batteries starts with cell
qualification. The cell voltage should be greater than
3.0V per cell before initiating a fast or high current
charge. If the cell voltage is less than 3.0V per cell, a
low value conditioning current is used to start the
charge cycle. Once the cell voltage is above the 3.0V
threshold, a fast charge or high current charge is
initiated (0.5C to 1.0C). As the battery cell voltage
rises, it reaches the maximum voltage value before it
reaches full capacity. As an example, most Li-Ion
batteries constant voltage level is 4.2V, where the
battery charger now transitions into a constant voltage
source (regulating voltage instead of current). The
charge cycle continues as the charge current
decreases while in the constant voltage mode. Once
the charge current decreases to about 7% of the fast
charge value, charge is terminated. Continuing the
charge cycle past this point can damage the battery so
the charge must be terminated. Once terminated a new
charge cycle can be initiated when the battery voltage
decreases to approximately 4.0V.
DS01137A-page 2
© 2007 Microchip Technology Inc.
AN1137
Li-Ion Charge Profile
Stage 1
Pre-Charge
Stage 2
Constant Current
VCELL
FIGURE 2:
ICH = 0.2C
Stage 4
Termination
4.2V
ICH = 1.0C
2.8V
VCELL (V)
ICH (A)
T(°C)
Stage 3
Constant Voltage
ICH = 0.07C
Pack T (°C)
ICH = 0C
Li-Ion Charge Profile.
Multi-Chemistry Charger
There are significant differences in the charge profile
between Ni batteries versus Li-Ion batteries. A multichemistry charger must be able to implement the
proper profile and proper termination methods. This
application note will demonstrate a charger that has the
capability to charge single or multiple cells in series.
THE POWER BEHIND CHARGING
BATTERIES
·
1 – Eff
P DISS = P OUT × ⎛ ----------------⎞
⎝ Eff ⎠
A switching charger solution operating at similar
conditions at 85% efficiency would dissipate
approximately 1.05 Watts, making it much easier to
cool. For high input voltage applications, switching
battery chargers are smaller and more cost effective.
A battery charger and power supply have a lot in
common, delivering a regulated output from a varying
input. Two solutions are prevalent, linear and switch
mode solutions. The linear solution is commonly used
for low input voltage or low power applications. Its main
drawback is internal power dissipation, calculated by
the following formula:
P DISS = ( V IN – V BATT ) × I CHARGE
For example, a +12V input linear charger would
dissipate 18 watts when charging a +3.0V Li-Ion battery
at 2A. Any power dissipation over a few watts is a
challenge to cool.
Cooling 18 watts of power dissipation is no easy task,
airflow and large heatsinks are required making a linear
solution impractical.
© 2007 Microchip Technology Inc.
DS01137A-page 3
AN1137
CHARGER POWER TOPOLOGY
• Primary Inductive Converter:
- The SEPIC converter topology has an inductor at the input, smoothing input current
reducing necessary filtering and generated
source noise.
• Single Low Side Switch:
- A single low side switch reduces MOSFET
drive and current limit protection complexity.
• Buck-Boost Capability:
- For applications where the input voltage can
be above or below the battery voltage a
SEPIC can buck or boost the input voltage.
Many switching regulator power topologies exist, buck,
boost, SEPIC and flyback are all used to develop
switching battery chargers (including others for very
high power applications). A SEPIC converter is
commonly used, it has advantages over buck and
boost converters when used in battery charger
applications.
• Capacitive Isolation:
- There is no direct dc path from input to output
providing isolation, this results in less power
components and a safer battery charger.
SEPIC Converter
Capacitive Isolation
Coupled Inductor
Blocking Diode
CC
+12V Input
+Vbatt
CIN
1
COUT
1
ISENSE
Switch
Batteries
VEXT
CS
Input Current
FIGURE 3:
SEPIC Topology.
MULTI-CHEMISTRY BATTERY
CHARGER DESIGN
The development of an intelligent multi-chemistry
battery charger starts with the microcontroller. By
implementing the charge algorithm in code, the charger
can be adapted for multi-chemistry, custom charge
profile and unique applications. For dc-dc converters,
switching at high frequency with high performance gate
drive capability, PWM control and high-speed protection, specialized analog circuitry is required. A new
high-speed analog PWM, the MCP1631HV was
developed for constant current SEPIC applications
(battery chargers and LED drivers). By implementing
the pulse width modulation, PWM, control using the
DS01137A-page 4
RLIMIT
MCP1631, the battery charger has the benefits of
analog speed and resolution. By controlling the charge
algorithm using the microcontroller, the battery charger
has the intelligence and flexibility to generate a profile
for all battery types using digital timers and
programmed algorithms.
As complex as this project sounds, it is really quite
simple if the SEPIC converter is thought of as a microcontroller controlled current source. To increase current, the microcontroller simply increases the VREF
input to the MCP1631HV and to decrease current, the
microcontroller decreases the VREF input to the
MCP1631HV. To generate a charge algorithm, the
microcontroller measures the battery voltage using an
© 2007 Microchip Technology Inc.
AN1137
analog to digital converter(A/D), computes the desired
charge current and adjusts the SEPIC controlled
current source up or down.
To develop the charge algorithm for the NiMH battery,
the microcontroller A/D converter is used to measure
the battery pack voltage, when the pack voltage is
within the desired range, the microcontroller sets the
proper current level. To terminate the charge, two A/D
inputs are used, one to sense the decreasing battery
voltage and one to sense the increasing battery pack
temperature. Charge termination will occur, if either
one or both are detected.
To develop the algorithm for charging Li-Ion batteries,
the A/D converter is used to measure pack voltage.
Depending on pack voltage, the microcontroller will set
the appropriate charge current. Once the pack voltage
reaches the constant voltage phase, the A/D converter
senses and regulates the pack voltage by adjusting the
amount of current into the battery. The current continues to decrease until is reaches about 7% of the fast
charge value. At this point, the microcontroller
terminates the charge.
VIN = +5.3V to +16.0V
AVDD_OUT = +5.0V
C
C
250 mA Available
MCP1631HV
µController
VDD Input
VIN = +3.8V to +16.0V
AVDD_OUT = +3.3V
C
C
250 mA Available
MCP1631HV
µController
VDD Input
The MCP1631HV Implementation
The MCP1631HV integrates the necessary blocks to
develop an intelligent, programmable battery charger
or constant current source used for driving high power
LED’s.
INPUT VOLTAGE AND BIAS GENERATION
The MCP1631HV provides a regulated bias voltage for
internal circuitry that is available for biasing the microcontroller and other components. It is available in two
regulated voltage options, +5.0V and +3.3V and can
handle a maximum output current of 250 mA. The
maximum input voltage range for the regulator is
+16.0V and can withstand transients to +18.0V. For
regulated input voltages or higher input voltage
applications, the MCP1631 device option without
internal regulator can be used. By using a high voltage
regulator to bias the MCP1631 and microcontroller, the
range of input voltage for the design is only limited by
the regulator maximum input and power train design.
AVDD_IN = USB +5.0V
C
MCP1631
µController
VDD Input
High Voltage
Input
High
Voltage
Linear
C
Regulator
Regulated
+3.3V or +5.0V
MCP1631
C
µController
FIGURE 4:
MCP1631HV and MCP1631
Bias Voltage Options.
© 2007 Microchip Technology Inc.
DS01137A-page 5
AN1137
HIGH SPEED ANALOG PWM OPERATION
ramping current is used for peak current mode control
CS signal. A filter is used on the CS input to remove the
leading edge turn on spike associated with the turn on
of the external power MOSFET. The driver P-Channel
MOSFET is powered using a separate PVDD pin
helping to keep switching noise off of the AVDD pin and
sensitive CS circuitry.
The high-speed analog PWM is used to control the
power train switch ON and OFF times to regulate the
output of the converter. Voltage or current can be
regulated depending on what is being sensed. For the
SEPIC Battery Charger application, the MCP1631HV
is always regulating current, the microcontroller is
programming this current.
The error amplifier is configured as an integrator, so
any difference between its inputs, VREF and VFB are
quickly removed. If the VFB input is high, the inverting
error amplifiers output, (COMP), will be pulled down,
lowering the peak current into the switch and lowering
duty cycle bringing the output back into regulation. The
external R and C used for compensation is used to control the speed of the error amplifiers output response. If
not compensated properly, the error amplifier output
will move to fast (unstable system with under damped
oscillations) or slow (over damped system with no
performance or response to changes). The VREF input
is set by the microcontroller to program the proper
charge current.
The analog PWM starts with the oscillator input,
typically a microcontroller PWM output or simple clock
output (50% duty cycle). When the oscillator input is
high, the VEXT output is pulled low, (N-Channel MOSFET Driver is ON). A new cycle is started when the
OSC_IN input transitions from a high to a low, the internal N-channel MOSFET driver turns off and the PChannel MOSFET turns on driving the VEXT pin high
turning on the external N-Channel MOSFET. Current
begins to ramp up in the external CS sense resistor
until it reaches 1/3 of the level of the error amplifier
output voltage (limited to 0.9V by error amplifier clamp).
The 0.9V limit is used as an overcurrent limit, the
OSC_IN
Low = Active Duty Cycle
High = VEXT OFF
V = COMP/3
PVDD
Latch
S
MOSFET P
CS INPUT
VEXT
Q
VREF
VFB
+
A1
+
C1
-
COMP 2R
-
R
PGND
R
Error Amplifier
and Compensation
MOSFET N
High Speed Comparator
Note 1: A1 output or COMP is clamped to 2.7V maximum to set current limit.
FIGURE 5:
DS01137A-page 6
Analog PWM Operation.
© 2007 Microchip Technology Inc.
AN1137
CURRENT REGULATION
MCP1631HV integrates an inverting 10V/V gain
amplifier to increase the battery current sense signal.
The microcontroller sets the VREF input to the desired
current level, the MCP1631HV uses the VREF input as
a reference for regulation.
To sense battery current for regulation in a SEPIC converter, the secondary winding of the coupled inductor
can be used. The average current flowing through the
secondary winding is equal to the current flowing into
the battery. As shown, this topology does not require
the sense resistor in series with the battery, removing
any power lost in series with the battery while running
the system. When sensing battery current, a low value
sense resistor is desired to minimize power loss, the
The resistor in series with the external SEPIC switch
provides a high speed current limit protecting the
switch and other power train components from a short
circuit or over current condition.
VIN
CIN
CS INPUT
VREF
VFB
+
COUT
+
C1
-
2R
A1
-
IBATT
R
BATT
IINPUT
10R
10X IBATT
FIGURE 6:
R
A2
+
R
Current Regulation Diagram.
SENSING BATTERY VOLTAGE
Using the internal microcontroller A/D converter to
sense battery voltage is a popular approach. An issue
with this technique is the A/D converter requires a low
source impedance to perform accurate readings. Low
source impedance requires low resistance values that
draw excessive quiescent current from the battery. The
MCP1631HV integrates a low current amplifier (A3),
configured as a unity gain buffer. The buffer output
impedance is low, driving the SAR A/D converter, while
consuming very little quiescent current. A high value
resistor divider is used to drop the battery voltage to an
acceptable range. R1, R2 and R3 values are selected
to minimize the drain on the batteries, typically drawing
on the order of 1 µA. The microcontroller reads the A/D
converter, calculates the current setting and adjusts the
VREF input to regulate current.
removed or opens. OV protection is typically required
for any current source application (battery chargers,
LED drivers).
The MCP1631HV integrates an internal high speed OV
comparator that has a 1.2V reference connected to its
inverting input. If the voltage on the OV_IN pin exceeds
the 1.2V threshold, the VEXT output is asychronously
terminated. Switching will resume after the voltage has
dropped more than the built in 50 mV of hysteresis. If a
battery is removed during the charge cycle, the charger
output voltage will be limited to a safe value.
Overvoltage (OV) protection is a common battery
charger protection feature. The OV protection is not
there to protect the battery, it is used to protect the
power train from excessive voltage if the battery is
© 2007 Microchip Technology Inc.
DS01137A-page 7
AN1137
CS INPUT
COUT
VREF
VFB
+
+
C1
-
2R
A1
-
To PWM
Latch H = PWM
OFF
BATT
R
R1
VS_OUT
to microcontroller
A/D Converter
+
A3
-
R2
OV COMP
+
C2
-
R3
+1.2V
Comp
FIGURE 7:
DS01137A-page 8
MCP1631HV Voltage Buffer and Overvoltage Comparator Setup.
© 2007 Microchip Technology Inc.
AN1137
System Level Block Diagram
The system level block diagram shown in Figure 7
represents all of the MCP1631HV internal blocks. The
SHDN input is used to turn off the charger and lower
the quiescent current draw to a 4.4 µA typical, the +5V
generated bias is available and A3 remain powered for
battery monitoring and microcontroller power.
MCP1631HV/VHV High-Speed Analog PWM
+3.3V or +5.0V
LDO
250 mA
VIN
VDD
Internal
1.2V VREF
VDD
AVDD_OUT / AVDD_IN
Shutdown Control
A3 Remains On
SHDN
Overvoltage Comp
w/ Hysteresis
C2
+
OVIN
PVDD
OSCDIS
VDD
100 kΩ
0.1 µA
OT
OSCIN
VEXT
UVLO
S
VDD
PGND
Q
VDD
+
C1
-
COMP
R
A2
+
A1
+
2R
ISIN
R
VDD
2.7V Clamp
Note 1: For Shutdown control, amplifier A3 remains functional so
battery voltage can be sensed during discharge phase.
A3
VSIN
-
R
AGND
R
ISOUT
+
VREF
100 kΩ
10R
VDD
VDD
FB
Q
-
CS/VRAMP
VSOUT
FIGURE 8:
MCP1631HV Block Diagram.
© 2007 Microchip Technology Inc.
DS01137A-page 9
AN1137
Charger Reference Board Design
charger application.
A battery charger reference design was developed for
the MCP1631HV to evaluate the device in a battery
Multi-cell, Multi-Chemistry Charger
VIN Range +5.5V to +16V
L1A
CC
SCHOTTKY
DIODE
COUT
CIN
L1B
MCP1631HV
+VDD_OUT
VEXT
VIN
ILIMIT
BATTERY
ISENSE
CS
AVDD_OUT
FSW SET
RTHERM
PGND
PVDD
OSCIN
ISIN
ISOUT
OVIN
NC
VSIN
FB
VREF
NC
SHDN
COMP
OSCDIS
AGND
VSOUT
0V PROTECTION
LOW IQ SHUTDOWN
PROGRAMMAGLE
CURRENT SOURCE
REFERENCE
R
C
PIC® Microcontroller
VDD
GP1/C
CCP1
GP3
GP4
GP5
GND
GP0/C
AVDD_OUT
LED
STATUS INDICATOR
FIGURE 9:
DS01137A-page 10
Charger Diagram.
© 2007 Microchip Technology Inc.
AN1137
A
K
A
K
A
K
A
K
A
K
A
K
1
2
3
4
5
6
7
8
K
A
28
27
26
25
24
23
22
21
20
19
18
17
16
15
1
2
3
4
5
6
7
8
9
10
11
12
13
14
K
A
FIGURE 10:
Detailed Schematic.
© 2007 Microchip Technology Inc.
DS01137A-page 11
AN1137
FIGURE 11:
DS01137A-page 12
Board Layout.
© 2007 Microchip Technology Inc.
AN1137
THE DESIGN DETAILS OF CHARGING
BATTERIES USING THE PIC®
MICROCONTROLLER AND
MCP1631HV WITH A SEPIC
TOPOLOGY
Design Example:
•
•
•
•
•
•
VIN = 12V
VBATT = 0V to 4.2V
IBATT = 200 mA Pre-Charge Current
IBATT = 2A Fast Charge Current
IBATT = 140 mA termination or “tail” current
Overvoltage Protection
SEPIC Power Train Design
• Calculate Maximum Output Power
P OUT = V BATT × I BATT
POUT = 4.2V X 2.0A or 8.4 Watts
P OUT
P IN = -------------------------Efficiency
• By making an efficiency estimate, the converter
input power can be estimated. The typical
efficiency of a SEPIC converter in this power
range using a schottky diode for the output
rectifier is around 85%.
• PIN = 8.4 Watts / 0.85 or 9.88 Watts
• IIN = PIN / IIN
- IIN = +12V / 0.88 Watts
- IIN = 1.21 A
With IIN and IBATT known, the average inductor current
for each winding is known.
Inductor Ripple Current
For the coupled inductor, the effective inductance is
twice the value of the inductor, this is a result of 2x the
voltage across 2x the number of turns. Since the value
of L is proportional to n2, the effective inductance is
twice the actual value of the inductor.
2×V
-----------2
n
© 2007 Microchip Technology Inc.
A 10 µH inductor looks like a 20 µH inductor (for
coupled inductors only). Larger inductance reduces
ripple current and operates in the continuous mode at
lighter loads, an advantage over non-coupled inductor
solutions.
The input and output inductor ripple current is equal to:
VL
ΔI L = ------ × t ON
L
Where TON is the amount of time the SEPIC switch is
turned on:
1
t ON = DutyCycle × ---------F SW
Where Duty Cycle for a SEPIC converter operating in
continuous conduction mode is equal to:
V OUT
DutyCycle = --------------------------V OUT + V IN
To derive the transfer function of the SEPIC converter,
start by balancing the inductor volt-time product in the
boost stage (W1).
Q1 Turned on (+ Slope):
ΔI W1 ⁄ t ON = V IN ⁄ L W1
Q1 Turned off (- Slope):
V C1 + V OUT – V IN
ΔI W1 ⁄ t OFF = -------------------------------------------L W1
Inductor slope’s must be equal for volt-time balance:
V IN
V C1 + V OUT – V IN
t ON × ---------- = t OFF × -------------------------------------------L W1
L W1
Multiply both sides by 1/(tON + tOFF):
V IN × D = ( V C1 + V OUT – V IN ) × ( 1 – D )
Solve for VC1: .
1
V C1 = V IN × ⎛ -------------⎞ – V OUT
⎝ 1 – D⎠
For the second stage, the inductor slopes must also be
equal.
Q1 Turned on (+ Slope):
ΔI W2
V C1
------------ = --------t ON
L W2
DS01137A-page 13
AN1137
Q1 Turned off (- Slope):
V OUT
ΔI W2
------------ = ------------t OFF
L W2
Inductor slope’s must me equal for volt-time balance:
V C1
V OUT
t ON × ---------- = t OFF × ------------L W2
L W2
Multiply both sides by 1/(tON + tOFF):
V C1 × D = V OUT × ( 1 – D )
Solving for VC1.
1–D
V C1 = V OUT × ⎛⎝ -------------⎞⎠
D
Set VC1 = VC1 for both the Boost stage and the BuckBoost stage:
1
1–D
V C1 = V IN × ⎛⎝ -------------⎞⎠ – V OUT = V OUT × ⎛⎝ -------------⎞⎠
1–D
D
Solving for VOUT/VIN:
V OUT
D
------------- = ⎛ -------------⎞
⎝ 1 – D⎠
V IN
Looking back, if D/(1-D) x VIN is substituted for VOUT, it
is shown that VC1 = VIN. This is true, if C1 is large
enough that the ripple voltage on C1 is low.
Now that the duty cycle is known as a VOUT/VIN
relationship, the duty cycle can be calculated for any
input output condition. Remember, this transfer
function is dependent upon the fact that inductor
current is continuous or never reached zero. If it does
reach zero, this transfer function is no longer true and
there is another state added to the operation.
DS01137A-page 14
© 2007 Microchip Technology Inc.
AN1137
Power Train Design
W1
IW1
+
VIN
C1
+
D
Q1
IW2
W2
COUT
+
ICHARGE
A
W1
+
IW1
+
VIN
C1
IW2
W2
COUT
+
ICHARGE
B
W1
IW1
+
VIN
C
+ 1
IW2
W2
COUT
+
ICHARGE
C
Sum of Winding Currents
Diode Current
IW2
Switch Current
IW1 + IW2
tON
tOFF
IOUT (Average)
IOUT x (VOUT/VIN)
IIN (AVERAGE)
IW1
tON
FIGURE 12:
tOFF
tON
tOFF
SEPIC Converter Inductor, Switch and Diode Currents.
© 2007 Microchip Technology Inc.
DS01137A-page 15
AN1137
Inductor Winding Current Calculation
The first step to calculating the inductor winding current
is to know the maximum output power. For this
constant current battery charger application, the output
power is simply the maximum output voltage times the
charge current.
P OUT = V OUT × I CHARGE
Maximum output voltage is equal to 4.2V (1 battery @
4.2V).
POUT =4.2V x 2 A or 8.4 Watts.
Since energy is conserved, the input power is equal to
the output power (assuming 100% efficiency). An
efficiency estimate can be used to closer approximate
the input current.
P IN = P OUT ⁄ ( Efficiency )
Where:
PIN
PIN
=
8.4 Watts / 85%; 85% used as a
typical efficiency estimate
=
9.88 Watts
The average input current is equal to the input power
divided by the input voltage:
I IN ( AVG ) ) = P IN ⁄ V IN
Where:
IINAVG
=
9.88 Watts/12V (Nominal)
IINAVG
=
824 mA. (Typical average input
current
The peak-to-peak W1 inductor current ripple
calculation was shown earlier. Given the derived
transfer function and the maximum voltage on the
output of the converter to be 4.2V, the switch on time is
estimated.
Switch On Time:
t ON
V OUT ⁄ ( V OUT + V IN )
= --------------------------------------------------F SW
For the 12V input and 4.2V output case, the switch ON
time is estimated to be approximately 519 ns. (500 kHz
switching frequency).
The input peak-to-peak ripple
calculated:.
GIVEN:
current
can
be
LW1 = LW2 = 20 µH (10 µH Coupled)
Input Peak-to-Peak Ripple Current (W1)
ΔIL(W1)
=
(12V / 20 µH) x tON = 311 mA
IL(W1)PK
=
IINAVG +1/2 x ΔIL(W1)
IL(W1)PK
=
980 mA for winding 1 (W1)
IL(W1)MIN
=
IINAVG -1/2 x ΔIL(W1)
IL(W1)MIN
=
669 mA for winding 1 (W1)
The ripple current in winding (W2) is calculated in a
similar fashion. The main difference is that the average
current in W2 is equal to IOUT or 2A in this application.
W2 Peak-to-Peak Ripple Current
ΔIL(W2) = (12V / 20 µH) x tON = 311 mA
IL(W2)PK = IOUTAVG +1/2 x ΔIL(W2)
IL(W2)PK = 2.16 A for winding 2 (W2)
IL(W2)MIN = IOUTAVG -1/2 x ΔIL(W2)
IL(W2)MIN = 1.85 A for winding 1 (W2)
Note:
In the case of VIN = VOUT, the current in
W1 = W2 (ripple and average).
The coupled inductor winding currents calculated
above are used to determine the size of the inductor
necessary. High switching frequency has several
advantages, smaller ripple current, lower peak and
RMS current and lower volt-time product on the
inductor core. This leads to a small, low-cost solution.
SEPIC Switch Current and Voltage Calculations
The switch current (IQ1) is equal to the combination of
the winding currents during the switch on time. When
the switch is turned on, it conducts the current in W1
and W2.
ISW = IW1 + IW2 = 2.82A (Average)
ISWPK = 2.82A + 311 mA = 3.14A
The minimum switch current is equal to:
ISWMIN = 2.82A - 311 mA = 2.51A
RMS of a Trapezoidal waveform
2
I SWRMS =
2
⎛ I A + I A × I B + I B⎞
D × ⎜ ---------------------------------------⎟
3
⎝
⎠
Where:
IA
=
2.51A = Minimum,
IB
=
3.14A = Maximum
The RMS value of the switch current is approximately
1.44 mA.
DS01137A-page 16
© 2007 Microchip Technology Inc.
AN1137
The peak switch voltage is equal to VIN + VOUT for the
SEPIC converter. Any leakage inductance voltage
spike is clamped through the output diode by the output
capacitor. A switch voltage rating for this application
should be a minimum of VIN(MAX) + VOUT(MAX).
VSW = 12V +4.2V
VSW = 16.2V
A 30V, 30 milli-ohm, logic-level switch is selected.
MOSFET switching losses should also be considered
when selecting the MOSFET switch. Low on resistance
switches tend to have high capacitance and will switch
slower, increasing switching losses. The lowest RDSON
MOSFET is not necessarily the best choice. When
using the SOIC-8 package for a 30V MOSFET, there
are many choices available.
SEPIC Diode Voltage and Current Calculations
A schottky diode is recommended for low-voltage
applications. For battery charger applications, the
SEPIC diode will block current flow from the battery
back to the input. The reverse leakage current of the
selected schottky diode can be a critical parameter, if
low battery drain is desired. Low schottky diode forward
drop is also a key parameter; the low drop improves
converter efficiency.
The maximum reverse voltage across the SEPIC diode
occurs during the switch on time. The cathode of the
schottky diode is connected to VOUT, the anode of the
schottky diode is connected to the SEPIC coupling
capacitor. The voltage across the coupling capacitor
voltage is equal to VIN; the voltage across the diode is
equal to VOUT - (-VIN) or VOUT + VIN.
The peak SEPIC diode current occurs when the switch
is turned off. The peak diode current is equal to the
peak current in W2, plus the peak current in W1 or
3.14A. The average diode current is equal to the output
current (IOUT), typical of all topologies with a series
diode in the path of the output.
SEPIC Coupling Capacitor (C1) RMS Current
Calculations and Voltage Rating
The RMS current in the SEPIC coupling capacitor is
mainly dependant upon output power with some
influence by inductor ripple current. As output power
increases, the capacitor ripple current will increase as
well. As shown in Figure 12 (during the switch on time),
the current in winding 2 (output current) is flowing
through the coupling capacitor C1. During the switch off
time, the C1 current is equal to the current in winding
number 1 (W1). As previously discussed, the W1
current is equal to the average input current. Therefore,
the worst case or maximum RMS current in the
coupling capacitor will occur at maximum output power
© 2007 Microchip Technology Inc.
and minimum input voltage. To estimate size for the
coupling capacitor, the capacitor derivative equation
can be used.
dV
I C = C × ------dt
The rate of change of voltage across the capacitor is
related to the amount of current through the capacitor
and the size or energy storage capability of the
capacitor.
For the SEPIC converter coupling capacitor, the
voltage is approximated to be a DC value when
deriving the duty cycle. The ripple voltage should be no
more than 5% of the voltage across the capacitor or the
input voltage. In this example, the input voltage and C1
DC voltage is 12V, so there should be no more than 5%
or 600 mV of ripple on the coupling capacitor.
In this example there is an average of 2A flowing
through the coupling capacitor during the switch on
time. The on time is approximately 26% or 520 ns. To
keep the capacitor voltage ripple less than 5% of VIN,
or 600 mV, the amount of capacitance is equal to (2A)
/ (600mV/520 ns) or 1.73 µF. For this application a
standard value 2.2 µF X7R 25V rated ceramic
capacitor should be used.
+IW2
IW2
IW1
0
IC1
- IW1
Note:
Area above 0 equals the area below
zero.
FIGURE 13:
C1 Ripple Current.
As shown in Figure 13, the coupling capacitor ripple
current is largely dependent upon output power and
input voltage. As the input voltage decreases, the
current in W1 increases. During the switch on time, the
current flowing in W2 is equal to the current flowing in
C1. When the switch turns off, the current quickly
changes magnitude and direction so that the current
flowing in C1 is equal to the current in W1, magnitude
and direction.
DS01137A-page 17
AN1137
As an approximation, the RMS current in C1:
I C1 ( RMS ) = I OUT × V OUT ⁄ V IN
For worst-case situations, the RMS current in the C1
coupling capacitor is equal to 2A x (4.2V / 12V)1/2 or
1.18A. The current rating for small multi-layer ceramic
capacitors is typically much higher than 1.18A. For
higher power applications, it may be necessary to use
multiple capacitors in parallel to keep the RMS current
within ratings.
DS01137A-page 18
CONCLUSION
For applications that require intelligent power management solutions like battery chargers, the combination of
a microcontroller and MCP1631 high-speed PWM is
very powerful. It brings the programmability benefits of
the microcontroller and adds the performance of a
high-speed analog PWM. The analog PWM will
respond to changes in input voltage and output current
very quickly. No code or execution time is necessary to
regulate or protect the circuit. The microcontroller is
used for programmability, establishing charge profile
conditions and monitoring the circuit for fault conditions
and taking the appropriate action, in the event of a
specific fault.
© 2007 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights.
Trademarks
The Microchip name and logo, the Microchip logo, Accuron,
dsPIC, KEELOQ, KEELOQ logo, microID, MPLAB, PIC,
PICmicro, PICSTART, PRO MATE, rfPIC and SmartShunt are
registered trademarks of Microchip Technology Incorporated
in the U.S.A. and other countries.
AmpLab, FilterLab, Linear Active Thermistor, Migratable
Memory, MXDEV, MXLAB, SEEVAL, SmartSensor and The
Embedded Control Solutions Company are registered
trademarks of Microchip Technology Incorporated in the
U.S.A.
Analog-for-the-Digital Age, Application Maestro, CodeGuard,
dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN,
ECONOMONITOR, FanSense, FlexROM, fuzzyLAB,
In-Circuit Serial Programming, ICSP, ICEPIC, Mindi, MiWi,
MPASM, MPLAB Certified logo, MPLIB, MPLINK, PICkit,
PICDEM, PICDEM.net, PICLAB, PICtail, PowerCal,
PowerInfo, PowerMate, PowerTool, REAL ICE, rfLAB, Select
Mode, Smart Serial, SmartTel, Total Endurance, UNI/O,
WiperLock and ZENA are trademarks of Microchip
Technology Incorporated in the U.S.A. and other countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
All other trademarks mentioned herein are property of their
respective companies.
© 2007, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
Printed on recycled paper.
Microchip received ISO/TS-16949:2002 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
© 2007 Microchip Technology Inc.
DS01137A-page 19
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10/05/07
DS01137A-page 20
© 2007 Microchip Technology Inc.
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