dm00189143

AN4710
Application note
HVLED001 – QR high power factor flyback controller for LED
drivers
Francesco Ferrazza
Introduction
The flexibility of the LED chips and their improving performances are more and more
exploited in many fields of applications. Besides the residential and bulb replacement
applications, whose input power spreads between 3 W and 15 W, a very wide range of
higher power applications are available for very different environment such office or store
lighting, professional or colored lighting and outdoor lighting. Those applications are
characterized by output powers up to 100 W.
Key factors of such applications are good efficiency, power controllability (e.g.: dimming)
and a small component count. In such condition, the single stage high power factor flyback
topology combines the advantages of the single stage (higher efficiency and low BOM
count) with the flexibility of the control algorithm (higher controllability).
The HVLED001 controller has been designed to optimize the control of a high power factor
flyback or buck-boost to be used in a LED driver. Nevertheless it is also able to efficiently
control the same topology to provide a constant output voltage exploiting a proprietary
primary side controlled algorithm. When driven by a PFC pre-regulator the HVLED001
device can be used as a DC/DC QR flyback converter.
This application note is intended to describe the HVLED001 features and to provide the
design guidelines to implement a single stage LED driver.
September 2015
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Contents
AN4710
Contents
1
HVLED001 features description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
1.1
Pin function summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
1.2
Operating modes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Start-up mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.2.2
Active mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.2.3
Stop mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.2.4
Low consumption mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.3
Device supply management . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
1.4
Peak current mode definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
1.5
1.6
1.4.1
Multiplier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
1.4.2
Current sense comparator system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
1.4.3
Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Smart zero current detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
1.5.1
Demagnetization detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
1.5.2
SMART ZCD detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
1.5.3
Frequency foldback . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Control loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
1.6.1
Primary side regulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
1.6.2
Secondary side regulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
1.6.3
Burst mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
1.7
Gate driving . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
1.8
Disabling features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
1.9
2/63
1.2.1
1.8.1
Instant disable by CTRL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
1.8.2
Instant disable by FB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
1.8.3
Timed disable by CTRL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
1.9.1
Input overvoltage and surge protection (IOVP) . . . . . . . . . . . . . . . . . . . 30
1.9.2
Overload and optocoupler failure management (OFP) . . . . . . . . . . . . . 31
1.9.3
Short-circuit detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
1.9.4
Brownout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
1.9.5
Magnetic saturation or rectifier short-circuit . . . . . . . . . . . . . . . . . . . . . . 33
1.9.6
VCC short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
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Contents
Designing a high power factor flyback LED driver . . . . . . . . . . . . . . . 34
2.1
Selecting the design input specifications . . . . . . . . . . . . . . . . . . . . . . . . . 34
2.2
Transformer design guide lines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
2.3
2.4
2.5
2.6
2.2.1
Primary inductance selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
2.2.2
Core size selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
2.2.3
Turn selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
2.2.4
Transformer design equation summary . . . . . . . . . . . . . . . . . . . . . . . . . 40
ZCD network definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
2.3.1
ZCD network to implement PSR (simple resistive network) . . . . . . . . . 41
2.3.2
ZCD network to implement PSR (network with derivative components) 42
2.3.3
ZCD network to operate without PSR . . . . . . . . . . . . . . . . . . . . . . . . . . 43
2.3.4
ZCD network driven by MOSFET's drain . . . . . . . . . . . . . . . . . . . . . . . . 43
Active components definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
2.4.1
Input rectifier bridge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
2.4.2
Secondary side rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
2.4.3
MOSFET selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
2.4.4
Clamping device selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Input and output filter definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
2.5.1
Output capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
2.5.2
Input capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
Control loops definition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
2.6.1
PSR control loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
2.6.2
Optocoupler based control loop (constant current) . . . . . . . . . . . . . . . . 51
2.7
Soft-starting the application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
2.8
Supplying the application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
2.8.1
Simple Zener regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
2.8.2
Linear voltage regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
Appendix A Theory of QR flyback topology fed by an AC line . . . . . . . . . . . . . 56
Appendix B List of abbreviations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
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List of figures
AN4710
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
Figure 34.
Figure 35.
Figure 36.
Figure 37.
Figure 38.
Figure 39.
Figure 40.
Figure 41.
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HVLED001 operation states diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
High voltage start-up block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Supply management unit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
First start-up time waveform . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Multiplier block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Current sense comparators system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
VCS,lim vs. Vctrl characteristic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
CTRL pin structures to be used for soft-start purpose . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
ZCD detection circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
ZCD related waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
TBLANK vs. TOFF voltage characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Example: TOFF voltage higher than 2 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Example: TOFF voltage lower than 2 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Example: TOFF voltage lower than 2 V and resulting in valley skipping . . . . . . . . . . . . . . 21
Example: TOFF voltage lower than 2 V with extinguished drain's resonance . . . . . . . . . . 22
PSR management logic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
OTA output equivalent model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
OTA output characteristic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
ZCD voltage divider settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Optocoupler operation typical arrangement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Disabling HVLED001 using CTRL pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Disabling HVLED001 using FB pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Timed disable of HVLED001 using CTRL pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Input overvoltage protection (IOVP) waveform . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
OFP protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Brownout protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Minimum primary inductance nomogram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Transformer's winding arrangement example. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
ZCD network (PSR - resistive) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
ZCD network (PSR - derivative) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
ZCD network to disable the PSR . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Capacitive coupled ZCD network without PSR. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
RCD clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Transil clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
PSR control loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
Secondary side error amplifier arrangement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
CTRL pin biasing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
Zener based VCC regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
Linear regulator for VCC supply. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
Estimated input current shape . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
THD vs. KV graph . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
DocID027926 Rev 1
AN4710
HVLED001 features description
1
HVLED001 features description
1.1
Pin function summary
Table 1. Pin function
Symbol Pin
HVSU
1
Function name
Section
High voltage start-up
Section 1.3: Device supply management on page 8.
Input voltage detection
Section 1.4: Peak current mode definition on page 10.
Input overvoltage detection
Section 1.9.1: Input overvoltage and surge protection (IOVP) on
page 30.
N.C.
2
Not connected pin
TOFF
3
ZCD blanking time setting
Section 1.5.3: Frequency foldback on page 17.
Input for optocoupler connection Section 1.6.2: Secondary side regulation on page 25.
FB
CTRL
ZCD
4
5
6
PSR E/A output connection
Section 1.6.1: Primary side regulation on page 23.
Opto failure protection
Section 1.9.2: Overload and optocoupler failure management
(OFP) on page 31.
Overload protection
Section 1.9.2.
Burst mode
Section 1.6.3: Burst mode on page 26.
Disable input (active low)
Section 1.8.2: Instant disable by FB on page 28.
Inrush limit / soft-start
Section 1.4.3 on page 14.
Disable input (active low)
Section 1.8.1: Instant disable by CTRL on page 26.
Timed disable (active high)
Section 1.8.3: Timed disable by CTRL on page 29.
ZCD detection
Section 1.5.2: SMART ZCD detection on page 17.
Vout sampling input for PSR
Section 1.6.1: Primary side regulation on page 23.
Brownout protection
Section 1.9.4: Brownout on page 32.
Current sense comparator input Section 1.4.2 on page 12.
CS
7
Overcurrent protection (OCP)
input
Section 1.9.5: Magnetic saturation or rectifier short-circuit on
page 33.
GND
8
Reference pin
-
GD
9
Gate driver output
Section 1.7: Gate driving on page 26.
VCC
10
Supply energy to the IC
Section 1.3: Device supply management on page 8.
Internal UVLO logic
Section 1.3.
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HVLED001 features description
1.2
AN4710
Operating modes
The HVLED001 device has four main operating modes: the start-up mode, active mode,
stop mode and low consumption mode.
1.2.1
Start-up mode
This state is entered to begin the switching activity after the application's turn-on or leaving
the low consumption state. The HVSU is involved into the mechanism of VCC charging; all
other peripherals, except from the UVLO and logic supply, are turned off to minimize the
start-up time.
During this state the CTLR pin is internally pulled to ground.
1.2.2
Active mode
It is the normal operational mode. During this state the external MOSFET is driven
accordingly to signals coming from the application in order to regulate the desired output
parameter in the closed loop (peak current control method).
The active mode is exit when abnormal conditions are present. The HVSU is inactive during
the active mode.
1.2.3
Stop mode
This state is intended to stop the switching activity without turning off the entire function set,
to quickly restart when abnormal or disabling conditions end. During this state the power
consumption is not minimized and the soft-start procedure is not enabled.
1.2.4
Low consumption mode
This state is intended to stop the switching activity reducing the power consumption to
a minimum level. During this state the VCC is kept between VCC,su and VCC,on by the
HVSU.
A simplified state diagram is reported in Figure 1.
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HVLED001 features description
Figure 1. HVLED001 operation states diagram
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HVLED001 features description
1.3
AN4710
Device supply management
[Involved pins 1: HVSU, 10: VCC]
The HVLED001 device embeds smart supply voltage management able to both prevent the
application from driving the MOSFET with insufficient energy and to maintain the precision
of the internal references.
A high voltage start-up unit, connected to the HVSU pin, provides the start-up current to
initiate the IC operations and maintain the IC on during low consumption modes (Figure 2).
Figure 2. High voltage start-up block diagram
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The HVSU starts its operation when the applied voltage is higher than 45 V (typ.). The
charging current ensures a quick start-up independent from the voltage applied to the HVSU
pin. Two different currents are generated depending on VCC voltage and the lower value is
generated when VCC is below the Vcc,su threshold. At the first start-up the VCC,su
threshold assumes its lower value (around 2 V).
Figure 3. Supply management unit
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The VCC management unit consists of three comparators (Figure 3). The lower threshold is
the Vcc,shd: this comparator is responsible to reset all timing and protection information
when VCC is lower than this voltage.
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HVLED001 features description
As soon as the VCC voltage reaches the turn-on threshold (Vcc,on), the HVSU is turned off
and pull-up sources of the pins FB, TOFF and CTRL are turned on. If the above mentioned
pins are not externally pulled down, the switching activity will start as soon as both the
operative conditions are reached (FB and CTRL above relevant disable thresholds) and
VCC is pulled up again to the Vcc,on threshold.
Figure 4. First start-up time waveform
VHVSU
VHVstart
Icharge
Icharge
IHV,su
VCC
Vcc,on
Vcc,su (at st-up)
ICC
Icc (swithing)
Icc (stop)
Iq
Start-up
Low cons.
Active mode
CTRL
VCTRL,dis
FB
VFB,dis
Switching
activity
AM039707
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HVLED001 features description
AN4710
During the switching activity the HVSU is kept off, in fact the IC is expected to be supplied
by the energy kept from either an auxiliary winding of the main magnetic component
(transformer or inductor) or an auxiliary converter. To be noted that any kind of supplying
mean (including auxiliary winding) must be decoupled from the VCC pin using a general
purpose diode having the proper reverse voltage rating: this good practice is necessary to
preserve the mutual functionality of HVSU and VCC.
If the supplying energy is insufficient the VCC voltage may drop below the Vcc,su threshold;
at this occurrence the switching activity is stopped and the device enters the low
consumption mode. Contemporarily the CTRL pin is internally pulled down: the CTRL
voltage could then be used to signal this state to external supervisors or to disable external
circuits.
During the low consumption mode the HVSU is active to maintain the VCC voltage between
VCC,on and VCC,su.
An internal clamping device is connected to the VCC pin to prevent spurious VCC
fluctuation from damaging internal structures.
1.4
Peak current mode definition
[Involved pins 1: HVSU, 4: FB, 5: CTRL, 7: CS]
1.4.1
Multiplier
The core of the HVLED001 device is a peak current mode controller that provides the turnoff command to the gate driver when the MOSFET's source current reaches a threshold
(VCS) provided by a multiplier block as per the following equation.
Equation 1
VCS  k p 
VHVSU
V
 VFB  VFB,ref 
HVSU,pk
At first the input voltage, connected to the HVSU pin, is scaled by an internal equivalent
voltage divider. Then it is normalized dividing the input voltage by its maximum value (VFF);
as a result the term VHVSU / VHVSU,pk is either a half sinusoid having amplitude of 1 V or
a unity DC voltage if the HVSU pin is connected respectively to a rectified mains (any value
between typ. 32 VAC to 305 VAC) or to a DC voltage (e.g.: PFC output).
The peak of the input voltage is obtained by a peak detector able to quickly react to an
abrupt mains change thanks to a proprietary internal structure able to operate without any
external storing element (capacitors). A positive going variation of the input voltage is
immediately processed by the peak holder block, while a negative going variation is
detected within typically 3 half sine waves and results into a limited variation (in time and
amplitude) of the application's output (current or voltage).
A detail of the blocks involved in this feature is shown in Figure 5.
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AN4710
HVLED001 features description
Figure 5. Multiplier block diagram
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The voltage present on the FB pin, representative of the energy required by the load, is
firstly purged by the term VFB,ref and then applied to another input of the multiplier.
Finally the multiplier block adapts the result of the product between the voltages
representative of input voltage and load energy to a maximum level for current sense input
equal to VCS,lim (around 0.75 V). The overall scaling factor of the multiplier is equal to kp
and is reported in the product datasheet. The linearity of the multiplier block is guaranteed
over a range of input voltages between 0 V and 480 Vdc. Higher input voltages may result
into a clipping of the multiplier results, but without affecting the safe operation of the device.
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HVLED001 features description
1.4.2
AN4710
Current sense comparator system
The internal current sense comparator compares the output of the multiplier with the voltage
present at the CS pin. In a typical application the CS pin is connected to a shunt resistor
(RCS) placed between the source of the MOSFET and common ground.
A summary of the structures involved into peak current detection is shown in Figure 6.
Figure 6. Current sense comparators system
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The maximum value of VCS is also related to the voltage present at the CTRL pin as per
following equation:
Equation 2
VCS,lim  0.366  VCTRL  0.016

VCTRL  VCTRL,dis , Veoss 
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HVLED001 features description
Figure 7. VCS,lim vs. Vctrl characteristic
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The value of RCS has to be selected in order to obtain VC,lim at a full load, i.e.: at
a maximum MOSFET's peak current Ipkp,max. The value of the Ipkp,max can be obtained
according to the implemented topology.
Equation 3
R CS 
VCS,lim
Ipkp,max
An internal LEB structure prevents the loop from reacting to gate driver's turn-on spikes;
when a proper design of the PCB layout is made, no further filter structure should be
necessary.
A minimum level of VCS is also present to guarantee a minimum value for the MOSFET
drain's current: this minimum current helps to reduce the THD and to guarantee a minimum
value for the demagnetization time. This minimum level is proportional to the peak of input
voltage according to the following equation.
Equation 4
V CS min = V HVSU  0.15m
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HVLED001 features description
1.4.3
AN4710
Soft-start
The dependence of VCS,lim on CTRL pin's voltage can be exploited either at the start-up or
when low consumption ends: in fact it limits the peak of the current transferred to the load
(inrush limit). The duration of this limitation is determined by the time the CTRL takes to
reach its steady value, i.e.: by the size of the capacitance (Css) placed between CTRL and
GND.
Figure 8 illustrates the equivalent model of the internal structures connected to the CTRL
pin that are involved in soft-start function and the suggested circuitry to activate this feature.
Figure 8. CTRL pin structures to be used for soft-start purpose
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Until the CTRL voltage is below VCTRL,dis the HVLED001 device is disabled. After the
device’s start and until CTRL pin's voltage is lower than Veoss many features are disabled:
namely the THD optimization and the timed protections. In fact, during the first operation
instants the control loop is providing the maximum power to the output in order to reach the
set value of the output variables, but this condition, without a proper masking, may trigger
said protections.
During the normal operation, after CTRL voltage rises above Veoss, all protections and
features are active and the CTRL voltage can be optionally reduced by external circuitry to
limit the maximum level of VCS,lim.
A proper choice of the optional resistor Rss allows to obtain a CTRL pin's voltage higher
than the disabling threshold immediately after the IC turn-on.
1.5
Smart zero current detection
[Involved pins 3: TOFF, 6: ZCD]
The ZCD pin of the HVLED001 device features the following functionalities:
14/63

Detection of the resonant's valleys associated with the transformer's demagnetization
instant to set the boundary mode conduction of the desired topology.

Measurement of the voltage present at the pin at the demagnetization instant, for the
primary side control purpose (described in Section 1.6.1 on page 23).

Measurement of the current sunk during the on time of the controlled MOSFET, for the
brownout protection purpose (described in Section 1.9.4 on page 32).
DocID027926 Rev 1
AN4710
1.5.1
HVLED001 features description
Demagnetization detection
The detection of the resonant valley is designed to implement the quasi resonant (QR)
mode of flyback or buck-boost topology. The QR mode consists on turning on the MOSFET
when the current into the primary side of the flyback transformer (or alternatively into the
inductor of the buck-boost) becomes zero. At this occurrence, the drain of the MOSFET
starts oscillating at a frequency set by the value of the inductance and the overall drain
capacitance.
Equation 5
fResonance 
1
2π CDrain  LPRI
The amplitude of the oscillation is given by the output voltage scaled by the primary-tosecondary turn ratio (or output voltage in case of buck-boost topology) and is superimposed
to input voltage. This oscillation (without DC component) is also present on the output of any
winding coupled with the primary inductance and then it can be fed to the ZCD pin, whose
internal block structure is illustrated in Figure 9.
Figure 9. ZCD detection circuitry
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The signals related to the zero current detection are illustrated in Figure 10.
Although the boundary mode operation is achieved when the MOSFET is turned on when
the demagnetization occurs, it is most convenient, in order to minimize the switching losses,
to turn-on the MOSFET to the minimum of the oscillation (valley switching).
The demagnetization instant (Tdemag) is captured by the DEMAG LOGIC block when the
derivative of the voltage becomes suddenly negative, while the valley switching (Tvalley) is
managed by the ZCD comparator waiting for the falling edge of the oscillation across zero
(VZCD,trig).
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HVLED001 features description
AN4710
VGD
Figure 10. ZCD related waveforms
Tdemag
Vdrain
Vin + VR
Vin
Vin - VR
Tvalley
IZCD,clamp
VZCD
Vaux
3 μs
VZCD,arm
VZCD,trig
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1.5.2
HVLED001 features description
SMART ZCD detection
The HVLED001 device is able to distinguish an effective demagnetization from abnormal
signals.
The HVLED001 considers an energy transfer phase correctly ongoing if the ZCD signal is
higher than an arming threshold (VZCD,arm) 3 µs (TBLANK,min) after the MOSFET turnoff. This strategy helps rejecting the high frequency oscillations associated with the leakage
inductance that normally ends in a couple of microseconds after the MOSFET turn-off.
This check is performed when the value of the Vcs threshold (or the output of the multiplier)
is higher than 300 mV (typ.). If a correct energy transfer occurs, the first falling edge
(VZCD,trig) subsequent the blanking time (defined by the voltage at the pin TOFF) is used
to trigger the MOSFET's turn-on instant.
At the start-up, when no ZCD signal is present, a 500 µs timer (starter) provides a triggering
signal for GD turn-on. The same unit provides a restarting attempt in case of absence of
a valid ZCD signal (e.g.: during short-circuit).
1.5.3
Frequency foldback
When the output power demand diminishes, the duty cycle of the MOSFET's activity is
reduced. This duty cycle reduction can be obtained either reducing the MOSFET's on time
interval or increasing the off time interval entering into the discontinuous conduction mode.
The valley skipping technique is an efficient approach to obtain both a longer off time and to
turn-on the MOSFET with a lower drain voltage. The resulting operating frequency is folded
back to a lower value. The HVLED001 performs this “valley skipping” technique adding
a further blanking time to TBLANK,min. The increase of the blanking time is adjustable up to
200 µs varying the voltage applied to the TOFF pin as described in the characteristic
illustrated in Figure 11 (typical values). The MOSFET is turned on in correspondence of the
first valley occurring after the end of the additional blanking time, while a proprietary
structure ensures a correct turn-on of the MOSFET's also after the oscillation amplitude is
fully decayed.
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AN4710
Figure 11. TBLANK vs. TOFF voltage characteristics
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The behavior of the HVLED001 in correspondence of the different TBLANK setting is also
illustrated from Figure 12 to Figure 15 on page 22.
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HVLED001 features description
Figure 12 is the case when TOFF voltage is higher than 2 V: the TBLANK is lower than
minimum blanking time and the first valley is used as a trigger signal to turn-on the MOSFET
gate driver.
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HVLED001 features description
AN4710
Figure 13 is the case when TOFF voltage is lower than 2 V, but the selected TBLANK is
shorter than the demagnetization time: the first valley subsequent to TBLANK is the first
valley and used as a trigger signal to turn-on the MOSFET gate driver.
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AN4710
HVLED001 features description
Figure 14 is the case when TOFF voltage is lower than 2 V, and the selected TBLANK is
longer than the demagnetization time: the first valley subsequent to TBLANK is used as
a trigger signal to turn-on the MOSFET gate driver resulting in valley skipping.
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Figure 14. Example: TOFF voltage lower than 2 V and resulting in valley skipping
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HVLED001 features description
AN4710
Figure 15 is the case when TOFF voltage is lower than 2 V, and the selected TBLANK ends
after the drain oscillation almost extinguished its amplitude: being the ZCD level above the
arming threshold after the very first 3 µs (Tblank,min), the HVLED001 waits for a triggering
signal, during the following 3 µs timeout. When the timeout elapses the internal starter
forces the gate driver turn-on.
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Figure 15. Example: TOFF voltage lower than 2 V with extinguished drain's resonance
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1.6
HVLED001 features description
Control loop
[Involved pins 4: FB, 6: ZCD]
The HVLED001 is able to operate either with an optocoupler based control loop or a primary
side regulated control loop (constant output voltage only).
The FB pin can be used respectively as a pull-up current generator for optocoupler output
and as output of the primary side regulator operational amplifier. In both cases a part of (or
whole) the compensation network is connected between the FB and GND pin. A block
diagram of the blocks involved in closing the control loop is depicted in Figure 16.
1.6.1
Primary side regulation
When the entire energy stored in the transformer is transferred to the load, the rectifier
current reaches zero as well as the voltage drop across the said diode.
As a consequence, the voltage across the secondary side at the demagnetization instant is
exactly equal to output voltage; the same voltage is also present across all other windings,
scaled by the relevant transfer ratio, in particular the auxiliary winding on the primary side is
monitored by the ZCD pin.
Figure 16. PSR management logic
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The ZCD pin DEMAG LOGIC acquires this value and feeds it to the inverting pin of the
internal error amplifier (E/A) to close the control loop.
The E/A is an operational transconductance amplifier (OTA) designed to operate either with
narrow or wide bandwidth so that the HVLED001 is able to control equally a high power
factor topology or a DC/DC topology (e.g.: flybacks fed by PFC pre-regulators).
The non-inverting pin of the E/A is connected to a high precision reference voltage (2.6 V).
The output of the E/A is connected to the FB pin, where the suitable compensation network
can be placed, referred to the common potential (GND); an equivalent small signal model of
this component, useful for a compensation network definition, is illustrated in Figure 17,
while the OTA characteristic curve is illustrated in Figure 18.
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HVLED001 features description
AN4710
Figure 17. OTA output equivalent model
Figure 18. OTA output characteristic
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The output voltage detector (S&H), driven by the demagnetization, is electrically equivalent
to a low pass filter having a time constant approximately equal to 500 ns. This behavior
could worse the output voltage regulation and can be reduced introducing either
a derivative term, having almost the same time constant, into the voltage divider connected
between the auxiliary winding and ZCD pin or reducing the equivalent resistance of the
voltage divider itself.
The simplest way to implement the derivative term consists on replacing the principle
voltage divider illustrated in Figure 19 (a) with a more effective structure, illustrated in
Figure 19 (b), that contains a reset diode that makes the derivative action independent from
the auxiliary voltage value during on time.
Alternatively the voltage divider with reduced equivalent resistance is illustrated in Figure 19
(c). The brownout protection prevents the use of low values of the upper resistor during on
time, so a bypass diode must be placed to differentiate between on time upper resistance
(Rzcd_bo + Rzcd) and off time upper resistance (Rzcd only). In this arrangement is highly
suggested to select a value for the lower resistance approximately equal to 2.7 k.
Figure 19. ZCD voltage divider settings
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1.6.2
HVLED001 features description
Secondary side regulation
When an optocoupler is used to transfer the error information from the error amplifier placed
on the secondary side the FB pin can be used as a pull-up for the photo transistor of an
optocoupler connected between FB and GND.
The pull-up current (around 1 mA) is active when the PSR error amplifier saturates high
(see Figure 18 illustrating the E/A output characteristics). To enter this mode of operation it
is sufficient to set the PSR regulated voltage to a value that is higher than the maximum
operating one.
Both the internal pull-up and the optocoupler output are ideal current sources; therefore they
cannot drive the same node without any compensation impedance. For this reason, besides
a noise canceling capacitor (CP, normally in the range 1 to 10 nF), a degeneration resistor
(RP) has to be connected between the FB pin and GND. Depending on the desired
bandwidth this resistor's value can be selected in a typical range between 4.7 k up
to 47 k.
If the degeneration resistor has a value lower than:
Equation 6
RP 
VFB, dis
0.75

 18.75k 
IFB, src (@lowconsu mption)
40μ
The HVLED001 risks to remain stuck in the low consumption mode during the start-up or
the low consumption mode itself. To prevent this occurrence a general purpose diode has to
be placed in series with the resistor (with an anode connected to the FB pin).
Figure 20. Optocoupler operation typical arrangement
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HVLED001 features description
1.6.3
AN4710
Burst mode
During the normal control operation the FB pin voltage is set (by the loop balance) between
VFB,bm (lighter load) and VOFP (heavier load).
When FB voltage drops below VFB,bm, i.e.: the load is very light, the burst mode operation
is entered: until the FB voltage stays in this condition, the controller generates a series of
4 GD pulses every 1 ms. The distance between each pulse is equal to minimum TBLANK
(3.0 µs).
In this phase the minimum power delivered to the transformer's outputs equals:
Equation 7
PBM 
V
1
4
1
2
 L PRI  Ipkp

  L PRI   CS,min
2
1ms 2
 R CS
2

4
 
 1ms

A corresponding minimum load has to be guaranteed on the transformer's outputs to
dissipate this minimum energy during no-load application and to prevent the output voltage
to increase without limitation.
1.7
Gate driving
[Involved pins 9: GD]
The gate driver consists on a push-pull stage whose output is able to drive an external
MOSFET with the 300 mA source and 600 mA sink capability.
To avoid undesired switch-on of the external MOSFET an internal pull-down circuit holds the
pin low. This circuit guarantees 2 V maximum on the pin (at Isink = 2 mA) when VCC is
below the Vcc,shd threshold. This allows omitting the “bleeder” resistor connected between
the gate and the source of the external MOSFET used for this purpose.
1.8
Disabling features
[Involved pins 4: FB, 5:CTRL]
The HVLED001 embeds a set of inputs able to externally control the device activity.
1.8.1
Instant disable by CTRL
The device can be disabled entering into the low consumption mode since the CTRL pin is
pulled below the VCTRL,dis threshold. At this occurrence the device becomes immediately
inactive and the internal structures are put in the low consumption mode. Both pull-ups
embedded into the FB pin and TOFF pin remains active while the driver drives the MOSFET
off.
The response time of the internal comparator is very fast acting on the device status in less
than 200 ns. A small hysteresis (50 mV typ) is also present to improve the noise immunity of
this comparator. During the inactive state, the HVSU block guarantees that the VCC
remains within the range VCC,su … VCC on. The power demand in this condition is
approximately equal to 120 mW at Vin = 480 Vdc.
When the disabling signal is released, the HVSU charges the VCC pin and then the
switching activity starts from the start-up state (Figure 21).
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HVLED001 features description
Figure 21. Disabling HVLED001 using CTRL pin
CTRL
IC enabled
VCTRL,bias
IC disabled
VCTRL,dis
VCC
VCC > Vcc,on
Vcc,on
Vcc,su
Start-up
A.M.
Low consumption
Active mode
Switching
activity
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1.8.2
AN4710
Instant disable by FB
The device can be disabled entering into the stop mode since the FB pin is pulled below the
VFB,dis threshold. In this condition the IC's functional blocks remains fully operating (except
for the GD driver that remains constantly in low position) and therefore the supply current is
unchanged and HVSU is inactive.
When the FB pin is released above the threshold the device restarts immediately entering
the active mode state.
This disabling mean has been designed to interrupt the device activity for short periods; in
fact the VCC hold-up time in this condition is shortened due to IC power demand.
In case the VCC drops below Vcc,su, the IC enters low consumption; as a consequence the
power consumption is lowered, the HVSU is activated, the CTRL pin is internally pulled
down and the IC behaves as explained in the previous paragraph (Figure 22).
Figure 22. Disabling HVLED001 using FB pin
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1.8.3
HVLED001 features description
Timed disable by CTRL
The device can be also disabled when the CTRL pin is constantly kept above the threshold
named Vadis for a time longer than 100 ms.
After this time the IC enters the low consumption mode activating the HVSU and the CTRL
pull down resistor. It is then restarted, from the start-up state, after 2.5 s (typ.).
If the CTRL pin drops below Vadis before the 100 ms the counting timer is reset to zero
(Figure 23).
Figure 23. Timed disable of HVLED001 using CTRL pin
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1.9
Protections
1.9.1
Input overvoltage and surge protection (IOVP)
[Involved pins 1: HVSU]
During the normal operation as the offline converter, the HVLED001 can be subjected to
voltage variations, bursts and surges whose amplitude can be easily greater than 600 V.
During the off time, the voltage of the MOSFET' s drain is the sum of the input voltage and
the reflected voltage: the input overvoltage protection (IOVP) immediately (200 ns typ.)
interrupts the switching activity of the HVLED001 device when the input voltage is higher
than Vsurge. As a result, the rating of both - the MOSFET and the secondary side diode is
less critical than in standard controllers.
When the protection is triggered the HVLED001 device enters the stop mode. If the HVSU
voltage falls below the threshold within 10 ms, the switching activity is immediately restored,
otherwise the controller enters the low consumption mode until the HVSU voltage is
reduced. In the last mode of operation the internal high voltage start-up unit maintains the
HVSU supplied and, contemporarily, slightly loads the input capacitor (Figure 24).
Figure 24. Input overvoltage protection (IOVP) waveform
VHVSU
Overvoltage
Vsurge
Without iOVP
HVSU on
VDRAIN,pk
VHVSU+VR
VR
VCC
VCC,on
VCC,shd
Active mode
Switching
activity
STOP
Low cons.
Active mode
10 ms
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1.9.2
HVLED001 features description
Overload and optocoupler failure management (OFP)
[Involved pins 4: FB]
There are some abnormal conditions that make the FB pin voltage to rise until the output of
the internal E/A saturates. Such conditions are namely:

Overload

Short-circuit

Optocoupler failure
While in the first two cases the E/A reacts to deliver more power to the output in order to
recover a loss of output voltage, in the third case is more critical, especially when the LED
string is directly connected to the application's output.
The LED load behaves as a voltage limiter so that an optocoupler failure results into an
increase of the output current without an increase of the output voltage: without a dedicated
protection the LED string would be destroyed.
The optocoupler failure protection (Figure 25) embedded into the HVLED001 stops the
switching activity and enters the low consumption mode if FB pin's voltage remains above
the OFP threshold for a time longer than 100 ms. The switching activity is auto restored after
2.5 s, during which the IC is supplied by the high voltage start-up unit.
Figure 25. OFP protection
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1.9.3
AN4710
Short-circuit detection
[Involved pins 6: ZCD]
In case of short-circuit on the output connector of the application, theoretically, the
amplitude of the ZCD signal is insufficient to arm and trigger the ZCD comparator.
Unfortunately the primary side's peak current, quite high due to the reaction of the control
loop, magnetizes the leakage inductance that, oscillating with drain's capacitance,
generates a signal whose amplitude is higher than the arming threshold.
To prevent a false triggering of the ZCD comparator by those spurious oscillations, the
arming threshold is checked after TBLANK,min. If the ZCD signal is lower than the arming
threshold, then the starter (500 µs) is invoked. This approach minimizes the stress of the
secondary side rectifier reducing the risk of hard switching.
A constant short-circuit condition either triggers the OFP protection or results in an
insufficient VCC power supply.
1.9.4
Brownout
[Involved pins 6: ZCD]
Brownout protection is intended to stop the switching activity if the input voltage is
constantly lower than a desired level. The input voltage is monitored by the ZCD pin using
the current sunk from its internal negative clamp during the on time.
During the on time the auxiliary winding is expected to provide a negative voltage whose
amplitude is proportional to the input voltage, scaled by the turn ratio between primary and
auxiliary winding (NAUX/NPRI).
In this condition, the upper resistance of the voltage divider that is connected to the ZCD pin
generates a current that is compared with a threshold value (100 µA):
Equation 8
IZCD 
NAUX
NPRI
R zcd
Vin 
A free running timer is reset anytime the current is higher than the threshold: when the count
reaches 100 ms (i.e.: the current has never been sufficient to reset it) the brownout
condition is met and the IC is stopped in the low consumption mode. After 2.5 s an internal
auto-restart mechanism restores the switching activity (Figure 26).
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HVLED001 features description
Figure 26. Brownout protection
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1.9.5
Magnetic saturation or rectifier short-circuit
[Involved pins 7: CS]
In case of either saturation of the primary side, secondary side rectifier short-circuit or the
continuous mode operation the voltage across the current sense pin quickly rises above the
threshold set by the multiplier. At this occurrence a second level comparator interrupts the
switching activity for 1 ms before restarting the switching activity. The inactive mode
associated with this protection is a special case of the stop mode.
1.9.6
VCC short-circuit protection
[Involved pins 1: HVSU, 10: VCC]
The overload or short-circuit on the VCC pin prevents the internal high voltage start-up unit
to rise the VCC voltage from 0 V. On the other hand a continuous operation of the charging
unit at the maximum current may result into permanent damage of the device.
For this reason, if the VCC is lower than 2 V, then the charging current of the high voltage
start-up unit is reduced to a safe value.
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Designing a high power factor flyback LED driver
2
AN4710
Designing a high power factor flyback LED driver
This section describes how to design an LED driver based on the single stage flyback
controlled by the HVLED001. Guidelines on how to correctly design the transformer to
obtain robust performances is presented as well as the control loop analysis of both - CC
and CV (PSR) loops are illustrated.
2.1
Selecting the design input specifications
At first the target specification of the application needs to be defined. A list of this
specification is reported in Table 2.
Table 2. Typical design specification list
Design specs.
Parameter Unit
Description
The range of this value can be defined in terms of the single range
(e.g.: 198 265), wide range (100  264) or universal range
(90  305). Within each range is very common to distinguish
VAC a narrower range of values where the optimal performances are
guaranteed.
The class of the components of the input filter is mainly affected by
this parameter.
Mains voltage
Vmains
Mains frequency
Fmains
Hz
The mains frequency is nominally 50 Hz or 60 Hz and is normally very
precise.
Output LED current
ILED
mA
The average current to supply the LED.
Output current ripple
ILED%
%
Amplitude of the ripple superimposed to the LED current ILED
expressed as a percentage.
In such kind of application it is quite large (± 30% or more).
LED forward voltage
VLED
V
LED total forward voltage. The range of this parameter has normally
a ratio between maximum and minimum value around 3 or 4.
LED dynamic resistance
Rd_LED

LED dynamical resistance. It is the inverse of the slope of the LED's
current/voltage curve and plays an important role in both
compensation and output capacitor selection.
Output voltage
Vout
V
Maximum regulated output voltage during the open circuit.
Expected efficiency

%
The efficiency of the application: its main contributors are the
transformer and the power semiconductors (input rectifier bridge,
MOSFET and secondary side rectifier).
Ambient temperature
Tamb
°C
This information is important to select the class of the electrolytic
capacitor and the size of the heat sinks (where needed).
Some additional parameters can be defined starting from the specifications.
In particular the first important step is to select the value for the reflected voltage:
Equation 9
VR  Vout  Vf  
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NPRI
NSEC
AN4710
Designing a high power factor flyback LED driver
This term is the voltage present at the primary side of the transformer when the MOSFET is
off. Its value affects the MOSFET's breakdown voltage, the MOSFET switching losses and
the input current distortion (THD, see Appendix A: Theory of QR flyback topology fed by an
AC line on page 56).
In order to obtain both - good efficiency (> 90%) and low THD (typ < 10% at Vmains max.),
a reflected voltage equal to VR = 240 V is suggested.
Higher reflected voltage leads to better efficiency and THD, but the MOSFET's breakdown
voltage increases and the PSR regulation is less accurate and vice-versa.
Given the reflected voltage is convenient to solve some preliminary equations to obtain both
auxiliary parameters and operating voltages and currents.
Table 3. Preliminary equations
Eq. no.
Description
Equation 10
Equation 11
KV 
sinθ 
1
F1K V  

1  K V sinθ  π
Equation 12
F2K V  
Equation 13
F3K V  
Equation 14
Equation 15
Equation 16
Equation 17

π
0
sin2 θ 
1

1 K V sinθ  π
sin3 θ 
1

1  K V sinθ  π
N

0.5  1.4  10 3  K V
sin2 θ 
dθ 
1 K V sinθ 
1 0.815  K V
Refer to AN1059 for
detailed description
0.424  5.7  10 4  K V
sin3 θ 
dθ 
1  K V sinθ 
1  0.862  K V
Refer to AN1059 for
detailed description
π
NPRI
VR

NSEC (Vout  Vf)
Vout  ILED
η
Pin 
Maximum input
power
2  Pin
Vinpk  F2K V 
Peak value of the
primary side current
IDCp 
1
Ipkp  F1K V 
2
DC value of the
primary side current
IRMSp  Ipkp  K V 
Equation 20
Transformer turn
ratio
Ipkp 
Equation 18
Equation 19
Auxiliary parameter
Refer to AN1059
(on www.st.com) for
detailed description
0
0
Vinpk
VR
sinθ 
0.637  4.6  10 3  K V
dθ 
1  K V sinθ 
1  0.729  K V

π
Note
Ipks 
IDCs 
F2K V 
3
2  ILED
K V  F2K V 
1
Ipkp  F1K V 
2
DocID027926 Rev 1
RMS value of the
primary side current
Peak value of the
secondary side
current
DC value of the
secondary side
current
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Designing a high power factor flyback LED driver
AN4710
Table 3. Preliminary equations (continued)
Eq. no.
Description
Note
Equation 21
IRMSs  Ipks  K V 
Equation 22
2.2
F3K V 
3
IACi  IRMS i2  IDC i2
i  p, s
RMS value of the
secondary side
current
AC value of primary /
secondary side
current
Transformer design guide lines
Given the preliminary calculation of operating parameters, the proper design of the flyback
transformer can be accomplished.
2.2.1
Primary inductance selection
Usually the transformer's primary inductance is selected to set the minimum switching
frequency to a suitable value.
Using the HVLED001 device a further constraint has to be considered at first to properly
operate the smart ZCD detection (see Section 1.5.2: SMART ZCD detection on page 17).
It must be guaranteed that the demagnetization time is longer than 3 µs when the output of
the multiplier is higher than 0.3 V, otherwise a false missed magnetization condition is
detected and an abnormal operation is obtained (500 µs of the inactive state is observed).
The following equations find the relation between the demagnetization time, output of
multiplier and primary inductance:
Equation 23
Tdemag 
L SEC  Ipks
Vout  Vf
Rewritten the previous equation using the turn ratio a primary side rated version can be
obtained:
Equation 24
Tdemag 
1 LPRI  Ipkp LPRI  Ipkp

N Vout  Vf 
VR
The peak of the primary current can be easily related to the CS pin voltage using the value
of the current sense resistor.
Equation 25
Tdemag 
LPRI  VCS
VR  R CS
The value of RCS is selected to exploit the entire VCS dynamic: therefore its value can be
expressed as the ratio between the maximum value of VCS (VCS,lim = 750 mV) and the
maximum value of the primary current (at maximum load and minimum input voltage)
Ipkp,max.
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Designing a high power factor flyback LED driver
Equation 26
R CS 
VCS, lim
Ipkp, max
Substituting Equation 26 into Equation 25, the demagnetization time can be written using
known parameters.
Equation 27
Tdemag 
LPRI  VCS  Ipkp, max
VR  VCS, lim
Re-arranging the terms and considering that 0.3 V / VCS,lim = 0.4, the minimum primary
inductance, given a desired reflected voltage and the maximum primary current, can be
written as:
Equation 28
LPRI,min 
3μ  VR
Ipkp, max  0.4
The equation is also expressed as nomogram as illustrated in Figure 27.
Figure 27. Minimum primary inductance nomogram
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The value of the Ipkp,max can be obtained using Equation 16 with maximum input power
and minimum input voltage.
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Designing a high power factor flyback LED driver
2.2.2
AN4710
Core size selection
The size of the flyback transformer can be differently approached to optimize different
parameters. Besides the different techniques, a valid starting point to select core's size is
the use of the area product calculation.
Two expressions for determining the minimum required core area-product (winding window
area times effective magnetic cross section in cm4) will be provided:
Equation 29
APmin


460  Pin, max


 fsw, min  1  K V   F2K V  
1.316
Equation 30
480  Pin max
AP min = -----------------------------------------------------------------------------fsw min   1 + K V   F2  K V 
1.585
2 0.66
  J H  K V   fsw min + J E  K V   fsw min 
where JH (KV) and JE (KV) are functions related to hysteresis and eddy current losses,
whose best fit approximation are respectively:
Equation 31
JH K V  
1.87  1.26  K V
 10  5
1  0.55  K V
JE K V  
1.88  1.06  K V
 10 10
1  0.34 K V
Equation 32
Equation 29 assumes that the maximum peak flux density inside the core is limited by core
saturation and that all transformer losses are located in the windings; Equation 30 assumes
that core losses limit the flux swing and the total dissipation are half due to core losses and
half to windings losses.
Common to both formulas are the following assumptions:
1.
the material is a typical power ferrite with a saturation flux density above 0.3 Tesla;
2.
the windings occupy 40% of the total window area to leave space for isolation layers,
creepage and clearance distances;
3.
primary and secondary winding wires are proportioned for equal RMS current density;
4.
core and/or copper losses result in 30 °C hot spot temperature rise (no forced cooling);
5.
skin and proximity effects are neglected, considering the frequency range involved.
For a given fsw,min, one should try both formulas (considering KV at minimum line voltage)
and use the higher resulting value.
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2.2.3
Designing a high power factor flyback LED driver
Turn selection
The number of primary side turn can be written considering the maximum allowable
magnetic flux (Bmax) and the core's effective area (listed into magnetic core datasheet).
Equation 33
NPRI 
LPRI  Ipkp, max
Bmax  Ae
The primary-to-secondary side turn ratio is defined by reflected voltage selection (Equation
14 on page 35), while the primary-to-auxiliary side turn ratio is selected based on both
power supply and ZCD triggering considerations.
Auxiliary winding can be coupled accordingly to primary side (flyback configuration) or to
primary side (forward configuration).
The first arrangement can be adopted when the output voltage range is small and the PSR
operation is needed to control Vout:
Equation 34
VLED,min VZCD,arm

Vout
Vref,PSR
In this case the auxiliary turn number has to be set in order to obtain a Vaux higher than
Vcc,shd when the LED voltage is minimum. Often a simple linear regulator is beneficial to
set the VCC voltage within the pin's voltage allowed range.
Equation 35
NPRI VLED,min NPRI


NAUX
Vaux
NSEC
When the output voltage range is wider than calculated in Equation 34 the ZCD signal must
be obtained using an alternate structure and/or the VCC has to be obtained connecting the
auxiliary winding in forward configuration.
In case of forward configuration auxiliary voltage variation has the same variation of the
input voltage and a linear regulator can be used, even in universal mains application, to limit
the VCC excursion. Being the rectified mains a half sinusoid, is preferable to consider, as
minimum input voltage for the turn ratio selection, the RMS value of the input mains rather
than its peak value. Doing so, some margin is automatically taken into account.
Equation 36
NPRI VinminRMS

NAUX
Vaux
Different winding arrangement can be done: one arrangement example that reduces the
primary side leakage inductance and gives good regulation facts is illustrated in Figure 28.
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AN4710
Figure 28. Transformer's winding arrangement example
2.2.4
Transformer design equation summary
Table 4 summarizes the equations to be used designing the transformer.
Table 4. Transformer design equations
Parameter
Primary
inductance
Equation
Equation 28
LPRI,min 
3μ  VR
Ipkp, max  0.4
Equation 29


460  Pin, max
APmin  

 fsw, min  1  K V   F2K V  
Area product
(must be higher
than the maximum Equation 30
between the two)
480  Pin max
AP min = -----------------------------------------------------------------------------fsw min   1 + K V   F2  K V 
1.585
1.316
2 0.66
  J H  K V   fsw min + J E  K V   fsw min 
Equation 33
Primary side turn
number
“Pri to sec” turn
ratio
NPRI 
LPRI  Ipkp, max
Bmax  Ae
Equation 14 on page 35
N
NPRI
VR

NSEC (Vout  Vf)
Equation 35
NPRI VLED,min NPRI


NAUX
Vaux
NSEC
Flyback configuration
“Pri to aux” turn
ratio
Equation 36
NPRI VinminRMS

NAUX
Vaux
Forward configuration
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AN4710
2.3
Designing a high power factor flyback LED driver
ZCD network definition
The ZCD network's functions are listed in Table 1 on page 5.
2.3.1
ZCD network to implement PSR (simple resistive network)
The ZCD network suitable to regulate output voltage with PSR function using resistors only
is illustrated in Figure 29.
Figure 29. ZCD network (PSR - resistive)
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Equation 37
Rfb  2.7kΩ
(suggested value)
Equation 38
 Vout - Vfzcd NAUX 

 1
Rzcd  Rfb  
 VREF,PSR NSEC 
Vfzcd is the voltage drop of the diode in parallel with Rzcd_bo.
Equation 39
Rzcd_bo 
Vin 
NAUX
NPRI  Rzcd
IBO
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Designing a high power factor flyback LED driver
2.3.2
AN4710
ZCD network to implement PSR (network with derivative components)
The ZCD network suitable to regulate output voltage with PSR function including the
derivative contribution to compensate the sample and hold capacitive behavior is illustrated
in Figure 30.
Figure 30. ZCD network (PSR - derivative)
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This network needs some fine tuning, but can work with higher resistance values.
Equation 40
Rzcd 
Vin 
NAUX
NPRI
IBO
Equation 41
Rzcd_su  Rzcd
Equation 42
Csu 
1μs
Rzcd
Equation 43
Rfb 
42/63
Rzcd
Vout
NAUX 


 1

 VREF,PSR NSEC 
DocID027926 Rev 1
AN4710
2.3.3
Designing a high power factor flyback LED driver
ZCD network to operate without PSR
The ZCD network suitable to disable the PSR function is illustrated in Figure 31: the diodes
(standard type, like 1N4148) allows to generate a ZCD signal having amplitude that is
higher than VZCD,arm and lower than VREF,PSR. This configuration sets the FB pin to
operate as a constant current source to bias the optocoupler output.
Figure 31. ZCD network to disable the PSR
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Equation 44
Rzcd 
2.3.4
Vin 
NAUX
NPRI
IBO
ZCD network driven by MOSFET's drain
An alternate connection of the ZCD network consists on connecting it to the MOSFET's
drain through an RC network as illustrated in Figure 32.
Figure 32. Capacitive coupled ZCD network without PSR
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This network is useful when, due to the auxiliary winding forward connection, the auxiliary
signal is not available to detect the demagnetization's oscillations.
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Designing a high power factor flyback LED driver
AN4710
The CZCD is charged at Vin: in fact the network is supplied by an almost square wave
whose amplitude is equal to:
NPRI
Vin  Vout  Vf sec  
 Vin  VR
NSEC
and whose duty cycle is equal to (1 - D), where D is the MOSFET's duty cycle and equates:
Equation 45
D
VR
Vin  VR 
Therefore the average capacitor's voltage can be derived averaging this square wave:
Equation 46
VR 

VCZCD  Vin  VR    1 
  Vin
Vin  VR 

The resistor can therefore be selected to satisfy the following relation:
Equation 47
Rzcd 
Vin, max
IBO
In order to guarantee correct valley detection, the time constant of this structure must be
longer that the maximum expected damping time of the drain node oscillations. In practical
flyback circuitries this time is normally around 40 µs.
Equation 48
CZCD 
40 
Rzcd
This network is normally used with or without PSR and the value of Rlo can be found
considering that VR is present across voltage divider:
Equation 49

VPSR, REF


 Rlo  Rzcd  
 PSR is active

 N  Vout  VPSR, REF 
Rlo 2  Standard diode
PSR disabled

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Designing a high power factor flyback LED driver
2.4
Active components definition
2.4.1
Input rectifier bridge
The input rectifier bridge should be able to withstand a reverse voltage greater than the
input voltage peak's value.
Equation 50
Vbr_rev  2  Vinpk, max
An integrated or non-integrated bridge can be used depending on thermal and
manufacturing considerations.
The maximum value of the RMS current flowing into the bridge is equal to:
Equation 51
Ibr_f 
Pin
Vmains, min
The power losses associated with the bridge can be estimated (with margin) as:
Equation 52
Pbr  Ibr_f  2  Vf_br
Where Vf_br is the forward voltage drop of a single diode of the bridge.
2.4.2
Secondary side rectifier
The output rectifier bridge should be able to withstand a reverse voltage greater than:
Equation 53
Vbr  Vout  Vinpk, max 
NSEC
NPRI
The maximum current that flows in the diode is Ipks (Equation 19 on page 35).
The diode's power losses are:
Equation 54
Pdsec  Vth  IDCs  Rdsec  IRMSs
The terms Vth and Rdsec can be found in the diode's datasheet.
2.4.3
MOSFET selection
The MOSFET must have a breakdown voltage greater than:
Equation 55
Vbrdss  Vinpk, max  VR  Vspike
Vspike in Equation 55 is the allowed amplitude of the oscillations occurring between the
transformer's leakage inductance and drain capacitance.
The Rds,on of the MOSFET can be selected as the best trade-off between cost and power
losses performances.
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AN4710
The losses associated with the MOSFET are mainly dominated by conduction losses
(Pcond) when the input voltage is low, while they are dominated by capacitive losses (Pcap)
when the input voltage is high.
Equation 56
Pcond  Rds, on  IRMSp 2
The capacitive losses are associated with the discharge of the capacitance present at the
MOSFET drain. Therefore it is proportional to overall drain capacitance (Cdrain) and
switching frequency. The last parameter varies over the half sinusoid: the average value of
said frequency can be used to evaluate the switching losses.
Equation 57

1
Vinpk
1
fsw avg = ----  ----------------------------  --------------------------------- d
 L PRI  Ipkp 1 + K V  sin 
0
Equation 58
3
2

 2  Vinpk  VR   
 2  Vinpk  VR   2 1
Pcap  fsw, avg  3.3  Coss  
C



Drain 
 

π
2
π
 




The sum the of power losses contribution has to be considered to refine the MOSFET's
selection and the size of the eventual heat sink.
2.4.4
Clamping device selection
Two different clamping structures can be applied in parallel with the transformer's primary
side to limit the overvoltage spikes due to the leakage inductance of the transformer
(Vspike): the RCD clamp (Figure 33) and Transil™ based clamp (Figure 34).
Figure 33. RCD clamp
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Considering the RCD clamp, the capacitor is selected so as to have Vspike (as a rule of the
thumb, half the reflected voltage) at turn-off such that the voltage rating of the MOSFET is
never exceeded.
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Designing a high power factor flyback LED driver
From energetic balance, it is possible to write:
Equation 59
Ccl 
Llk  Ipkp, max 2
Vspike  Vspike  2  VR 
The term Llk in Equation 59 is the leakage inductance of the transformer. Normally, it can be
minimized between approximately 1% and 3% of LPRI. The capacitor undergoes large
current spikes and therefore it should be a very low ESR type with a polypropylene or
polystyrene dielectric film.
The minimum resistor value can be found by imposing that the voltage on the capacitor at
the beginning of each switching cycle never falls below the reflected voltage:
Equation 60
Rcl 
1
Vspike 

fsw, min  Ccl  ln 1 

VR 

The term fsw,min in Equation 60 can be derived by Equation 57 calculated when the
minimum input voltage is applied. The power rating of this resistor can be estimated by
considering the DC dissipation due to the reflected voltage and the leakage inductance
energy:
Equation 61
2
2
fsw min
VR
P  Rcl  = -----------------------  L lk  lpkp max   1 + K Vmin   F 2  K Vmin  + ----------Rcl
2
The diode will be rated for repetitive peak currents equal to Ipkp,max, and with a breakdown
voltage greater than Vpk,max + VR.
Considering the Transil based clamp, its clamping voltage can be approximated with its
breakdown voltage. In fact, the peak current is quite small and it is possible to neglect the
contribution due to the dynamic resistance. The breakdown voltage, which should account
for the drift due to the temperature rise, will then be:
Equation 62
Vbr_tsl  VR  Vspike
The steady-state power dissipation capability must be at least:
Equation 63
Vbr – tsl
fsw min
2
P  Tsl  = -----------------------  Llk  lpkp max   1 + K Vmin   F 2  K Vmin   ---------------------------------Vbr – tsl – V R
2
The rating of the diode follows the same rules of the RCD-based Transil structure.
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Designing a high power factor flyback LED driver
AN4710
2.5
Input and output filter definition
2.5.1
Output capacitor
The output capacitor undergoes the AC component of the secondary current. The current
ripple (similarly to voltage ripple) has two components: a high frequency ripple due to finite
ESR of the capacitor and the low frequency ripple due to the twice line frequency envelope.
The high frequency ripple can be expressed as (it has been supposed that ESR << Rd_LED
that is the LED dynamical resistance):
Equation 64
∆Io HF  Ipks 
ESR
Rd_LED
To calculate the low frequency ripple an auxiliary variable needs to be defined (see AN1059
for further details):






V


V

 
 
K
 
3
K
0
1 4
7
5 0
.
.
1
1
5 1
2
.
0

θ
d
θ
2 θ
s n
o i
c s
V
θ K
2
n 1
i
s

0
V

π
K
2
H

1 π
Equation 65
The low frequency ripple can be expressed as:
D
E
L
I



t
u
o
C
s
n
i
a
m
F


V
V

K
2
F
F
L

π
o
I
Δ
K
2
H
1
Equation 66

The Cout voltage rating must be higher than Vout plus some additional margin.
2.5.2
Input capacitor
The input capacitor undergoes the high frequency component of the input (line) current. The
goal is to prevent this high frequency from its transferring back to the line.
Once defined the maximum allowed percentage ripple (f_HF), the capacitor value can be
found as:
Equation 67
∆Iin HF 
1
1

2π r_HF  Pin  fsw, min
The rating of the capacitor must be at least higher than Vinpk,max.
An X2 type film capacitor is suggested.
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DocID027926 Rev 1
AN4710
Designing a high power factor flyback LED driver
2.6
Control loops definition
2.6.1
PSR control loop
In order to compensate the PSR loop, the open loop transfer function has to be derived
starting from its elementary components (see Section 1.6.1: Primary side regulation on
page 23). The compensation network will be placed between the FB pin and GND.
Figure 35. PSR control loop
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DocID027926 Rev 1
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63
Designing a high power factor flyback LED driver
AN4710
At first the plant transfer function can be written as:
Equation 68




 
 











S
1 C
R
m
k

t
tu
u
oo V
CC K
T Γ
RU
SO
1
ER
s s
1 1
T
U
O
R
N 2


V


V
D
O
M
K V
2 K
F Γ
1
K
G

t
l
u
M
S
P
 
G
s
G
s
p
A
 







The term Г(KV) in Equation 68 is equal to:

V

V

3
 

K
V

K
8
.
0
1
K
Γ

0
1
1
1
Equation 69
The plant transfer function has a low frequency pole and high frequency zero: the zero could
be neglected, being the narrow bandwidth a requirement to obtain a good PF and THD.
The effect of ZCD input circuitry can be written as:
Equation 70
GEA_in s   G ZCD

Rfb
1
 NAUX
 
 G SH (s)  


 NSEC Rfb  Rzcd   1  s 4π
fsw






Depending on the ZCD voltage divider, the Rzcd term assumes different means - please
refer to Section 2.3: ZCD network definition on page 41 for a complete description of the
possibilities.
The internal sample and hold introduces a pole whose frequency depends on the operating
frequency.
Finally the compensation network connected between the FB and GND allows obtaining the
desired bandwidth tuning the singularities position. Its transfer function is:
Equation 71
GFB s   gm  ROTA 
1  s  RS  CS
1  s  ROTA  CS   1  s  RS  CS  COTA 
ROTA and COTA have been introduced in Figure 17 on page 24 and are respectively equal
to 2 M and 350 pF.
HINTS:
It can be convenient to set the zero's frequency equal to the low frequency pole of the plant
transfer function and act on the poles of the compensation network to obtain the desired
bandwidth.
In case the PSR loop is used as an output voltage limit of a constant output current
application, the bandwidth is wider than in a HPF application. This aspect has to be taken
into account when setting the zero/pole position.
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AN4710
2.6.2
Designing a high power factor flyback LED driver
Optocoupler based control loop (constant current)
In order to compensate the loop based on the optocoupler (constant output current), the
open loop transfer function has to be derived.
The plant transfer function can be written as:
Equation 72


 

︳
 











S
1 C
R
m
k

t
u
to
u
oC V
CD K
RE Γ
SL
Ed 1
sR
1 s
1
N 2


V


V
D
O
M
K V
2 K
F Γ
1
K
G

t
l
u
M
S
P
 
G
s
G
s
p
A
 







The error amplifier section is on the secondary side and includes the relevant compensation
network [GEA(s)] the optocoupler and relevant biasing means [Hp(s)] as illustrated in
Figure 36.
Figure 36. Secondary side error amplifier arrangement
Equation 73
Rsense
RP  ROTA
Hp s  
 CTR 
RBias
RP  ROTA




1


 1  s  RP  ROTA  CP  COTA  
RP  ROTA


The pole of Equation 73 is normally set on relatively high frequency, but the DC gain of said
frequency represents an important tuning factor to set the bandwidth to a low value.
The RP can eventually be placed in series with a diode if its value is lower than
VFB,dis / IFBpu = 15 k.
The compensation network is finally designed according to its transfer function:
Equation 74
GEA s  
1  s  R1  R2   C2
s  C2  R2
The zero of GEA can be used to cancel the plant transfer function's low frequency pole.
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Designing a high power factor flyback LED driver
2.7
AN4710
Soft-starting the application
When the application is turned on, the output capacitor needs to be charged to reach the
operating voltage. During this phase the control loop is saturated high to transfer the
maximum energy to the secondary side resulting into two main effects: the initial rectifier
current is very high (inrush current) and an overshoot appears due to the narrow loop's
bandwidth.
The capacitor connected to the CTRL pin can be used to limit the inrush current and,
eventually, to reduce the overshooting by means of the reduction of the maximum primary
side current. It is convenient to add a resistor in series with the capacitor as illustrated in
Figure 8 on page 14 and Figure 37 to bypass the CTRL disable feature (Section 1.8.1:
Instant disable by CTRL on page 26).
Figure 37. CTRL pin biasing
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Said RC network can be designed as follows:
Equation 75
Rss 
VCTRL, dis
Ictrl, bias
The soft-start phase is accomplished when CTRL voltage reaches Veoss. The size of Css
can be selected to obtain the desired soft-start time (Tss):
Equation 76
Css 
52/63
Tss
Rctrl  Rss 

1

Rss  
Veoss  
 ln  1 

  1 
Rctrl  
Vctrl, bias  

DocID027926 Rev 1
AN4710
2.8
Designing a high power factor flyback LED driver
Supplying the application
The device has to be supplied by a generator having an average current capability higher
than operating ICC: normally the auxiliary winding of the transformer is sufficient (selfsupplied applications), but different requirements may need the implementation of separate
controllers (separate supplied applications - e.g.: when the application belongs to a complex
subsystem). Whatever the supplying method is adopted, a decoupling diode between the
supplying source and the VCC pin is required for a proper functionality of the HVSU.
The present section is focused on self-supplied applications only. The different load
characteristics lead to different constraints or an external circuitry approach.
2.8.1
Simple Zener regulator
A constant output voltage application, normally, can be supplied directly by the auxiliary
winding as illustrated in Figure 38: this structure has to be designed to limit the maximum
VCC voltage during overshooting thanks to the Zener diode DzVcc. The Zener clamping
voltage must be lower than the VCC pin's absolute maximum rating and its current has to be
limited by Rvcc; said resistor represents an additional voltage drop between Vaux and VCC.
Equation 77
Vcc, min  Vaux  Vf, Dvcc  ICC, max  Rvcc
Figure 38. Zener based VCC regulator
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The size of the capacitor CVCC can be estimated considering a trade-off between the
required hold-up time (Thold-up, at least 10 ms to sustain the VCC during the stop mode
occurring at the beginning of an input overvoltage event) and the start-up time.
Equation 78
CVcc 
Thold  up  ICC, max
Vcc, min  Vcc, su max
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63
Designing a high power factor flyback LED driver
AN4710
The resulting start-up time can be estimated as:
Equation 79
 Vcc, su start - up Vcc, on - Vcc, su start - up
Tstart  up  CVcc  

Icharge
 Icharge, su


1
VHVstart 
  2  Arcsin 


2π
Fmains
Vinpk,
min 



The second term in Equation 79 represents the additional delay time that can be present
when the application is turned when Vin is lower than VHV,start.
2.8.2
Linear voltage regulator
There are several situations when the simple Zener regulator is inappropriate. As an
example, when the aux. voltage varies over a wide range of values (LED drivers) the Zener
diode must dissipate any extra voltage exceeding its rating. As a result both very huge
dissipation and regulated voltage increasing could occur when aux. voltage is at maximum
value.
Linear regulator (Figure 39) helps reducing the Zener dissipation.
Figure 39. Linear regulator for VCC supply
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The Vbe rating of QVCC must be higher than maximum aux. voltage, while its package
depends on the associated thermal resistance that allows to keep the QVCC junction
temperature below the desired value.
The diode DVCC2 prevents the base to emitter the junction of QVCC from being reverse
biased by the start-up network (a high voltage start-up unit of the HVLED001).
DzVCC value can be set as the sum of the operating VCC value and twice standard diode's
forward voltage drop (2 x 0.7 V).
54/63
DocID027926 Rev 1
AN4710
Designing a high power factor flyback LED driver
Finally the value for RVCC has to be set in order to bias both - the base current and Zener
diode:
Equation 80
Rvcc 
Vaux, min  VfDvcc  VDzVcc
hfe  ICC, max  IDzVcc
The Zener biasing current (IDzVcc) is usually in the range of few tenths of µA, while hfe is
reported in the NPN transistor datasheet.
The CVCC value is set using Equation 78 and Equation 79. In case the resulting start-up
time is too long a trick can be adopted: selecting a lower CVCC value and increasing the
value of the Caux capacitor to a suitable level. This arrangement ensures contemporarily
a quick start-up (the HVSU charges CVCC only) and a longer hold-up time (the energy is
stored into Caux).
DocID027926 Rev 1
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63
Theory of QR flyback topology fed by an AC line
Appendix A
AN4710
Theory of QR flyback topology fed by an AC
line
It can be demonstrated that the flyback topology operating in the discontinuous conduction
mode is intrinsically able to operate with a high power factor. A particular case of DCM is the
quasi resonant (QR) flyback that, in respect of the standard DCM, optimizes the efficiency
and reduces the high frequency output current ripple. A unity power factor is obtained when
the input current is sinusoidal and in phase with the input voltage.
To obtain almost sinusoidal input current absorption, the HVLED001 device modulates
sinusoidally the peak of the primary side current, but, at the same time, also the duty cycle
(D) is modulated. As a result: the PF and the THD of the input current cannot be
respectively 1 and 0, but their theoretical limit can be calculated. At first, some preliminary
conditions need to be assumed:

The multiplier is perfectly linear so that Ipkp(ɵ) = Ipkp ·sin(ɵ)

The feedback signal (FB voltage) is perfectly constant over a mains cycle

The transformer is ideal so that perfect winding coupling can be assumed

The MOSFET is turned on immediately after demagnetization (Tdemag)
The current absorbed by a flyback converter can be written as:
Equation 81
1
Iint   Ipkpt   D
2
It is convenient to express the currents and voltages in terms of the angular phase to extend
the equations to any input frequency:
The expression for the duty cycle can be obtained writing the expressions for Ton and Toff
periods.
Equation 82
TON 
LPRI  Ipkpθ LPRI  Ipkp

Vinθ
Vinpk
Equation 83
TOFF
LPRI
 N  Ipkpθ
2
L SEC  Ipksθ
L  Ipkpθ LPRI  Ipkpθ

 N
 PRI

Vout  Vf
Vout  Vf
NVout  Vf 
VR
Combining Equation 81 and Equation 84 it is possible to write the expression of the input
current.
Equation 84
D
56/63
TON
1
1


TON  TOFF 1  Vinpk  sinθ 1  K V  sinθ
VR
DocID027926 Rev 1
AN4710
Theory of QR flyback topology fed by an AC line
Equation 85
Iinθ  
sinθ 
1
Ipkp 
2
1  K V sinθ 
Equation 85 contains a distortion term (the denominator of the fractional term) that is equal
to one only if KV equals zero. Practically the ratio between the peak and reflected voltage
cannot be zero.
Considering that the reflected voltage is typically fixed between 150 V and 250 V and the
output voltage could also be variable over a range from 4 to 1, the minimum value of the
parameter KV can be around 0.5 in a wide range applications at minimum line voltage.
The maximum value of this parameter is reached at maximum input voltage and minimum
output voltage: in a wide range applications KV > 7 can be easily reached.
Figure 40 illustrates, in the same graph, the normalized input line voltage (red line) and the
input line currents at different condition. The worst case is a wide range application having
an input voltage range from 90 Vac (KV = 0.5) to 305 Vac. Output voltage can vary by
a factor 4 obtaining, at maximum output voltage a KV varying between around 2 and 7.
The input current amplitudes are normalized to the same RMS values if output condition is
constant, and are adjusted if output voltage decreases.
Figure 40. Estimated input current shape
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,7
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The expression for THD and PF associated with the input current can be explicitly obtained.
Equation 86
PF 
IinRMS,1st_harmonic
IinRMS
DocID027926 Rev 1
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63
Theory of QR flyback topology fed by an AC line
AN4710
Equation 87
IinRMS,1st_harmonic 
Pin
2  Pin

Vin
Vinpk
Equation 88
Pin 
1
sin2 θ 
1
Vinpk  Ipkp
 Vinpk  Ipkp  F2K V 
2
1  K V sinθ  2
Equation 89
1
1
IinRMS   Ipkp 
π
2

sinθ 

 dθ
1  K V sinθ 
2
π
0
Using Equation 87, Equation 88 and Equation 89 into Equation 86 and exploiting numerical
analysis an approximate expression for PF, depending on KV can be found:
Equation 90
PF K V   1  8.1  10 3 K V  3.4  10 4 K 2V
Neglecting the distortion related with input capacitance, the THD can be related to PF using
the following expression:
Equation 91
THDK V  
1
PFK V 2
1
Equation 91 gives a measurement of the distortion already noticed looking at Figure 41.
Figure 41. THD vs. KV graph
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,7
".
58/63
DocID027926 Rev 1
AN4710
List of abbreviations
Appendix B
List of abbreviations
The list of the abbreviations adopted in this application note is reported in this appendix. The
symbols corresponding to HVLED001 electrical characteristics can be found in the relevant
datasheet.
Table 5. List of abbreviations
Symbol
Description
C2
Capacitor of secondary side compensation network belonging to feedback network (CC mode)
Ccl
Primary side clamp capacitor
Cdrain
Cin
Total capacitance of the drain's node
Input capacitor connected after the rectifier bridge
Coss
MOSFET's drain to source capacitance
Cout
Output capacitor
CP
Capacitor of primary side compensation network directly placed between FB and GND
CS
Capacitor of the R/C of the primary side compensation network
Css
Capacitor of the typical R/C network connected to CTRL pin
Csu
Speed up capacitor of the ZCD network
CVCC
D
Dcl
VCC bulk capacitor
Duty cycle of MOSFET
Primary side clamp rectifier
DSec
Secondary side rectifier
DVCC
VCC rectifier diode
DZ_clamp
Primary side clamp Zener
DzVcc
VCC limiting Zener
Fmains
Mains frequency
fResonance
Frequency of the main resonance of the flyback topology
fsw
Switching frequency (avg., min. or max.)
Iin
Input current
ILED
Average value of LED current
Ipkp,max
Peak of primary current - max. value
Ipkp,min
Peak of primary current - min. value
IZCD,on
ZCD pin
KV
Ratio between input peak and reflected voltage
Llk
Transformer's leakage inductance
LPRI
Primary inductance of flyback transformer
LSEC
Secondary side inductance of flyback transformer
N
Primary-to-secondary turn ratio
DocID027926 Rev 1
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63
List of abbreviations
AN4710
Table 5. List of abbreviations (continued)
Symbol
Description
NAUX
Number of turns of transformer's auxiliary (primary side) winding
NPRI
Number of turns of transformer's primary side winding
NSEC
Number of turns of transformer's secondary side winding
PBM
Minimum deliverable power during burst mode
Pin
Input power
Pout
Output power
R1
Resistor of secondary side compensation network connecting E/A with shunt (CC mode)
R2
Resistor of secondary side compensation network belonging to feedback network (CC mode)
Rbias
Optocoupler input biasing resistor
Rcl
Primary side clamp resistor
RCS
Primary side shunt resistor connected in series with the source of the MOSFET
Rds,on
Rd_LED
Rfb
MOSFET's Rds,on
LED dynamical resistance
Lower resistor of the ZCD network
Rhvsu
Resistor in series with high voltage start-up pin
Rmin
Secondary side bleeder resistor
ROUT
Equivalent output capacitor
RP
Resistor of primary side compensation network directly placed between FB and GND
RS
Resistor of the R/C of the primary side compensation network
Rshunt
Secondary side shunt resistor
Rss
Resistor of the typical R/C network connected to CTRL pin
Rzcd
Upper resistor of the ZCD network
Rzcd_bo
Upper resistor of the ZCD network bypassed by diode when low impedance voltage divider is
implemented
Rzcd_su
Upper resistor of the ZCD network bypassed by speed - up capacitor and/or diode
Tamb
Ambient temperature
Tcase
Case temperature
Tdemag
Tj
Demagnetization time of the transformer
Junction case
TON
MOSFET's on time
TOFF
MOSFET's off time
Tvalley
Time interval between MOSFET's turn-off and minimum of the first resonance oscillation
Tvalley_n
Time interval between MOSFET's turn-off and minimum of the n-th resonance oscillation
Vf
VCS
60/63
Secondary side diode forward voltage
Voltage of CS pin
DocID027926 Rev 1
AN4710
List of abbreviations
Table 5. List of abbreviations (continued)
Symbol
Vin
Description
Input voltage
Vinpk
Peak value of input voltage
VLED
LED forward voltage
Vmains
Vout
VR
Mains (AC) voltage applied to the application
Output voltage
Reflected voltage
Vref_CC
Reference voltage for SSR - CC loop
Vref_CV
Reference voltage for SSR - CV loop
Vspike
Primary side overvoltage due to leakage inductance
ILED
Amplitude of the LED current ripple (absolute)
ILED%
Amplitude of the LED current ripple (%)
ILEDpp
Amplitude of the LED current ripple (peak to peak)

Efficiency (%)
ɵ
Angular phase of the mains voltage
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Revision history
AN4710
Revision history
Table 6. Document revision history
62/63
Date
Revision
01-Sep-2015
1
Changes
Initial release.
DocID027926 Rev 1
AN4710
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