TI TPS61040DBV

TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
LOW POWER DC/DC BOOST CONVERTER IN SOT-23 PACKAGE
FEATURES
•
•
•
•
•
•
•
•
DESCRIPTION
1.8 V to 6 V Input Voltage Range
Adjustable Output Voltage Range up to 28 V
400 mA (TPS61040) and 250 mA (TPS61041)
Internal Switch Current
Up to 1 MHz Switching Frequency
28 µA Typical No Load Quiescent Current
1 µA Typical Shutdown Current
Internal Softstart
Available in a Tiny 5-Pin SOT23 Package
The TPS61040/41 is a high-frequency boost converter dedicated for small to medium LCD bias supply
and white LED backlight supplies. The device is ideal
to generate output voltages up to 28 V from a dual
cell NiMH/NiCd or a single cell Li-Ion battery. The
part can also be used to generate standard 3.3 V/5 V
to 12 V power conversions.
The TPS61040/41 operates with a switching frequency up to 1 MHz. This allows the use of small
external components using ceramic as well as tantalum output capacitors. Together with the tiny SOT23
package, the TPS61040/41 gives a very small overall
solution size. The TPS61040 has an internal 400 mA
switch current limit, while the TPS61041 has a 250
mA switch current limit, offering lower output voltage
ripple and allows the use of a smaller form factor
inductor for lower power applications. The low quiescent current (typically 28 µA) together with an
optimized control scheme, allows device operation at
very high efficiencies over the entire load current
range.
APPLICATIONS
•
•
•
•
•
•
•
LCD Bias Supply
White-LED Supply for LCD Backlights
Digital Still Camera
PDAs, Organizers and Handheld PCs
Cellular Phones
Internet Audio Player
Standard 3.3 V/5 V to 12 V Conversion
DBV PACKAGE
(TOP VIEW)
SW
1
GND
2
FB
3
5
VIN
4
EN
TYPICAL APPLICATION
EFFICIENCY
vs
OUTPUT CURRENT
5 V
IN
SW
FB
D1
R1
1
VI = 5 V
86
84
CFF
CO
1 µF
3
VO = 18 V
88
VOUT
VIN to 28 V
Efficiency - %
VIN
1.8 V to 6.0 V
CIN
4.7 µF
90
L1
10 µH
VI = 3.6 V
82
80
VI = 2.4 V
78
76
4
EN
GND
2
R2
74
72
70
0.10
1
10
IO - Output Current - mA
100
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2002–2004, Texas Instruments Incorporated
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
TA
SWITCH CURRENT LIMIT
SOT23 PACKAGE
PACKAGE MARKING
400 mA
TPS61040DBV
PHOI
250 mA
TPS61041DBV
PHPI
-40 to 85°C
(1)
(1)
The DBV package is available in tape & reel. Add R suffix (DBVR) to order quantities of 3000 parts.
FUNCTIONAL BLOCK DIAGRAM
SW
Under Voltage
Lockout
Bias Supply
VIN
400 ns Min
Off Time
Error Comparator
FB
-
S
+
RS Latch
Logic
Gate
Driver
Power MOSFET
N-Channel
VREF = 1.233 V
R
Current Limit
EN
6 µs Max
On Time
+
_
RSENSE
Soft
Start
GND
2
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
Terminal Functions
TERMINAL
NAME
NO.
I/O
I
DESCRIPTION
SW
1
Connect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to the drain of the
internal power MOSFET.
GND
2
FB
3
I
This is the feedback pin of the device. Connect this pin to the external voltage divider to program the desired output
voltage.
EN
4
I
This is the enable pin of the device. Pulling this pin to ground forces the device into shutdown mode reducing the
supply current to less than 1 µA. This pin should not be left floating and needs to be terminated.
VIN
5
I
Supply voltage pin
Ground
DETAILED DESCRIPTION
OPERATION
The TPS61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up to
28 V. The device operates in a pulse frequency modulation (PFM) scheme with constant peak current control.
This control scheme maintains high efficiency over the entire load current range, and with a switching frequency
up to 1 MHz, the device enables the use of very small external components.
The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference voltage
of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon as the
inductor current reaches the internally set peak current of typically 400 mA (TPS61040) or 250 mA (TPS61041).
Refer to the section peak current control for more information. The second criteria that turns off the switch is the
maximum on-time of 6 µs (typical). This is just to limit the maximum on-time of the converter to cover for extreme
conditions. As the switch is turned off the external Schottky diode is forward biased delivering the current to the
output. The switch remains off for a minimum of 400 ns (typical), or until the feedback voltage drops below the
reference voltage again. Using this PFM peak current control scheme the converter operates in discontinuous
conduction mode (DCM) where the switching frequency depends on the output current, which results in very high
efficiency over the entire load current range. This regulation scheme is inherently stable, allowing a wider
selection range for the inductor and output capacitor.
PEAK CURRENT CONTROL
The internal switch turns on until the inductor current reaches the typical dc current limit (ILIM) of 400 mA
(TPS61040) or 250 mA (TPS61042). Due to the internal propagation delay of typical 100 ns, theactualcurrent
exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated:
I
I
Vin 100 ns
peak(typ)
LIM
L
I
400 mA Vin 100 ns for the TPS61040
peak(typ)
L
Vin
100 ns for the TPS61041
I
250 mA peak(typ)
L
(1)
The higher the input voltage and the lower the inductor value, the greater the peak.
By selecting the TPS61040 or TPS61041, it is possible to tailor the design to the specific application current limit
requirements. A lower current limit supports applications requiring lower output power and allows the use of an
inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output
voltage ripple as well.
SOFTSTART
All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This
can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut
down.
3
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
DETAILED DESCRIPTION (continued)
I LIM
The TPS61040/41 limits this inrush current by increasing the current limit in two steps starting from 4
for 256
I LIM
cycles to
2 for the next 256 cycles, and then full current limit (refer to Figure 14).
ENABLE
Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 µA (typical). Since
there is a conductive path from the input to the output through the inductor and Schottky diode, the output
voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should not be
left floating. Using a small external transistor disconnects the input from the output during shutdown as shown in
Figure 18.
UNDERVOLTAGE LOCKOUT
An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V.When the input
voltage is below the undervoltage threshold the main switch is turned off.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature (unless otherwise noted)
(1)
UNIT
Supply voltages on pin VIN
Voltages on pins EN, FB
(2)
-0.3 V to 7 V
(2)
Switch voltage on pin SW
-0.3 V to VIN + 0.3 V
(2)
30 V
Continuous power dissipation
See Dissipation Rating Table
TJ
Operating junction temperature
-40°C to 150°C
TStg
Storage temperature
-65°C to 150°C
Lead temperature (soldering 10 seconds)
(1)
(2)
260°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltage values are with respect to network ground terminal.
DISSIPATION RATING TABLE (1)
PACKAGE
TA ≤ 25°C
POWER RATING
DERATING FACTOR
ABOVE TA = 25°C
TA = 70°C
POWER RATING
TA = 85°C
POWER RATING
DBV
357 mW
3.5 mW/°C
192 mW
140 mW
(1)
The thermal resistance junction to ambient of the 5-pin SOT23 is 250°C/W.
RECOMMENDED OPERATING CONDITIONS
MIN
Vin
Input voltage range
VOUT
Output voltage range
L
Inductor (1)
f
Switching frequency (1)
Cin
Input capacitor
MAX
6
28
2.2
10
UNIT
V
V
µH
1
(1)
(1)
TYP
1.8
4.7
MHz
µF
COUT
Output capacitor
TA
Operating ambient temperature
-40
85
°C
TJ
Operating junction temperature
-40
125
°C
(1)
4
Refer to application section for further information
1
µF
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
ELECTRICAL CHARACTERISTICS
Vin = 2.4 V, EN = VIN, TA = -40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
6
V
50
µA
SUPPLY CURRENT
Vin
Input voltage range
IQ
Operating quiescent current
IOUT= 0 mA, not switching, VFB = 1.3 V
1.8
28
ISD
Shutdown current
EN=GND
0.1
1
µA
VUVLO
Under-voltage lockout threshold
1.5
1.7
V
ENABLE
VIH
EN high level input voltage
VIL
EN low level input voltage
II
EN input leakage current
1.3
EN = GND or VIN
V
0.1
0.4
V
1
µA
30
V
POWER SWITCH AND CURRENT LIMIT
Vsw
Maximum switch voltage
toff
Minimum off time
250
400
550
ns
ton
Maximum on time
4
6
7.5
µs
RDS(ON)
MOSFET on-resistance
Vin = 2.4 V; Isw = 200 mA; TPS61040
600
1000
mΩ
RDS(ON)
MOSFET on-resistance
Vin = 2.4 V; Isw = 200 mA; TPS61041
750
1250
mΩ
MOSFET leakage current
Vsw = 28 V
1
10
µA
ILIM
MOSFET current limit
TPS61040
350
400
450
mA
ILIM
MOSFET current limit
TPS61041
215
250
285
mA
28
V
OUTPUT
VOUT
Adjustable output voltage range
Vref
Internal voltage reference
IFB
Feedback input bias current
VFB= 1.3 V
VFB
Feedback trip point voltage
1.8 V ≤ Vin ≤ 6.0 V
Line regulation
(1)
Load regulation (1)
(1)
Vin
1.233
1.208
1.233
V
1
µA
1.258
V
1.8 V≤ Vin ≤ 6.0 V; Vout = 18 V; Iload = 10 mA
Cff = not connected
0.05
%/V
Vin = 2.4 V; Vout = 18 V; 0 mA≤ Iout ≤ 30 mA
0.15
%/mA
The line and load regulation depend on the external component selection. Refer to the application section for further information.
5
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Load current
1, 2, 3
vs Input voltage
4
Quiescent current
vs Input voltage and temperature
5
VFB
Feedback voltage
vs Temperature
6
ISW
Switch current limit
vs Temperature
7
vs Supply voltage, TPS61041
8
vs Supply voltage, TPS61040
9
vs Temperature
10
vs Supply voltage
11
η
Efficiency
IQ
ICL
Switch current limit
RDSon
RDSon
Line transient response
12
Load transient response
13
Start-up behavior
14
EFFICIENCY
vs
OUTPUT CURRENT
EFFICIENCY
vs
LOAD CURRENT
90
90
VO = 18 V
88
88
VI = 5 V
86
82
80
VI = 2.4 V
78
78
74
74
72
72
1
10
IO - Output Current - mA
100
TPS61041
80
76
Figure 1.
6
82
76
70
0.10
TPS61040
84
VI = 3.6 V
Efficiency - %
Efficiency - %
84
86
L = 10 µH
VO = 18 V
70
0.10
1
10
IL - Load Current - mA
Figure 2.
100
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
EFFICIENCY
vs
LOAD CURRENT
EFFICIENCY
vs
INPUT VOLTAGE
90
88
90
VO = 18 V
86
IO = 10 mA
86
L = 10 µH
84
IO = 5 mA
84
L = 3.3 µH
82
Efficiency - %
Efficiency - %
L = 10 µH
VO = 18 V
88
80
78
82
80
78
76
76
74
74
72
72
70
70
0.10
1
10
IL - Load Current - mA
1
100
2
3
4
5
6
VI - Input Voltage - V
Figure 3.
Figure 4.
TPS61040
QUIESCENT CURRENT
vs
INPUT VOLTAGE
FEEDBACK VOLTAGE
vs
FREE-AIR TEMPERATIRE
1.24
40
TA = 85°C
35
VFB - Feedback Voltage - V
Quiescent Current - µ A
1.238
30
TA = 27°C
25
TA = -40°C
20
15
10
1.236
VCC = 2.4 V
1.234
1.232
5
0
1.8
2.4
3
3.6
4.2
4.8
VI - Input Voltage - V
Figure 5.
5.4
6
1.23
-40
-20
0
20
40
60
80
TA - Temperature - °C
100
120
Figure 6.
7
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
TPS61040/41
SWITCH CURRENT LIMIT
vs
FREE-AIR TEMPERATURE
TPS61041
CURRENT LIMIT
vs
SUPPLY VOLTAGE
260
430
TPS61040
258
390
I (CL) - Current Limit - mA
256
370
350
330
310
290
250
248
246
250
242
230
-40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90
TA - Temperature - °C
240
3
3.6
4.2
4.8
5.4
TPS61040/41
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
FREE-AIR TEMPERATURE
405
400
395
390
385
3
3.6
4.2
4.8
VCC - Supply Voltage - V
Figure 9.
5.4
6
6
VCC - Supply Voltage - V
TPS61040
CURRENT LIMIT
vs
SUPPLY VOLTAGE
TA = 27°C
2.4
2.4
Figure 8.
410
380
1.8
1.8
Figure 7.
415
I (CL) - Current Limit - mA
TA = 27°C
252
TPS61041
420
8
254
244
270
r
− Static Drain-Source On-State Resistance − mΩ
DS(on)
I (SW) - Switch Current Limit - mA
410
1200
1000
TPS61041
800
600
TPS61040
400
200
0
−40 −30 −20 −10 0 10 20 30 40 50 60 70 80 90
TA − Temperature − °C
Figure 10.
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
rDS(on) − Static Drain-Source On-State Resistance − mΩ
TPS61040/41
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
SUPPLY VOLTAGE
1000
VO = 18 V
900
VI
2.4 V to 3.4 V
800
TPS61041
700
600
TPS61040
500
400
VO
100 mV/div
300
200
100
0
1.8
2.4
3
3.6
4.2
4.8
5.4
200 µS/div
6
VCC − Supply Voltage − V
Figure 11.
Figure 12. Line Transient Response
VO = 18 V
VO = 18 V
VO
100 mA/div
VO
5 V/div
EN
1 V/div
VO
1 mA to 10 mA
II
50 mA/div
200 µS/div
Figure 13. Load Transient Tresponse
Figure 14. Start-Up Behavior
9
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
APPLICATION INFORMATION
INDUCTOR SELECTION, MAXIMUM LOAD CURRENT
Since the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability of
the regulator. The selection of the inductor together with the nominal load current, input and output voltage of the
application determines the switching frequency of the converter. Depending on the application, inductor values
between 2.2 µH up to 47 µH are recommended. The maximum inductor value is determined by the maximum on
time of the switch, typically 6 µs. The peak current limit of 400 mA/250mA (typically) should be reached within
this 6-µs period for proper operation.
The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor
value that ensures the maximum switching frequency at the converter maximum load current is not exceeded.
The maximum switching frequency is calculated by the following formula:
Vin
(Vout–Vin)
min
fS max I L Vout
P
(2)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Vinmin = The highest switching frequency occurs at the minimum input voltage
If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step
is to calculate the switching frequency at the nominal load current using the following formula:
2I
(Vout–Vin Vd)
load
fS I
load
I 2L
P
(3)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Iload = Nominal load current
Vd = Rectifier diode forward voltage (typically 0.3V)
A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.
The inductor value has less effect on the maximum available load current and is only of secondary order. The
best way to calculate the maximum available load current under certain operating conditions is to estimate the
expected converter efficiency at the maximum load current. This number can be taken out of the efficiency
graphs shown in Figures 1, 2, 3, and 4. The maximum load current can then be estimated as follows:
I 2 L fSmax
I
P
load max
2 (Vout Vin)
(4)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
fSmax = Maximum switching frequency as calculated previously
η = Expected converter efficiency. Typically 70% to 85%
10
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
APPLICATION INFORMATION (continued)
The maximum load current of the conveter is the current at the operation point where the coverter starts to enter
the continuous conduction mode. Usually the converter should always operate in discontinuous conduction
mode.
Last, the selected inductor should have a saturation current that meets the maximum peak current of the
converter (as calculated in the peak current control section). Use the maximum value for ILIMfor this calculation.
Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency
of the converter. Refer to the Table 1 and the typical applications for the inductor selection.
Table 1. Recommended Inductor for Typical LCD Bias Supply (see Figure 15)
DEVICE
TPS61040
TPS61041
INDUCTOR VALUE
COMPONENT SUPPLIER
COMMENTS
10 µH
Sumida CR32-100
High efficiency
10 µH
Sumida CDRH3D16-100
High efficiency
10 µH
Murata LQH4C100K04
High efficiency
4.7 µH
Sumida CDRH3D16-4R7
Small solution size
4.7 µH
Murata LQH3C4R7M24
Small solution size
10 µH
Murata LQH3C100K24
High efficiency
Small solution size
SETTING THE OUTPUT VOLTAGE
The output voltage is calculated as:
V out 1.233 V 1 R1
R2
(5)
For battery powered applications a high impedance voltage divider should be used with a typical value for R2 of
≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of
the feedback pin.
A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the
error comparator. Without a feedforward capacitor, or one whose value is too small, the TPS61040/41 shows
double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage
ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.
The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good
starting point is to use a 10 pF feedforward capacitor. As a first estimation, the required value for the feedforward
capacitor at the operation point can also be calculated using the following formula:
1
C
FF
2 fS R1
20
(6)
Where:
R1 = Upper resistor of voltage divider
fS = Switching frequency of the converter at the nominal load current (See previous section for calculating
the switching frequency)
CFF = Choose a value that comes closest to the result of the calculation
11
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for line
regulation is paramount, the selected feedforward capacitor should be as small as possible. See the next section
for more information about line and load regulation.
LINE AND LOAD REGULATION
The line regulation of the TPS61040/41 depends on the voltage ripple on the feedback pin. Usually a 50 mV
peak-to-peak voltage ripple on the feedback pin FB gives good results.
Some applications require a very tight line regulation and can only allow a small change in output voltage over a
certain input voltage range. If no feedforward capacitor CFF is used across the upper resistor of the voltage
feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage
ripple is higher because the TPS61040/41 shows output voltage bursts instead of single pulses on the switch pin
(SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage
ripple.
If a larger output capacitor value is not an option, a feedforward capacitor CFF can be used as described in the
previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the
feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation.
There are two ways to improve the line regulation further:
1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple,
as well as the voltage ripple on the feedback pin.
2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback pin
down to 50 mV again. As a starting point, the same capacitor value as selected for the feedforward capacitor
CFF can be used.
OUTPUT CAPACITOR SELECTION
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value but tantalum capacitors can be used as well, depending on the application.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output
voltage ripple can be calculated as:
I
V out out Cout
I L
1
P
–
fS(Iout) Vout Vd–Vin
I ESR
P
(7)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Iout = Nominal load current
fS (Iout) = Switching frequency at the nominal load current as calculated previously
Vd = Rectifier diode forward voltage (typically 0.3V)
Cout = Selected output capacitor
ESR = Output capacitor ESR value
Refer to Table 2 and typical applications section for choosing the output capacitor.
Table 2. Recommended Input and Output Capacitors
DEVICE
TPS61040/41
12
CAPACITOR
VOLTAGE RATING
COMPONENT SUPPLIER
COMMENTS
4.7 µF/X5R/0805
6.3 V
Tayo Yuden JMK212BY475MG
CIN/COUT
10 µF/X5R/0805
6.3 V
Tayo Yuden JMK212BJ106MG
CIN/COUT
1.0 µF/X7R/1206
25 V
Tayo Yuden TMK316BJ105KL
COUT
1.0 µF/X5R/1206
35 V
Tayo Yuden GMK316BJ105KL
COUT
4.7 µF/X5R/1210
25 V
Tayo Yuden TMK325BJ475MG
COUT
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
INPUT CAPACITOR SELECTION
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 µF ceramic input capacitor
is sufficient for most of the applications. For better input voltage filtering this value can be increased. Refer to
Table 2 and typical applications for input capacitor recommendations.
DIODE SELECTION
To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the
peak current rating of the converter as it is calculated in the section peak current control. Use the maximum
value for ILIM for this calculation. Refer to Table 3 and the typical applications for the selection of the Schottky
diode.
Table 3. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 15)
DEVICE
REVERSE VOLTAGE
COMPONENT SUPPLIER
30 V
ON Semiconductor MBR0530
TPS61040/41
20 V
ON Semiconductor MBR0520
20 V
ON Semiconductor MBRM120L
30 V
Toshiba CRS02
COMMENTS
High efficiency
LAYOUT CONSIDERATIONS
Typical for all switching power supplies, the layout is an important step in the design; especially at high peak
currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems
and duty cycle jitter.
The input capacitor should be placed as close as possible to the input pin for good input voltage viltering. The
inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into
other circuits. Since the feedback pin and network is a high impedance circit the feedback network should be
routed away from the inductor. The feedback pin and feedback network should be shielded with a ground plane
or trace to minimize noise coupling into this circuit.
Wide traces should be used for connections in bold as shown in Figure 15. A star ground connection or ground
plane minimizes ground shifts and noise.
D1
L1
VO
VIN
VIN
SW
CFF
R1
CO
FB
CIN
EN
GND
R2
Figure 15. Layout Diagram
13
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
L1
10 µH
VIN
1.8 V to 6 V
D1
VOUT
18 V
TPS61040
VIN
CFF
22 pF
R1
2.2 M
SW
C2
1 µF
FB
C1
4.7 µF
EN
GND
L1:
D1:
C1:
C2:
R2
160 k
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 16. LCD Bias Supply
L1
10 µH
D1
VO
18 V
TPS61040
VIN
1.8 V to 6 V
VIN
CFF
22 pF
R1
2.2 M
SW
C2
1 µF
FB
C1
4.7 µF
EN
GND
DAC or Analog Voltage
0 V = 25 V
1.233 V = 18 V
R2
160 k
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden GMK316BJ105KL
Figure 17. LCD Bias Supply With Adjustable Output Voltage
R3
200 k
VIN
1.8 V to 6 V
L1
10 µH
TPS61040
VIN
C1
4.7 µF
SW
FB
EN
GND
BC857C
D1
VOUT
18 V / 10 mA
R1
2.2 M
C2
1 µF
R2
160 k
CFF
22 pF
C3
0.1 µF
(Optional)
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 18. LCD Bias Supply With Load Disconnect
14
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
D3
V2 = -10 V/15 mA
D2
L1
6.8 µH
C4
4.7 µF
C3
1 µF
D1
V1 = 10 V/15 mA
TPS61040
VIN
VIN = 2.7 V to 5 V
SW
R1
1.5 M
CFF
22 pF
C2
1 µF
FB
C1
4.7 µF
EN
GND
L1:
D1, D2, D3:
C1:
C2, C3, C4:
R2
210 k
Murata LQH4C6R8M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden EMK316BJ105KF
Figure 19. Positive and Negative Output LCD Bias Supply
L1
6.8 µH
D1
VO = 12 V/35 mA
TPS61040
VIN 3.3 V
C1
10 µF
VIN
SW
R1
1.8 M
CFF
4.7 pF
C2
4.7 µF
FB
EN
GND
L1:
D1:
C1:
C2:
R2
205 k
Murata LQH4C6R8M04
Motorola MBR0530
Tayo Yuden JMK212BJ106MG
Tayo Yuden EMK316BJ475ML
Figure 20. Standard 3.3-V to 12-V Supply
D1
3.3 µH
5 V/45 mA
TPS61040
1.8 V to 4 V
VIN
SW
R1
620 k
FB
C1
4.7 µF
EN
GND
R2
200 k
CFF
3.3 pF
C2
4.7 µF
L1:
D1:
C1, C2:
Murata LQH4C3R3M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Figure 21. Dual Battery Cell to 5V/50mA Conversion
Efficiency Aprox. Equals 84% at Vin = 2.4 V to Vo = 5 V/45 mA
15
TPS61040
TPS61041
www.ti.com
SLVS413B – OCTOBER 2002 – REVISED MAY 2004
L1
10 µH
VCC = 2.7 V to 6 V
VIN
D1
SW
C1
4.7 µF
D2
24 V
(Optional)
FB
EN
PWM
100 Hz to 500 Hz
C2
1 µF
GND
RS
82 Ω
L1:
D1:
C1:
C2:
Murata LQH4C100K04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 22. White LED Supply With Adjustable Brightness Control
Using a PWM Signal on the Enable Pin Efficiency Aprox. Equals 86% at Vin = 3 V, ILED = 15 mA
L1
10 µH
VCC = 2.7 V to 6 V
C1
4.7 µF
VIN
SW
D2
24 V
(Optional)
C2†
100 nF
FB
EN
R1
120 k
GND
Analog Brightness Control
3.3 V≅ Led Off
0 V≅ Iled = 20 mA
A.
D1
MBRM120L
R2 160 k
RS
110 L1:
D1:
C1:
C2:
Murata LQH4C3R3M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Standard Ceramic Capacitor
A smaller output capacitor value for C2 will cause a larger LED ripple
Figure 23. White LED Supply With Adjustable Brightness Control
Using an Analog Signal on the Feedback Pin
16
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