ETC CS5301/D

CS5301
Three-Phase
Buck Controller with
Integrated Gate Drivers and
Power Good
The CS5301 is a three–phase step down controller which
incorporates all control functions required to power high performance
processors and high current power supplies. Proprietary multi–phase
architecture guarantees balanced load current distribution and reduces
overall solution cost in high current applications. Enhanced V2
control architecture provides the fastest possible transient response,
excellent overall regulation, and ease of use.
The CS5301 multi–phase architecture reduces output voltage and
input current ripple, allowing for a significant reduction in inductor
values and a corresponding increase in inductor current slew rate. This
approach allows a considerable reduction in input and output capacitor
requirements, as well as reducing overall solution size and cost.
Features
• Enhanced V2 Control Method
• 5–Bit DAC with 1% Accuracy
• Adjustable Output Voltage Positioning
• 6 On–Board Gate Drivers
• 200 kHz to 800 kHz Operation Set by Resistor
• Current Sensed through Buck Inductors, Sense Resistors, or V–S
Control
• Hiccup Mode Current Limit
• Individual Current Limits for Each Phase
• On–Board Current Sense Amplifiers
• 3.3 V, 1.0 mA Reference Output
• 5.0 V and/or 12 V Operation
• On/Off Control (through COMP Pin)
• Power Good Output
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SO–32L
DW SUFFIX
CASE 751P
32
1
MARKING DIAGRAM
32
CS5301
AWLYYWW
1
A
WL, L
YY, Y
WW, W
= Assembly Location
= Wafer Lot
= Year
= Work Week
PIN CONNECTIONS
1
32
ROSC
VCCL
VCCL1
GATE(L)1
GND1
Gate(H)1
VCCH12
Gate(H)2
GND2
Gate(L)2
VCCL23
Gate(L)3
GND3
Gate(H)3
VCCH3
LGND
COMP
VFB
VDRP
CS1
CS2
CS3
CSREF
PWRGD
VID0
VID1
VID2
VID3
VID4
PWRGDS
ILIM
REF
ORDERING INFORMATION
Device
 Semiconductor Components Industries, LLC, 2001
May, 2001 – Rev. 9
1
Package
Shipping
CS5301GDW32
SO–32L
22 Units/Rail
CS5301GDWR32
SO–32L
1000 Tape & Reel
Publication Order Number:
CS5301/D
CS5301
+12 V
+5.0 V
+12 V
D1
BAT54S
D3
BAT54S
L4
300 nH
D2
BAS16LT1
C12
1.0 nF
C1
1.0 µF
R10
10 K
C3
1.0 µF
C2
1.0 µF
+5.0 V
C11
1.0 nF
C4
R3
0.1 µF
56.2 K
C14
1.0 µF
U1
R8
80.6 k
COMP
VFB
VDRP
CS1
CS2
CS3
CSREF
PWRGD
VID0
VID1
VID2
VID3
VID4
PWRGDS
ILIM
REF
CS5301
R11
10 k
+5.0 V
PWRGD
R12
20 K
VID0
VID1
VID2
Q1
CS1
L1
R4
1.0 Ω
C10
470 pF
R9
19.1 k
+ C15,16
2 × 16SP180M
ROSC
VCCL
VCCL1
Gate(L)1
GND1
Gate(H)1
VCCH12
Gate(H)2
GND2
Gate(L)2
VCCL23
Gate(L)3
GND3
Gate(H)3
VCCH3
LGND
1.0 µF
Q2
C17–24
8 × 4SP560M
Q3
CS2
L2
1.0 µF
Q4
C25–32
8 × 10 µF
R1
VID3
VID4
Q5
C5
0.1 µF
3.09 k
R2
L3
R7
R6
30 k
C6
.01 µF
CS1
VCORE
1.0 µF
1.40 k
R5
30 k
CS3
Q6
30 k
C7
.01 µF
Q1–Q6, NTD4302
CS3
C8
.01 µF
CS2
C9
.01 µF
Figure 1. Application Diagram for Intel Pentium 4 Processor
12 V to 1.7 V, 42 A
ABSOLUTE MAXIMUM RATINGS*
Rating
Value
Unit
150
°C
–65 to +150
°C
ESD Susceptibility (Human Body Model)
2.0
kV
Thermal Resistance, Junction–to–Case, RΘJC
14
°C/W
Thermal Resistance, Junction–to–Ambient, RΘJA
70
°C/W
230 peak
°C
Operating Junction Temperature
Storage Temperature Range
Lead Temperature Soldering:
Reflow: (SMD styles only) (Note 1.)
1. 60 second maximum above 183°C.
*The maximum package power dissipation must be observed.
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2
CS5301
ABSOLUTE MAXIMUM RATINGS
Pin Name
Pin Symbol
VMAX
VMIN
ISOURCE
ISINK
Power for Logic
VCCL
16 V
–0.3 V
N/A
70 mA DC
Power Good Sense
PWRGDS
6.0 V
–0.3 V
1.0 mA
1.0 mA
Power Good Output
PWRGD
6.0 V
–0.3 V
1.0 mA
20 mA
Return for Logic
LGND
N/A
N/A
2.0 A, 1.0 µs 200 mA DC
N/A
Power for Gate(L)1
VCCL1
16 V
–0.3 V
N/A
1.5 A, 1.0 µs 200 mA DC
Power for Gate(L)2 and Gate(L)3
VCCL23
16 V
–0.3 V
N/A
1.5 A, 1.0 µs 200 mA DC
Power for Gate(H)1 and Gate(H)2
VCCH12
20 V
–0.3 V
N/A
1.5 A, 1.0 µs 200 mA DC
Power for Gate(H)3
VCCH3
20 V
–0.3 V
N/A
1.5 A, 1.0 µs 200 mA DC
Voltage Feedback Compensation
Network
COMP
6.0 V
–0.3 V
1.0 mA
1.0 mA
Voltage Feedback Input
VFB
6.0 V
–0.3 V
1.0 mA
1.0 mA
Output for Adjusting Adaptive
Voltage Positioning
VDRP
6.0 V
–0.3 V
1.0 mA
1.0 mA
Frequency Resistor
ROSC
6.0 V
–0.3 V
1.0 mA
1.0 mA
Reference Output
REF
6.0 V
–0.3 V
1.0 mA
50 mA
High Side FET Drivers
GH1–3
20 V
–0.3 V DC
–2.0 V for
100 ns
1.5 A, 1.0 µs 200 mA DC
1.5 A, 1.0 µs 200 mA DC
Low Side FET Drivers
GL1–3
16 V
–0.3 V DC
–2.0 V for
100 ns
1.5 A, 1.0 µs 200 mA DC
1.5 A, 1.0 µs 200 mA DC
Return for #1 Driver
GND1
0.3 V
–0.3 V
2.0 A, 1.0 µs 200 mA DC
N/A
Return for #2 Driver
GND2
0.3 V
–0.3 V
2.0 A, 1.0 µs 200 mA DC
N/A
Return for #3 Driver
GND3
0.3 V
–0.3 V
2.0 A, 1.0 µs 200 mA DC
N/A
Current Sense for Phases 1–3
CS1–CS3
6.0 V
–0.3 V
1.0 mA
1.0 mA
Current Limit Set Point
ILIM
6.0 V
–0.3 V
1.0 mA
1.0 mA
Output Voltage
CSREF
6.0 V
–0.3 V
1.0 mA
1.0 mA
Voltage ID DAC Inputs
VID0–4
6.0 V
–0.3 V
1.0 mA
1.0 mA
ELECTRICAL CHARACTERISTICS (0°C < TA < 70°C; 0°C < TJ < 125°C; 4.7 V < VCCL < 14 V; 8.0 V < VCCH < 20 V;
CGATE(H) = 3.3 nF, CGATE(L) = 3.3 nF, RROSC = 53.6 k, CCOMP = 0.1 µF, CREF = 0.1 µF, DAC Code 10000, CVCC = 1.0 µF, ILIM ≥ 1.0 V;
unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
± 1.0
%
Voltage Identification DAC (0 = Connected to GND; 1 = Open or Pull–up to internal 3.3 V or external 5.0 V)
Accuracy (all codes)
VID code – 125 mV
Connect VFB to COMP,
Measure COMP
–
–
VID4
VID3
VID2
VID1
VID0
VID Voltage
DACOUT Voltage
–
1
1
1
1
1
–
FAULT Mode–Output Off
–
1
1
1
1
0
1.100
0.965
0.975
0.985
V
1
1
1
0
1
1.125
0.990
1.000
1.010
V
1
1
1
0
0
1.150
1.015
1.025
1.035
V
1
1
0
1
1
1.175
1.040
1.050
1.061
V
1
1
0
1
0
1.200
1.064
1.075
1.086
V
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CS5301
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; 4.7 V < VCCL < 14 V; 8.0 V < VCCH < 20 V;
CGATE(H) = 3.3 nF, CGATE(L) = 3.3 nF, RROSC = 53.6 k, CCOMP = 0.1 µF, CREF = 0.1 µF, DAC Code 10000, CVCC = 1.0 µF, ILIM ≥ 1.0 V;
unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Voltage Identification DAC (0 = Connected to GND; 1 = Open or Pull–up to internal 3.3 V or external 5.0 V)
1
1
0
0
1
1.225
1.089
1.100
1.111
V
1
1
0
0
0
1.250
1.114
1.125
1.136
V
1
0
1
1
1
1.275
1.139
1.150
1.162
V
1
0
1
1
0
1.300
1.163
1.175
1.187
V
1
0
1
0
1
1.325
1.188
1.200
1.212
V
1
0
1
0
0
1.350
1.213
1.225
1.237
V
1
0
0
1
1
1.375
1.238
1.250
1.263
V
1
0
0
1
0
1.400
1.263
1.275
1.288
V
1
0
0
0
1
1.425
1.287
1.300
1.313
V
1
0
0
0
0
1.450
1.312
1.325
1.338
V
0
1
1
1
1
1.475
1.337
1.350
1.364
V
0
1
1
1
0
1.500
1.361
1.375
1.389
V
0
1
1
0
1
1.525
1.386
1.400
1.414
V
0
1
1
0
0
1.550
1.411
1.425
1.439
V
0
1
0
1
1
1.575
1.436
1.450
1.465
V
0
1
0
1
0
1.600
1.460
1.475
1.490
V
0
1
0
0
1
1.625
1.485
1.500
1.515
V
0
1
0
0
0
1.650
1.510
1.525
1.540
V
0
0
1
1
1
1.675
1.535
1.550
1.566
V
0
0
1
1
0
1.700
1.560
1.575
1.591
V
0
0
1
0
1
1.725
1.584
1.600
1.616
V
0
0
1
0
0
1.750
1.609
1.625
1.641
V
0
0
0
1
1
1.775
1.634
1.650
1.667
V
0
0
0
1
0
1.800
1.658
1.675
1.692
V
0
0
0
0
1
1.825
1.683
1.700
1.717
V
0
0
0
0
0
1.850
1.708
1.725
1.742
V
Input Threshold
VID4, VID3, VID2, VID1, VID0
1.00
1.25
1.50
V
Input Pull–up Resistance
VID4, VID3, VID2, VID1, VID0
25
50
100
kΩ
3.15
3.30
3.45
V
Pull–up Voltage
–
Power Good Output
Upper Threshold
Force PWRGDS
1.9 (–5%)
2.0
2.1 (+5%)
V
Lower Threshold
Force PWRGDS
0.95 × (VID –
125 mV)
or –2.6%
from nominal
PWRGD
Threshold
0.975 × (VID –
125 mV)
VID –
125 mV
or +2.6%
from nominal
PWRGD
Threshold
V
0.926
0.951
0.975
V
VID4
VID3
VID2
VID1
VID0
1
1
1
1
0
–
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CS5301
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; 4.7 V < VCCL < 14 V; 8.0 V < VCCH < 20 V;
CGATE(H) = 3.3 nF, CGATE(L) = 3.3 nF, RROSC = 53.6 k, CCOMP = 0.1 µF, CREF = 0.1 µF, DAC Code 10000, CVCC = 1.0 µF, ILIM ≥ 1.0 V;
unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Power Good Output
1
1
1
0
1
–
0.950
0.975
1.000
V
1
1
1
0
0
–
0.974
1.000
1.025
V
1
1
0
1
1
–
0.998
1.024
1.050
V
1
1
0
1
0
–
1.021
1.048
1.075
V
1
1
0
0
1
–
1.045
1.073
1.100
V
1
1
0
0
0
–
1.069
1.097
1.125
V
1
0
1
1
1
–
1.093
1.122
1.150
V
1
0
1
1
0
–
1.116
1.146
1.175
V
1
0
1
0
1
–
1.140
1.170
1.200
V
1
0
1
0
0
–
1.164
1.195
1.225
V
1
0
0
1
1
–
1.188
1.219
1.250
V
1
0
0
1
0
–
1.211
1.243
1.275
V
1
0
0
0
1
–
1.235
1.268
1.300
V
1
0
0
0
0
–
1.259
1.292
1.325
V
0
1
1
1
1
–
1.283
1.316
1.350
V
0
1
1
1
0
–
1.306
1.341
1.375
V
0
1
1
0
1
–
1.330
1.365
1.400
V
0
1
1
0
0
–
1.354
1.389
1.425
V
0
1
0
1
1
–
1.378
1.414
1.450
V
0
1
0
1
0
–
1.401
1.438
1.475
V
0
1
0
0
1
–
1.425
1.463
1.500
V
0
1
0
0
0
–
1.449
1.487
1.525
V
0
0
1
1
1
–
1.473
1.511
1.550
V
0
0
1
1
0
–
1.496
1.536
1.575
V
0
0
1
0
1
–
1.520
1.560
1.600
V
0
0
1
0
0
–
1.544
1.584
1.625
V
0
0
0
1
1
–
1.568
1.609
1.650
V
0
0
0
1
0
–
1.591
1.633
1.675
V
0
0
0
0
1
–
1.615
1.658
1.700
V
0
0
0
0
0
–
1.639
1.682
1.725
V
Switch Leakage Current
PWRGD = 5.5 V
PWRGDS = 1.60 V
–
0.1
10.0
µA
Delay
PWRGDS low to PWRGD low
25
50
125
µs
Output Low Voltage
PWRGDS = 1.0 V
IPWRGD = 4.0 mA
–
0.15
0.40
V
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CS5301
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; 4.7 V < VCCL < 14 V; 8.0 V < VCCH < 20 V;
CGATE(H) = 3.3 nF, CGATE(L) = 3.3 nF, RROSC = 53.6 k, CCOMP = 0.1 µF, CREF = 0.1 µF, DAC Code 10000, CVCC = 1.0 µF, ILIM ≥ 1.0 V;
unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Voltage Feedback Error Amplifier
VFB Bias Current, (Note 1.)
0.9 V < VFB < 1.8 V
5.5
6.0
6.5
µA
COMP Source Current
COMP = 0.5 V to 2.0 V;
VFB = 1.6 V; DAC = 00000
15
30
60
µA
COMP Sink Current
COMP = 0.5 V to 2.0 V;
VFB = 1.8 V; DAC = 00000
15
30
60
µA
0.20
0.27
0.34
V
–
32
–
mmho
–
2.5
–
MΩ
COMP Discharge Threshold Voltage
Transconductance
–
–10 µA < ICOMP < +10 µA
Output Impedance
–
Open Loop DC Gain
Note 2.
60
90
–
dB
Unity Gain Bandwidth
0.01 µF COMP Capacitor
–
400
–
kHz
–
70
–
dB
PSRR @ 1.0 kHz
–
COMP Max Voltage
VFB = 1.65 V COMP Open;
DAC = 00000
2.4
2.7
–
V
COMP Min Voltage
VFB = 1.8 V COMP Open;
DAC = 00000
–
0.1
0.2
V
–
2
5.0
10
µA
Minimum Pulse Width
Measured from CSx to GATE(H),
V(VFB) = V(CSREF) = 0 V,
V(COMP) = 0.5 V, 60 mV step
applied between VCSX and VCREF
–
350
515
ns
Channel Startup Offset
V(CS1) = V(CS2) = V(CS3) =
V(VFB) = V(CSREF) = 0 V;
Measure V(COMP) when
GATE1(H), 2(H), 3(H) switch high
0.3
0.4
0.5
V
High Voltage (AC)
Note 2. Measure VCCLX – GATE(L)
or VCCHX – GATE(H)
–
0
1.0
V
Low Voltage (AC)
Note 2. Measure GATE(L)
or GATE(H)
–
0
0.5
V
Rise Time GATE(H)x
1.0 V < GATE < 8.0 V; VCCHX = 10 V
–
35
80
ns
Rise Time GATE(L)x
1.0 V < GATE < 8.0 V; VCCLX = 10 V
–
35
80
ns
Fall Time GATE(H)x
8.0 V > GATE > 1.0 V; VCCHX = 10 V
–
35
80
ns
Fall Time GATE(L)x
8.0 V > GATE > 1.0 V; VCCLX = 10 V
–
35
80
ns
GATE(H) to GATE(L) Delay
GATE(H) < 2.0 V, GATE(L) > 2.0 V
30
65
110
ns
GATE(L) to GATE(H) Delay
GATE(L) < 2.0 V, GATE(H) > 2.0 V
30
65
110
ns
GATE Pull–down
Force 100 µA into GATE Driver
with VCCHX = VCCLX = 2.0 V
–
1.2
1.6
V
Hiccup Latch Discharge Current
PWM Comparators
GATE(H) and GATE(L)
1. The VFB Bias Current changes with the value of RROSC per Figure 4.
2. Guaranteed by design. Not tested in production.
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CS5301
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; 4.7 V < VCCL < 14 V; 8.0 V < VCCH < 20 V;
CGATE(H) = 3.3 nF, CGATE(L) = 3.3 nF, RROSC = 53.6 k, CCOMP = 0.1 µF, CREF = 0.1 µF, DAC Code 10000, CVCC = 1.0 µF, ILIM ≥ 1.0 V;
unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Switching Frequency
Measure any phase (RROSC = 32.4 k)
Note 3.
300
400
500
kHz
Switching Frequency
Measure any phase (RROSC = 53.6 k)
220
250
280
kHz
Switching Frequency
Measure any phase (RROSC = 16.2 k)
Note 3.
600
800
1000
kHz
ROSC Voltage
–
–
1.00
–
V
Phase Delay
–
105
120
135
deg
Oscillator
Adaptive Voltage Positioning
VDRP Output Voltage to DACOUT Offset
CS1 = CS2 = CS3 = CSREF,
VFB = COMP,
Measure VDRP– COMP
–15
–
15
mV
Maximum VDRP Voltage
|(CS1 = CS2 = CS3) = CSREF| =
50 mV, VFB = COMP,
|Measure VDRP– COMP|
360
465
570
mV
–
2.7
3.1
3.5
V/V
–
–
0.6
2.0
µA
–
0.2
2.0
µA
3.7
4.2
4.7
V/V
–5.0
–
5.0
mV
0
–
VCCL – 2.0
V
Current Sense Amp to VDRP Gain
Current Sensing and Sharing
CSREF Input Bias Current
CS1–CS3 Input Bias Current
V(CSx) = V(CSREF) = 0 V
Current Sense Amplifiers Gain
–
Current Sense Amp Mismatch
0 V ≤ (CSx–CSREF) ≤ 50 mV
Current Sense Amplifiers Input
Common Mode Range Limit
Note 3.
Current Sense Input to ILIM Gain
0.25 V < ILIM < 1.20 V
5.0
6.5
8.0
V/V
Current Limit Filter Slew Rate
Note 3.
4.0
10
26
mV/µs
ILIM Bias Current
0 V < ILIM < 1.0 V
–
0.1
1.0
µA
–
75
90
115
mV
Note 3.
1.0
–
–
MHz
0 mA < I(VREF) < 1.0 mA
3.15
3.25
3.35
V
Single Phase Pulse by Pulse Current
Limit: V(CSx) – V(CSREF)
Current Sense Amplifier Bandwidth
Reference Output
VREF Output Voltage
3. Guaranteed by design. Not tested in production.
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CS5301
ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < 70°C; 0°C < TJ < 125°C; 4.7 V < VCCL < 14 V; 8.0 V < VCCH < 20 V;
CGATE(H) = 3.3 nF, CGATE(L) = 3.3 nF, RROSC = 53.6 k, CCOMP = 0.1 µF, CREF = 0.1 µF, DAC Code 10000, CVCC = 1.0 µF, ILIM ≥ 1.0 V;
unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
General Electrical Specifications
VCCL Operating Current
VFB = COMP(no switching)
–
22
26
mA
VCCL1 Operating Current
VFB = COMP(no switching)
–
4.0
5.5
mA
VCCL23 Operating Current
VFB = COMP(no switching)
–
8.0
11
mA
VCCH12 Operating Current
VFB = COMP(no switching)
–
5.5
7.0
mA
VCCH3 Operating Current
VFB = COMP(no switching)
–
2.5
3.5
mA
VCCL Start Threshold
GATEs switching, COMP charging
4.05
4.50
4.70
V
VCCL Stop Threshold
GATEs stop switching, COMP
discharging
3.75
4.30
4.60
V
VCCL Hysteresis
GATEs not switching, COMP not
charging
100
200
300
mV
VCCH12 Start Threshold
–
3.2
3.5
3.8
V
VCCH12 Stop Threshold
–
2.9
3.2
3.5
V
VCCH12 Start Hysteresis
–
200
300
400
mV
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CS5301
PACKAGE PIN DESCRIPTION
PACKAGE PIN #
32 Lead SO Wide
PIN SYMBOL
1
COMP
2
VFB
Voltage Feedback Pin. To use Adaptive Voltage Positioning
(AVP) select an offset voltage at light load and connect a
resistor between VFB and VOUT. The input bias current of the
VFB pin and the resistor value determine output voltage offset
for zero output current. Short VFB to VOUT for no AVP.
3
VDRP
Current sense output for AVP. The offset of this pin above the
DAC voltage is proportional to the output current. Connect a
resistor from this pin to VFB to set amount AVP or leave this
pin open for no AVP.
4–6
CS1–CS3
Current sense inputs. Connect current sense network for the
corresponding phase to each input.
7
CSREF
Reference for current sense amplifiers. To balance input
offset voltages between the inverting and noninverting inputs
of the Current Sense Amplifiers, connect a resistor between
CSREF and the output voltage. The value should be 1/3 of
the value of the resistors connected to the CSx pins.
8
PWRGD
Power Good Output. Open collector output goes low when
CSREF is out of regulation.
9–13
VID4–VID0
Voltage ID DAC inputs. These pins are internally pulled up to
3.3 V if left open.
14
PWRGDS
Power Good Sense. Connect to Output.
15
ILIM
Sets threshold for current limit. Connect to reference through
a resistive divider.
16
REF
Reference output. Decouple with 0.1 µF to LGND.
17
LGND
Return for internal control circuits and IC substrate connection.
18
VCCH3
Power for GATE(H)3.
19
Gate(H)3
20
GND3
21
Gate(L)3
22
VCCL23
23
Gate(L)2
Low side driver #2.
24
GND2
Return for #2 driver.
25
Gate(H)2
High side driver #2.
26
VCCH12
27
Gate(H)1
28
GND1
Return #1 drivers.
29
Gate(L)1
Low side driver #1.
30
VCCL1
Power for GATE(L)1.
31
VCCL
Power for internal control circuits and UVLO Sense for Logic.
32
ROSC
A resistor from this pin to ground sets operating frequency
and VFB bias current.
FUNCTION
Output of the error amplifier and input for the PWM
comparators.
High side driver #3.
Return for #3 drivers.
Low side driver #3.
Power for GATE(L)2 and GATE(L)3.
Power for GATE(H)1 and GATE(H)2. UVLO Sense for High
Side Driver supply connects to this pin.
High side driver #1.
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VCCL
VCCL
VCCH12
VCCH3
VCCH12
ILIM
+
VID1
–
START
STOP
START
STOP
+
+
+
4.5 V
4.3 V
–
DAC
PH1
–
–
S
–
3.5 V
3.2 V
PWMC1
+
VID4
Gate(L)1
GND1
MAXC1
DACOUT
FAULT
VCCL1
+
125 mV
–
Gate(H)1
Gate
Nonoverlap
R
CO1
+
VID3
VCCH12
Reset
Dominant
3.3 V
REF
VID0
VID2
VCCL23
–
+
S
S
0.4 V
–
–
OVIC
PWMC2
+
R
–
Gate(H)2
Gate
Nonoverlap
VCCL23
Gate(L)2
R
CO2
+
VCCH12
FAULT
+
–
CO2
+
–
–
–
–
+
–
CS1
VITotal
1
+
2
CO3
CO1
5.0 µA
+
CO2
+
Current
Source
Gen
0.4 V
1
CO3
FAULT
×4
2
EA
OSC
DACOUT
VDRP
COMP
PH 2
PH 3
DACOUT
LGND
BIAS
PH 1
+
CSREF
CSA3
FAULT
× 0.75
–
–
GND3
0.4 V
+
+
AVPA
CS3
–
–
–
CS2
Gate(L)3
MAXC3
–
–
VCCL23
+
FAULT
+
CSA2
Gate(H)3
Gate
Nonoverlap
R
CO3
+
DAC X
97.5 %
CSA1
0.4 V
–
+
+
PWRGDS
S
PWMC3
0.25 V
VCCH3
FAULT
–
2.0 V +
DLY
–
+
PH3
Reset
Dominant
× 1.5
RESC
PWRGD
GND2
MAXC2
VFB ROSC
CS5301
10
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Figure 2. Block Diagram
Set
Dominant
CO1
PH2
Reset
Dominant
REF
VCCL
VCCL1
CS5301
TYPICAL PERFORMANCE CHARACTERISTICS
900
25
VFB Bias Current, µA
800
Frequency, kHz
700
600
500
400
300
20
15
10
5
200
100
10
20
30
40
50
60
0
10
70
20
30
RROSC Value, kΩ
70
80
Figure 4. VFB Bias Current vs. RROSC Value
120
120
100
100
80
80
Time, ns
Time, ns
60
RROSC Value, kΩ
Figure 3. Oscillator Frequency
60
60
40
40
20
20
0
0
0
2
4
6
8
10
12
14
0
16
2
4
6
8
10
12
14
16
Load Capacitance, nF
Load Capacitance, nF
Figure 5. Gate(H) Rise–time vs. Load Capacitance
measured from 1.0 V to 4.0 V with VCC at 5.0 V.
Figure 6. Gate(H) Fall–time vs. Load Capacitance
measured from 4.0 V to 1.0 V with VCC at 5.0 V.
120
120
100
100
80
80
Time, ns
Time, ns
50
40
60
60
40
40
20
20
0
0
0
2
4
6
8
10
12
14
16
0
2
4
6
8
10
12
14
16
Load Capacitance, nF
Load Capacitance, nF
Figure 7. Gate(L) Rise–time vs. Load Capacitance
measured from 4.0 V to 1.0 V with VCC at 5.0 V.
Figure 8. Gate(L) Fall–time vs. Load Capacitance
measured from 4.0 V to 1.0 V with VCC at 5.0 V.
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CS5301
APPLICATIONS INFORMATION
FIXED FREQUENCY MULTI–PHASE CONTROL
comparator rises and terminates the PWM cycle. If the
inductor starts the cycle with a higher current, the PWM
cycle will terminate earlier providing negative feedback.
The CS5301 provides a CSx input for each phase, but the
CSREF, VFB and COMP inputs are common to all phases.
Current sharing is accomplished by referencing all phases to
the same VFB and COMP pins, so that a phase with a larger
current signal will turn off earlier than phases with a smaller
current signal.
Including both current and voltage information in the
feedback signal allows the open loop output impedance of
the power stage to be controlled. If the COMP pin is held
steady and the inductor current changes, there must also be
a change in the output voltage. Or, in a closed loop
configuration when the output current changes, the COMP
pin must move to keep the same output voltage. The required
change in the output voltage or COMP pin depends on the
scaling of the current feedback signal and is calculated as
In a multi–phase converter, multiple converters are
connected in parallel and are switched on at different times.
This reduces output current from the individual converters
and increases the apparent ripple frequency. Because several
converters are connected in parallel, output current can ramp
up or down faster than a single converter (with the same
value output inductor) and heat is spread among multiple
components.
The CS5301 uses a three–phase, fixed frequency,
Enhanced V2 architecture. Each phase is delayed 120° from
the previous phase. Normally GATE(H) transitions high at
the beginning of each oscillator cycle. Inductor current
ramps up until the combination of the current sense signal
and the output ripple trip the PWM comparator and bring
GATE(H) low. Once GATE(H) goes low, it will remain low
until the beginning of the next oscillator cycle. While
GATE(H) is high, the enhanced V2 loop will respond to line
and load transients. Once GATE(H) is low, the loop will not
respond again until the beginning of the next cycle.
Therefore, constant frequency Enhanced V2 will typically
respond within 1/3 of the off–time for a three–phase
converter.
The Enhanced V2 architecture measures and adjusts
current in each phase. An additional input (CSx) for inductor
current information has been added to the V2 loop for each
phase as shown in Figure 9.
SWNODE
L
RL
CSX
+
CSA
RS
OFFSET
CSREF
The single–phase power stage output impedance is:
Single Stage Impedance VI RS CSA Gain.
The multi–phase power stage output impedance is the
single–phase output impedance divided by the number of
phases. The output impedance of the power stage determines
how the converter will respond during the first few µs of a
transient before the feedback loop has repositioned the
COMP pin.
The peak output current of each phase can also be
calculated from;
V
VFB VOFFSET
Ipkout (per phase) COMP
RS CSA Gain
+
+
+
VOUT
V RS CSA Gain I
Figure 10 shows the step response of a single phase with
the COMP pin at a fixed level. Before T1 the converter is in
normal steady state operation. The inductor current provides
the PWM ramp through the Current Share Amplifier. The
PWM cycle ends when the sum of the current signal, voltage
signal and OFFSET exceed the level of the COMP pin. At
T1 the output current increases and the output voltage sags.
The next PWM cycle begins and the cycle continues longer
than previously while the current signal increases enough to
make up for the lower voltage at the VFB pin and the cycle
ends at T2. After T2 the output voltage remains lower than
at light load and the current signal level is raised so that the
sum of the current and voltage signal is the same as with the
original load. In a closed loop system the COMP pin would
move higher to restore the output voltage to the original
level.
PWM
COMP
+
VFB
+
DACOUT
+
E.A.
+
COMP
Figure 9. Enhanced V2 Feedback and Current
Sense Scheme
The inductor current is measured across RS, amplified by
CSA and summed with the OFFSET and Output Voltage at
the non–inverting input of the PWM comparator. The
inductor current provides the PWM ramp and as inductor
current increases the voltage on the positive pin of the PWM
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CS5301
considered when setting the ILIM threshold. If a more
accurate current sense is required than inductive sensing can
provide, current can be sensed through a resistor as shown
in Figure 9.
SWNODE
Current Sharing Accuracy
VFB (VOUT)
PCB traces that carry inductor current can be used as part
of the current sense resistance depending on where the
current sense signal is picked off. For accurate current
sharing, the current sense inputs should sense the current at
the same point for each phase and the connection to the
CSREF should be made so that no phase is favored. (In some
cases, especially with inductive sensing, resistance of the
pcb can be useful for increasing the current sense
resistance.) The total current sense resistance used for
calculations must include any pcb trace between the CSx
inputs and the CSREF input that carries inductor current.
Current Sense Amplifier Input Mismatch and the value of
the current sense element will determine the accuracy of
current sharing between phases. The worst case Current
Sense Amplifier Input Mismatch is 5.0 mV and will
typically be within 3.0 mV. The difference in peak currents
between phases will be the CSA Input Mismatch divided by
the current sense resistance. If all current sense elements are
of equal resistance a 3.0 mV mismatch with a 2.0 mΩ sense
resistance will produce a 1.5 A difference in current between
phases.
CSA Out
COMP – Offset
CSA Out + VFB
T1
T2
Figure 10. Open Loop Operation
Inductive Current Sensing
For lossless sensing, current can be sensed across the
inductor as shown in Figure 11. In the diagram L is the output
inductance and RL is the inherent inductor resistance. To
compensate the current sense signal the values of R1 and C1
are chosen so that L/RL = R1 × C1. If this criteria is met the
current sense signal will be the same shape as the inductor
current, the voltage signal at CSx will represent the
instantaneous value of inductor current and the circuit can be
analyzed as if a sense resistor of value RL was used as a sense
resistor (RS).
Operation at > 50% Duty Cycle
For operation at duty cycles above 50% Enhanced V2 will
exhibit subharmonic oscillation unless a compensation
ramp is added to each phase. A circuit like the one on the left
side of Figure 12 can be added to each current sense network
to implement slope compensation. The value of R1 can be
varied to adjust the ramp size.
R1
SWNODE
CSX
L
C1
RL
VOUT
+
CSA
OFFSET
CSREF
+
+
+
+
PWM
COMP
DACOUT
Switch Node
Gate(L)X
VFB
E.A.
+
COMP
R1
3.0 k
25 k
CSX
Figure 11. Lossless Inductive Current Sensing with
Enhanced V2
1.0 nF
0.1 µF
When choosing or designing inductors for use with
inductive sensing tolerances and temperature effects should
be considered. Cores with a low permeability material or a
large gap will usually have minimal inductance change with
temperature and load. Copper magnet wire has a
temperature coefficient of 0.39% per °C. The increase in
winding resistance at higher temperatures should be
.01 µF
CSREF
MMBT2222LT1
Slope Comp
Circuit
Existing Current
Sense Circuit
Figure 12. External Slope Compensation Circuit
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CS5301
Ramp Size and Current Sensing
Because the current ramp is used for both the PWM ramp
and to sense current, the inductor and sense resistor values
will be constrained. A small ramp will provide a quick
transient response by minimizing the difference over which
the COMP pin must travel between light and heavy loads,
but a steady state ramp of 25 mVp–p or greater is typically
required to prevent pulse skipping and minimize pulse width
jitter. For resistive current sensing, the combination of the
inductor and sense resistor values must be chosen to provide
a large enough steady state ramp. For large inductor values
the sense resistor value must also be increased.
For inductive current sensing, the RC network must meet
the requirement of L/RL = R × C to accurately sense the AC
and DC components of the current the signal. Again the
values for L and RL will be constrained in order to provide
a large enough steady state ramp with a compensated current
sense signal. A smaller L, or a larger RL than optimum might
be required. But unlike resistive sensing, with inductive
sensing, small adjustments can be made easily with the
values of R and C to increase the ramp size if needed.
If RC is chosen to be smaller (faster) than L/RL, the AC
portion of the current sensing signal will be scaled larger
than the DC portion. This will provide a larger steady state
ramp, but circuit performance will be affected and must be
evaluated carefully. The current signal will overshoot during
transients and settle at the rate determined by R × C. It will
eventually settle to the correct DC level, but the error will
decay with the time constant of R × C. If this error is
excessive it will effect transient response, adaptive
positioning and current limit. During transients the COMP
pin will be required to overshoot along with the current
signal in order to maintain the output voltage. The VDRP pin
will also overshoot during transients and possibly slow the
response. Single phase overcurrent will trip earlier than it
would if compensated correctly and hiccup mode current
limit will have a lower threshold for fast rise step loads than
for slowly rising output currents.
The waveforms in Figure 13 show a simulation of the
current sense signal and the actual inductor current during a
positive step in load current with values of L = 500 nH,
RL = 1.6 mΩ, R1 = 20 k and C1 = .01 µF. For ideal current
signal compensation the value of R1 should be 31 kΩ. Due
to the faster than ideal RC time constant there is an overshoot
of 50% and the overshoot decays with a 200 µs time
constant. With this compensation the ILIM pin threshold
must be set more than 50% above the full load current to
avoid triggering hiccup mode during a large output load
step.
Figure 13. Inductive Sensing waveform during a Step
with Fast RC Time Constant (50 µs/div)
Current Limit
Two levels of overcurrent protection are provided. Any
time the voltage on a Current Sense pin exceeds CSREF by
more than the Single Phase Pulse by Pulse Current Limit, the
PWM comparator for that phase is turned off. This provides
fast peak current protection for individual phases. The
outputs of all the currents are also summed and filtered to
compare an averaged current signal to the voltage on the
ILIM pin. If this voltage is exceeded, the fault latch trips and
the Soft Start capacitor is discharged by a 5.0 µA source
until the COMP pin reaches 0.2 V. Then Soft Start begins.
The converter will continue to operate in this mode until the
fault condition is corrected.
Overvoltage Protection
Overvoltage protection (OVP) is provided as a result of
the normal operation of the Enhanced V2 control topology
with synchronous rectifiers. The control loop responds to an
overvoltage condition within 400 ns, causing the top
MOSFET’s to shut off and the synchronous MOSFET’s to
turn on. This results in a “crowbar” action to clamp the
output voltage and prevent damage to the load. The regulator
will remain in this state until the overvoltage condition
ceases or the input voltage is pulled low.
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CS5301
UVLO
voltage is not repositioned quickly enough after current is
stepped up and the upper limit is exceeded.
The CS5301 has undervoltage lockout functions connected
to two pins. One intended for the logic and low–side drivers
with a 4.5 V turn–on threshold is connected to the VCCL pin.
A second for the high side drivers has a 3.5 V threshold and
is connected to the VCCH12 pin.
The UVLO threshold for the high side drivers was chosen
at a low value to allow for flexibility in the part. In many
applications this function will be disabled or will only check
that the applicable supply is on – not that is at a high enough
voltage to run the converter.
For the 12 VIN converter (see Figure 1) the UVLO pin for
the high side driver is pulled up by the 5.0 V supply (through
two diode drops) and the function is not used. The diode
between the COMP pin and the 12 V supply holds the COMP
pin near GND and prevents start–up while the 12 V supply
is off. In an application where a higher UVLO threshold is
necessary a circuit like the one in Figure 15 will lock out the
converter until the 12 V supply exceeds 8.0 V.
Normal
Fast Adaptive Positioning
Slow Adaptive Positioning
Limits
Figure 14. Adaptive Positioning
The CS5301 uses two methods to provide fast and
accurate adaptive positioning. For low frequency
positioning the VFB and VDRP pins are used to adjust the
output voltage with varying load currents. For high
frequency positioning, the current sense input pins can be
used to control the power stage output impedance. The
transition between fast and slow positioning is adjusted by
the error amp compensation.
The CS5301 can be configured to adjust the output
voltage based on the output current of the converter, as
shown in Figure 1.
To set the no–load positioning, a resistor (R9) is placed
between the output voltage and VFB pin. The VFB bias
current will develop a voltage across the resistor to decrease
the output voltage. The VFB bias current is dependent on the
value of RROSC, as shown in Figure 4.
During no load conditions the VDRP pin is at the same
voltage as the VFB pin, so none of the VFB bias current flows
through the VDRP resistor (R8). When output current
increases the VDRP pin increases proportionally and the
VDRP pin current offsets the VFB bias current and causes the
output voltage to further decrease.
The VFB and VDRP pins take care of the slower and DC
voltage positioning. The first few µs are controlled primarily
by the ESR and ESL of the output filter. The transition
between fast and slow positioning is controlled by the ramp
size and the error amp compensation. If the ramp size is too
large or the error amp too slow there will be a long transition
to the final voltage after a transient. This will be most
apparent with lower capacitance output filters.
Note: Large levels of adaptive positioning can cause pulse
width jitter.
VID Codes and Power Good
The internal VID and DACOUT levels are set up so that the
reference for the control loop is nominally 125 mV below
the VID code (see the block diagram). The nominal lower
Power Good threshold is 2.5% below the DACOUT level.
The nominal upper Power Good threshold is fixed at 2.0 V
for all VID codes. This scheme is intended to select the VID
level as the maximum output voltage and the DACOUT level
as the minimum output voltage.
TRANSIENT RESPONSE AND ADAPTIVE
POSITIONING
For applications with fast transient currents the output
filter is frequently sized larger than ripple currents require in
order to reduce voltage excursions during transients.
Adaptive voltage positioning can reduce peak–peak output
voltage deviations during load transients and allow for a
smaller output filter. The output voltage can be set higher at
light loads to reduce output voltage sag when the load
current is stepped up and set lower during heavy loads to
reduce overshoot when the load current is stepped up. For
low current applications a droop resistor can provide fast
accurate adaptive positioning. However, at high currents the
loss in a droop resistor becomes excessive. For example; in
a 50 A converter a 1.0 mΩ resistor to provide a 50 mV
change in output voltage between no load and full load
would dissipate 2.5 Watts.
Lossless adaptive positioning is an alternative to using a
droop resistor, but must respond quickly to changes in load
current. Figure 14 shows how adaptive positioning works.
The waveform labeled normal shows a converter without
adaptive positioning. On the left, the output voltage sags
when the output current is stepped up and later overshoots
when current is stepped back down. With fast (ideal)
adaptive positioning the peak to peak excursions are cut in
half. In the slow adaptive positioning waveform the output
Error Amp Compensation
The transconductance error amplifier can be configured to
provide both a slow soft–start and a fast transient response.
C4 in Figure 1 controls soft–start. A 0.1 µF capacitor with
the 30 µA error amplifier output capability will allow the
output to ramp up at 0.3 V/ms or
1.5 V in 5.0 ms.
R10 is connected in series with C4 to allow the error
amplifier to slew quickly over a narrow range during load
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CS5301
transients. Here the 30 µA error amplifier output capability
works against 10 kΩ (R10) to limit the window of fast
slewing to 300 mV – enough to allow for fast transients, but
not enough to interfere with soft–start. This window will be
noticeable as a step in the COMP pin voltage at startup. The
size of this step must be kept smaller than the Channel
Startup Offset (nominally 0.4 V) for proper soft–start
operation. If adaptive positioning is used the R9 and R8 form
a divider with the VDRP end held at the DAC voltage during
startup, which effectively makes the Channel Startup Offset
larger.
C12 is included for error amp stability. A capacitive load
is required on the error amp output. Use of values less than
1.0 nF may result in error amp oscillation of several MHz.
C11 and the parallel resistance of the VFB resistor (R9)
and the VDRP resistor (R8) are used to roll off the error amp
gain. The gain is rolled off at high enough frequency to give
a quick transient response, but low enough to cross zero dB
well below the switching frequency to minimize ripple and
noise on the COMP pin.
Voltage feedback should be taken from a point of the
output or the output filter that doesn’t favor any one phase.
If the feedback connection is closer to one inductor than the
others the ripple associated with that phase may appear
larger than the ripple associated with the other phases and
poor current sharing can result.
The current sense signal is typically tens of milli–volts.
Noise pick–up should be avoided wherever possible.
Current feedback traces should be routed away from noisy
areas such as switch nodes and gate drive signals. The paths
should be matched as well as possible. It is especially
important that all current sense signals be picked off at
similar points for accurate current sharing. If the current
signal is taken from a place other than directly at the inductor
any additional resistance between the pick–off point and the
inductor appears as part of the inherent inductor resistance
and should be considered in design calculations. Capacitors
for the current feedback networks should be placed as close
to the current sense pins as practical.
DESIGN PROCEDURE
+12 V
Current Sensing, Power Stage and
Output Filter Components
+5.0 V
1. Choose the output filter components to meet peak
transient requirements. The formula below can be
used to provide an approximate starting point for
capacitor choice, but will be inadequate to calculate
actual values.
50 k
COMP
100 k
VPEAK (IT) ESL I ESR
100 k
Ideally the output filter should be simulated with
models including ESR, ESL, circuit board parasitics
and delays due to switching frequency and converter
response. Typically both bulk capacitance
(electrolytic, Oscon, etc.,) and low impedance
capacitance (ceramic chip) will be required. The bulk
capacitance provides “hold up” during the converter
response. The low impedance capacitance reduces
steady state ripple and bypasses the bulk capacitance
during slewing of output current.
2. For inductive current sensing (only) choose the
current sense network RC to provide a 25 mV
minimum ramp during steady state operation.
Figure 15. External UVLO Circuit
Layout Guidelines
With the fast rise, high output currents of microprocessor
applications, parasitic inductance and resistance should be
considered when laying out the power, filter and feedback
signal sections of the board. Typically, a multi–layer board
with at least one ground plane is recommended. If the layout
is such that high currents can exist in the ground plane
underneath the controller or control circuitry, the ground
plane can be slotted to reroute the currents away from the
controller. The slots should typically not be placed between
the controller and the output voltage or in the return path of
the gate drive. Additional power and ground planes or
islands can be added as required for a particular layout.
Output filter components should be placed on wide planes
connected directly to the load to minimize resistive drops
during heavy loads and inductive drops and ringing during
transients. If required, the planes for the output voltage and
return can be interleaved to minimize inductance between
the filter and load.
R (VIN VOUT) VOUTVIN
f C 25 mV
Then choose the inductor value and inherent
resistance to satisfy L/RL = R × C.
For ideal current sense compensation the ratio of L and
RL is fixed, so the values of L and RL will be a
compromise typically with the maximum value RL
limited by conduction losses or inductor temperature
rise and the minimum value of L limited by ripple
current.
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CS5301
Adaptive Positioning
3. For resistive current sensing choose L and RS to
provide a steady state ramp greater than 25 mV.
LRS (VIN VOUT) 7. To set the amount of voltage positioning above the
DAC setting at no load connect a resistor (RVFB)
between the output voltage and the VFB pin. Choose
RVFB as;
VOUTVIN
f 25 mV
RVFB NL PositionVFB Bias Current
Again the ratio of L and RL is fixed and the values of
L and RS will be a compromise.
4. Calculate the high frequency output impedance
(ConverterZ) of the converter during transients. This
is the impedance of the Output filter ESR in parallel
with the power stage output impedance (PwrstgZ)
and will indicate how far from the original level
(∆VR) the output voltage will typically recover to
within one switching cycle. For a good transient
response ∆VR should be less than the peak output
voltage overshoot or undershoot.
See Figure 4 for VFB Bias Current.
8. To set the difference in output voltage between no
load and full load, connect a resistor (RVDRP)
between the VDRP and VFB pins. RVDRP can be
calculated in two steps. First calculate the difference
between the VDRP and VFB pin at full load. (The VFB
voltage should be the same as the DAC voltage during
closed loop operation.) Then choose the RVDRP to
source enough current across RVFB for the desired
change in output voltage.
VR ConverterZ IOUT
VVDRP R IOUT CS to VDRP Gain
PwrstgZ ESR
ConverterZ PwrstgZ ESR
where:
R = RL or RS for one phase;
IOUT is the full load output current.
where:
PwrstgZ RS CSA Gain3
RVDRP VVDRP RV(FB)VOUT
Multiply the converterZ by the output current step
size to calculate where the output voltage should
recover to within the first switching cycle after a
transient. If the ConverterZ is higher than the value
required to recover to where the adaptive positioning
is set the remainder of the recovery will be controlled
by the error amp compensation and will typically
recover in 10–20 µs.
DESIGN EXAMPLE
Choose the component values for lossless current sensing,
adaptive positioning and current limit for a 250 kHz, 1.55 V,
60 A converter. The VID code is set to 1.6 V. Adaptive
positioning is set for 100 mV above DACOUT (or 25 mV
below VID) at no load and 75 mV below the no load position
with a 60 A load. The peak output voltage transient should
be less than 100 mV during a 60 A step current. The
overcurrent limit is nominally 75 A.
VR IOUT ConverterZ
Make sure that ∆VR is less than the expected peak
transient for a good transient response.
5. Adjust L and RL or RS as required to meet the best
combination of transient response, steady state output
voltage ripple and pulse width jitter.
Current Sensing, Power Stage and
Output Filter Components
1. Assume 1.5 mΩ of output filter ESR.
2. Choose C 0.01 F
VOUTVIN
f C 25 mV
1.5512
(12 1.55) 250 k 0.01 F 25 mV
21.5 k Choose 20 k
R (VIN VOUT) Current Limit
When the sum of the Current Sense amplifiers (VITOTAL)
exceeds the voltage on the ILIM pin the part will enter hiccup
mode. For inductive sensing the ILIM pin voltage should be
set based on the inductor resistance (or current sense
resistor) at max temperature and max current. To set the level
of the ILIM pin:
6. VILIM R IOUT(LIM) CS to ILIM Gain
where:
R is RL or RS;
IOUT(LIM) is the current limit threshold.
For the overcurrent to work properly the inductor
time constant (L/R) should be ≤ the Current sense RC.
If the RC is too fast, during step loads the current
waveform will appear larger than it is (typically for a
few hundred µs) and may trip the current limit at a
level lower than the DC limit.
LRL R C 20 k 0.01 F 200 s
Choose RL 2.0 m
L RL R C 2.0 m 200 s 400 nH
3. n/a
4. PwrstgZ RL CSA Gain3
2.0 m 4.23.0 2.8 m
ConverterZ PwrstgZ ESR
PwrstgZ ESR
2.8 m 1.5 m 1.0 m
2.8 m 1.5 m
VR ConverterZ IOUT
1.0 m 60 A 60 mV
5. n/a
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17
CS5301
8. VDRP RL IOUT
Current Limit
Current Sense to VDRP Gain
2.0 m 60 A 3.1
372 mV
6. VILIM RL IOUT(LIM)
CS to ILIM Gain
2.0 m 75 A 6.5
975 mV
RVDRP VDRP RVFBVOUT
372 mV 16.7 k75 mV
82 k
Adaptive Positioning
7. RVFB NL PositionVFB Bias Current
100 mV6.0 A 16.7 k
+12 V
+5.0 V
+12 V
D1
BAT54S
D3
BAT54S
L4
300 nH
D2
BAS16LT1
C12
1.0 nF
C1
1.0 µF
R10
10 K
C3
1.0 µF
C2
1.0 µF
+5.0 V
C11
1.0 nF
C4
R3
0.1 µF
56.2 K
C10
1.0 nF
R8
191 k
COMP
VFB
VDRP
CS1
CS2
CS3
CSREF
PWRGD
VID0
VID1
VID2
VID3
VID4
PWRGDS
ILIM
REF
CS5301
R11
10 k
+5.0 V
PWRGD
R12
20 K
VID0
VID1
VID2
Q1
CS1
L1
R4
1.0 Ω
C14
1.0 µF
U1
R9
36.5 k
+ C15,16
2 × 16SP180M
ROSC
VCCL
VCCL1
Gate(L)1
GND1
Gate(H)1
VCCH12
Gate(H)2
GND2
Gate(L)2
VCCL23
Gate(L)3
GND3
Gate(H)3
VCCH3
LGND
1.0 µF
Q2
C17–24
8 × 4SP560M
Q3
CS2
L2
1.0 µF
Q4
C25–32
8 × 10 µF
R1
VID3
3.09 k
VID4
R2
Q5
C5
0.1 µF
L3
1.0 µF
2.00 k
R7
R5
30 k
R6
30 k
C6
.01 µF
CS1
Q6
30 k
C7
.01 µF
CS3
Q1–Q6, NTD4302
CS3
C8
.01 µF
CS2
C9
.01 µF
Figure 16. Additional Application Diagram,
12 V to 1.75 V, 45 A for AMD Athlon Processor
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VCORE
CS5301
PACKAGE DIMENSIONS
SO–32L
DW SUFFIX
CASE 751P–01
ISSUE O
–X–
D
32
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS D AND E DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 PER
SIDE.
5. DIMENSION B DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 TOTAL IN EXCESS
OF B DIMENSION AT MAXIMUM MATERIAL
CONDITION.
17
H
E
0.25
–Y–
M
Y
M
16
1
PIN 1 IDENT
A
0.10
A1
B
0.25
G
M
T X
S
Y
–T–
L
SEATING
PLANE
C
M
S
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19
DIM
A
A1
B
C
D
E
G
H
L
M
MILLIMETERS
MIN
MAX
2.29
2.54
0.10
0.25
0.36
0.51
0.15
0.32
20.57
20.88
7.42
7.60
1.27 BSC
10.29
10.64
0.53
1.04
0°
8°
CS5301
V2 is a trademark of Switch Power, Inc.
Pentium is a registered trademark of Intel Corporation.
AMD Athlon is a trademark of Advanced Micro Devices, Inc.
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are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes
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CS5301/D