a FEATURES High CMV Isolation: 2500 V rms Continuous 63500 V Peak Continuous Small Size: 1.00" 3 2.10" 3 0.350" Three-Port Isolation: Input, Output, and Power Low Nonlinearity: 60.012% max Wide Bandwidth: 20 kHz Full-Power (–3 dB) Low Gain Drift: 625 ppm/8C max High CMR: 120 dB (G = 100 V/V) Isolated Power: 615 V @ 65 mA Uncommitted Input Amplifier APPLICATIONS Multichannel Data Acquisition High Voltage Instrumentation Amplifier Current Shunt Measurements Process Signal Isolation GENERAL DESCRIPTION The AD210 is the latest member of a new generation of low cost, high performance isolation amplifiers. This three-port, wide bandwidth isolation amplifier is manufactured with surface-mounted components in an automated assembly process. The AD210 combines design expertise with state-of-the-art manufacturing technology to produce an extremely compact and economical isolator whose performance and abundant user features far exceed those offered in more expensive devices. The AD210 provides a complete isolation function with both signal and power isolation supplied via transformer coupling internal to the module. The AD210’s functionally complete design, powered by a single +15 V supply, eliminates the need for an external DC/DC converter, unlike optically coupled isolation devices. The true three-port design structure permits the AD210 to be applied as an input or output isolator, in single or multichannel applications. The AD210 will maintain its high performance under sustained common-mode stress. Providing high accuracy and complete galvanic isolation, the AD210 interrupts ground loops and leakage paths, and rejects common-mode voltage and noise that may other vise degrade measurement accuracy. In addition, the AD210 provides protection from fault conditions that may cause damage to other sections of a measurement system. PRODUCT HIGHLIGHTS The AD210 is a full-featured isolator providing numerous user benefits including: High Common-Mode Performance: The AD210 provides 2500 V rms (Continuous) and ± 3500 V peak (Continuous) common- Precision, Wide Bandwidth 3-Port Isolation Amplifier AD210 FUNCTIONAL BLOCK DIAGRAM FB 16 –IN 17 +IN 19 ICOM 18 +VISS 14 –VISS 15 INPUT OUTPUT T1 DEMOD FILTER MOD T2 1 VO 2 OCOM 3 +VOSS 4 –VOSS T3 POWER OUTPUT POWER SUPPLY INPUT POWER SUPPLY POWER OSCILLATOR 30 29 PWR PWR COM AD210 mode voltage isolation between any two ports. Low input capacitance of 5 pF results in a 120 dB CMR at a gain of 100, and a low leakage current (2 µA rms max @ 240 V rms, 60 Hz). High Accuracy: With maximum nonlinearity of ± 0.012% (B Grade), gain drift of ± 25 ppm/°C max and input offset drift of (± 10 ± 30/G) µV/°C, the AD210 assures signal integrity while providing high level isolation. Wide Bandwidth: The AD210’s full-power bandwidth of 20 kHz makes it useful for wideband signals. It is also effective in applications like control loops, where limited bandwidth could result in instability. Small Size: The AD210 provides a complete isolation function in a small DIP package just 1.00" × 2.10" × 0.350". The low profile DIP package allows application in 0.5" card racks and assemblies. The pinout is optimized to facilitate board layout while maintaining isolation spacing between ports. Three-Port Design: The AD210’s three-port design structure allows each port (Input, Output, and Power) to remain independent. This three-port design permits the AD210 to be used as an input or output isolator. It also provides additional system protection should a fault occur in the power source. Isolated Power: ± 15 V @ 5 mA is available at the input and output sections of the isolator. This feature permits the AD210 to excite floating signal conditioners, front-end amplifiers and remote transducers at the input as well as other circuitry at the output. Flexible Input: An uncommitted operational amplifier is provided at the input. This amplifier provides buffering and gain as required and facilitates many alternative input functions as required by the user. REV. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703 AD210–SPECIFICATIONS (typical @ +258C, and V = +15 V unless otherwise noted) S Model AD210AN AD210BN OUTLINE DIMENSIONS AD210JN Dimensions shown in inches and (mm). GAIN Range Error vs. Temperature(0°C to +70°C) (–25°C to +85°C) vs. Supply Voltage 1 Nonlinearity 1 V/V – 100 V/V ± 2% max +25 ppm/°C max ± 50 ppm/°C max ± 0.002%/V ± 0.025% max * ± 1% max * * * ± 0.012% max * * * * * * INPUT VOLTAGE RATINGS Linear Differential Range Maximum Safe Differential Input Max. CMV Input-to-Output ac, 60 Hz, Continuous dc, Continuous Common-Mode Rejection 60 Hz, G = 100 V/V RS ≤ 500 Ω Impedance Imbalance Leakage Current Input-to-Output @ 240 V rms, 60 Hz ± 10 V ± 15 V * 2500 V rms ± 3500 V peak * * 120 dB * 2 µA rms max * * * * * * 1500 V rms ± 2000 V peak * * * * INPUT IMPEDANCE Differential Common Mode l012 Ω 5 GΩi5 pF * * * * INPUT BIAS CURRENT Initial, @ +25°C 30 pA typ (400 pA max) * vs. Temperature (0°C to +70°C) 10 nA max * (–25°C to +85°C) 30 nA max * * * * INPUT DIFFERENCE CURRENT Initial, @ +25°C vs. Temperature(0°C to + 70°C) (–25°C to +85°C) 5 pA typ (200 pA max) 2 nA max 10 nA max * * * * * * INPUT NOISE Voltage (l kHz) (10 Hz to 10 kHz) Current (1 kHz) 18 nV/√Hz 4 µV rms 0.01 pA/√Hz * * * * * * * * * * * * * * * * FREQUENCY RESPONSE Bandwidth (–3 dB) * G = 1 V/V 20 kHz G = 100 V/V 15 kHz Settling Time (± 10 mV, 20 V Step) * G = 1 V/V 150 µs G = 100 V/V 500 µs Slew Rate (G = 1 V/V) 1 V/µs OFFSET VOLTAGE (RTI)2 Initial, @ +25°C vs. Temperature (0°C to +70°C) (–25°C to +85°C) ± 15 ± 45/G) mV max (± 10 ± 30/G) µV/°C (± 10 ± 50/G) µV/°C (± 5 ±15/G) mV max * * * * * RATED OUTPUT3 Voltage, 2 kΩ Load Impedance Ripple (Bandwidth = 100 kHz) ± 10 V min 1 Ω max 10 mV p-p max * * * * * * ISOLATED POWER OUTPUTS 4 Voltage, No Load Accuracy Current Regulation, No Load to Full Load Ripple ± 15 V ± 10% ± 5 mA See Text See Text * * * * * * * * * * POWER SUPPLY Voltage, Rated Performance Voltage, Operating Current, Quiescent Current, Full Load – Full Signal +15 V dc ± 5% +15 V dc ± 10% 50 mA 80 mA * * * * * * * * TEMPERATURE RANGE Rated Performance Operating Storage –25°C to +85°C –40°C to +85°C –40°C to +85°C * * * * * * PACKAGE DIMENSIONS Inches Millimeters 1.00 × 2.10 × 0.350 25.4 × 53.3 × 8.9 * * * * NOTES *Specifications same as AD210AN. 1 Nonlinearity is specified as a % deviation from a best straight line.. 2 RTI – Referred to Input. 3 A reduced signal swing is recommended when both ± VISS and ± VOSS supplies are fully loaded, due to supply voltage reduction. 4 See text for detailed information. _ Specifications subject to change without notice. –2– AC1059 MATING SOCKET AD210 PIN DESIGNATIONS Pin Designation Function 1 2 3 4 14 15 16 17 18 19 29 30 VO OCOM +VOSS –VOSS +VISS –VISS FB –IN ICOM +IN Pwr Com Pwr Output Output Common +Isolated Power @ Output –Isolated Power @ Output +Isolated Power @ Input –Isolated Power @ Input Input Feedback –Input Input Common +Input Power Common Power Input WARNING! ESD SENSITIVE DEVICE CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD210 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. REV. A AD210 RF INSIDE THE AD210 The AD210 basic block diagram is illustrated in Figure 1. A +15 V supply is connected to the power port, and ± 15 V isolated power is supplied to both the input and output ports via a 50 kHz carrier frequency. The uncommitted input amplifier can be used to supply gain or buffering of input signals to the AD210. The fullwave modulator translates the signal to the carrier frequency for application to transformer T1. The synchronous demodulator in the output port reconstructs the input signal. A 20 kHz, three-pole filter is employed to minimize output noise and ripple. Finally, an output buffer provides a low impedance output capable of driving a 2 kΩ load. FB 16 –IN 17 +IN 19 ICOM 18 +VISS 14 –VISS 15 INPUT OUTPUT T1 DEMOD FILTER MOD T2 1 VO 2 OCOM 3 +VOSS 4 –VOSS 16 ( 1 17 VSIG AD210 2 18 14 +VISS 15 –VISS 30 30 29 PWR PWR COM +VOSS 3 –VOSS 4 29 +15V Figure 3. Input Configuration for G > 1 Figure 4 shows how to accommodate current inputs or sum currents or voltages. This circuit configuration can also be used for signals greater than ± 10 V. For example, a ± 100 V input span can be handled with RF = 20 kΩ and RS1 = 200 kΩ. IS RF 16 OUTPUT POWER SUPPLY INPUT POWER SUPPLY POWER OSCILLATOR 1 17 AD210 RS2 RS1 VS2 VS1 19 2 18 USING THE AD210 The AD210 is very simple to apply in a wide range of applications. Powered by a single +15 V power supply, the AD210 will provide outstanding performance when used as an input or output isolator, in single and multichannel configurations. 14 +VISS 15 –VISS Input Configurations: The basic unity gain configuration for signals up to ± 10 V is shown in Figure 2. Additional input amplifier variations are shown in the following figures. For smaller signal levels Figure 3 shows how to obtain gain while maintaining a very high input impedance. 16 1 17 VOUT (±10V) 19 AD210 2 18 14 VOUT +VISS +VOSS 30 VOUT = –RF VOUT AD210 Figure 1. AD210 Block Diagram VSIG ±10V ) 19 RG T3 POWER VOUT R = VSIG 1+ F RG VS1 V ( RS1 + RS2 S2 ) + IS + ... +VOSS 3 –VOSS 4 29 +15V Figure 4. Summing or Current Input Configuration Adjustments When gain and offset adjustments are required, the actual circuit adjustment components will depend on the choice of input configuration and whether the adjustments are to be made at the isolator’s input or output. Adjustments on the output side might be used when potentiometers on the input side would represent a hazard due to the presence of high common-mode voltage during adjustment. Offset adjustments are best done at the input side, as it is better to null the offset ahead of the gain. Figure 5 shows the input adjustment circuit for use when the input amplifier is configured in the noninverting mode. This offset adjustment circuit injects a small voltage in series with the GAIN 3 47.5kΩ 15 –VISS –VOSS 30 16 4 5kΩ 29 1 17 19 +15V Figure 2. Basic Unity Gain Configuration VSIG The high input impedance of the circuits in Figures 2 and 3 can be maintained in an inverting application. Since the AD210 is a three-port isolator, either the input leads or the output leads may be interchanged to create the signal inversion. AD210 RG HI LO 2 18 200Ω 14 +VISS +VOSS 3 –VOSS 4 100kΩ 50kΩ 15 OFFSET –VISS 30 29 +15V Figure 5. Adjustments for Noninverting Input REV. A –3– VOUT AD210 low side of the signal source. This will not work if the source has another current path to input common or if current flows in the signal source LO lead. To minimize CMR degradation, keep the resistor in series with the input LO below a few hundred ohms. CHANNEL OUTPUTS Figure 5 also shows the preferred gain adjustment circuit. The circuit shows RF of 50 kΩ, and will work for gains of ten or greater. The adjustment becomes less effective at lower gains (its effect is halved at G = 2) so that the pot will have to be a larger fraction of the total RF at low gain. At G = 1 (follower) the gain cannot be adjusted downward without compromising input impedance; it is better to adjust gain at the signal source or after the output. 3 2 1 0.1" GRID POWER Figure 6 shows the input adjustment circuit for use when the input amplifier is configured in the inverting mode. The offset adjustment nulls the voltage at the summing node. This is preferable to current injection because it is less affected by subsequent gain adjustment. Gain adjustment is made in the feedback and will work for gains from 1 V/V to 100 V/V. GAIN 47.5kΩ 5kΩ 1 17 RS VOUT 19 2 18 +VISS +VOSS 3 –VOSS 4 100kΩ 15 OFFSET –VISS 30 RG RF 3 2 CHANNEL INPUTS Synchronization: The AD210 is insensitive to the clock of an adjacent unit, eliminating the need to synchronize the clocks. However, in rare instances channel to channel pick-up may occur if input signal wires are bundled together. If this happens, shielded input cables are recommended. 50kΩ 14 1 Figure 8. PCB Layout for Multichannel Applications with Gain AD210 200Ω VSIG RG RF RG RF 16 29 +15V PERFORMANCE CHARACTERISTICS Figure 6. Adjustments for Inverting Input Figure 7 shows how offset adjustments can be made at the output, by offsetting the floating output port. In this circuit, ± 15 V would be supplied by a separate source. The AD210’s output amplifier is fixed at unity, therefore, output gain must be made in a subsequent stage. Common-Mode Rejection: Figure 9 shows the commonmode rejection of the AD210 versus frequency, gain and input source resistance. For maximum common-mode rejection of unwanted signals, keep the input source resistance low and carefully lay out the input, avoiding excessive stray capacitance at the input terminals. 180 16 G=1 O =0 Ω 140 AD210 RL 50kΩ 200Ω +VISS +VOSS –VISS –VOSS 30 29 CMR – dB 2 18 15 RL VOUT 19 14 G = 100 160 1 17 0.1µF 3 +15V O =0 Ω 100 RL O =1 0kΩ 100k OFFSET 4 O =5 00Ω RL 120 RL 80 O =1 0kΩ –15V 60 +15V Figure 7. Output-Side Offset Adjustment 40 10 PCB Layout for Multichannel Applications: The unique pinout positioning minimizes board space constraints for multichannel applications. Figure 8 shows the recommended printed circuit board layout for a noninverting input configuration with gain. 20 50 60 100 200 500 1k 2k 5k 10k FREQUENCY – Hz Figure 9. Common-Mode Rejection vs. Frequency –4– REV. A 0 40 –20 φG = 1 GAIN – dB 20 –40 φG = 100 0 –60 –20 –80 –40 –100 –60 –120 –80 10 100 1k 10k PHASE SHIFT – Degrees 60 ERROR – % Phase Shift: Figure 10 illustrates the AD210’s low phase shift and gain versus frequency. The AD210’s phase shift and wide bandwidth performance make it well suited for applications like power monitors and controls systems. +0.04 +8 +0.03 +6 +0.02 +4 +0.01 +2 0 0 –0.01 –2 –0.02 –4 –0.03 –6 ERROR – mV AD210 –8 –0.04 –10 –8 –6 –4 –2 0 +2 +4 +6 +8 +10 OUTPUT VOLTAGE SWING – Volts Figure 12. Gain Nonlinearity Error vs. Output 0.01 100 –140 100k 90 0.009 80 0.008 70 0.007 60 0.006 50 0.005 40 0.004 30 0.003 20 0.002 10 0.001 ERROR – ppm of Signal Swing Figure 10. Phase Shift and Gain vs. Frequency Input Noise vs. Frequency: Voltage noise referred to the input is dependent on gain and signal bandwidth. Figure 11 illustrates the typical input noise in nV/√Hz of the AD210 for a frequency range from 10 to 10 kHz. 60 NOISE – nV/√Hz 50 0.000 0 40 0 2 4 6 8 10 12 14 16 18 20 TOTAL SIGNAL SWING – Volts Figure 13. Gain Nonlinearity vs. Output Swing 30 Gain vs. Temperature: Figure 14 illustrates the AD210’s gain vs. temperature performance. The gain versus temperature performance illustrated is for an AD210 configured as a unity gain amplifier. 20 10 400 0 10 100 1k 10k 200 FREQUENCY – Hz G=1 0 GAIN ERROR – ppm of Span Figure 11. Input Noise vs. Frequency Gain Nonlinearity vs. Output: Gain nonlinearity is defined as the deviation of the output voltage from the best straight line, and is specified as % peak-to-peak of output span. The AD210B provides guaranteed maximum nonlinearity of ± 0.012% with an output span of ± 10 V. The AD210’s nonlinearity performance is shown in Figure 12. Gain Nonlinearity vs. Output Swing: The gain nonlinearity of the AD210 varies as a function of total signal swing. When the output swing is less than 20 volts, the gain nonlinearity as a fraction of signal swing improves. The shape of the nonlinearity remains constant. Figure 13 shows the gain nonlinearity of the AD210 as a function of total signal swing. –200 –400 –600 –800 –1000 –1200 –1400 –1600 –25 0 +25 +50 +70 TEMPERATURE – °C Figure 14. Gain vs. Temperature REV. A ERROR – % of Signal Swing FREQUENCY – Hz –5– +85 AD210 Isolated Power: The AD210 provides isolated power at the input and output ports. This power is useful for various signal conditioning tasks. Both ports are rated at a nominal ± 15 V at 5 mA. The load characteristics of the isolated power supplies are shown in Figure 15. For example, when measuring the load rejection of the input isolated supplies VISS, the load is placed between +VISS and –VISS. The curves labeled VISS and VOSS are the individual load rejection characteristics of the input and the output supplies, respectively. The isolated power supplies exhibit some ripple which varies as a function of load. Figure 16a shows this relationship. The AD210 has internal bypass capacitance to reduce the ripple to a point where performance is not affected, even under full load. Since the internal circuitry is more sensitive to noise on the negative supplies, these supplies have been filtered more heavily. Should a specific application require more bypassing on the isolated power supplies, there is no problem with adding external capacitors. Figure 16b depicts supply ripple as a function of external bypass capacitance under full load. There is also some effect on either isolated supply when loading the other supply. The curve labeled CROSSLOAD indicates the sensitivity of either the input or output supplies as a function of the load on the opposite supply. RIPPLE – Peak-Peak Volts 1V 30 VOLTAGE CROSSLOAD 100mV +V ( +VISS OSS ) 10mV ISS ( –V –VOSS ) 25 1mV 0.1µF VOSS VOSS Figure 16b. Isolated Power Supply Ripple vs. Bypass Capacitance (Volts p-p, 1 MHz Bandwidth, 5 mA Load) VISS SIMULTANEOUS 20 5 100µF 10µF CAPACITANCE VISS 0 1µF SIMULTANEOUS APPLICATIONS EXAMPLES 10 CURRENT – mA Figure 15. Isolated Power Supplies vs. Load Lastly, the curves labeled VOSS simultaneous and VISS simultaneous indicate the load characteristics of the isolated power supplies when an equal load is placed on both supplies. The AD210 provides short circuit protection for its isolated power supplies. When either the input supplies or the output supplies are shorted to input common or output common, respectively, no damage will be incurred, even under continuous application of the short. However, the AD210 may be damaged if the input and output supplies are shorted simultaneously. Noise Reduction in Data Acquisition Systems: Transformer coupled isolation amplifiers must have a carrier to pass both ac and dc signals through their signal transformers. Therefore, some carrier ripple is inevitably passed through to the isolator output. As the bandwidth of the isolator is increased more of the carrier signal will be present at the output. In most cases, the ripple at the AD210’s output will be insignificant when compared to the measured signal. However, in some applications, particularly when a fast analog-to-digital converter is used following the isolator, it may be desirable to add filtering; otherwise ripple may cause inaccurate measurements. Figure 17 shows a circuit that will limit the isolator’s bandwidth, thereby reducing the carrier ripple. +VOSS +VISS 16 100 R RIPPLE – mV p-p VSIG +VOSS 75 AD542 R 1 17 –VOSS 19 AD210 0.001µF VOUT 0.002µF 2 18 –VISS 50 –VOSS 14 +VISS 15 –VISS +VOSS 3 –VOSS 4 R (kΩ) = 25 30 0 0 1 2 3 4 5 6 Under any circumstances, care should be taken to ensure that the power supplies do not accidentally become shorted. C 29 +15V 7 Figure 17. 2-Pole, Output Filter LOAD – mA Figure 16a. Isolated Supply Ripple vs. Load (External 4.7 µ F Bypass) ( f 112.5 ) (kHz) Self-Powered Current Source The output circuit shown in Figure 18 can be used to create a self-powered output current source using the AD210. The 2 kΩ resistor converts the voltage output of the AD210 to an equiva–6– REV. A AD210 lent current VOUT/2 kΩ. This resistor directly affects the output gain temperature coefficient, and must be of suitable stability for the application. The external low power op amp, powered by +VOSS and –VOSS, maintains its summing junction at output common. All the current flowing through the 2 kΩ resistor flows through the output Darlington pass devices. A Darlington configuration is used to minimize loss of output current to the base. FDH333 16 +VOSS 2kΩ VSIG 0-10V 1 17 2N3906 (2) LF441 monitors the input terminal (cold-junction). Ambient temperature changes from 0°C to +40°C sensed by the AD590, are cancelled out at the cold junction. Total circuit gain equals 183; 100 and 1.83, from A1 and the AD210 respectively. Calibration is performed by replacing the thermocouple junction with plain thermocouple wire and a millivolt source set at 0.0000 V (0°C) and adjusting RO for EOUT equal to 0.000 V. Set the millivolt source to +0.02185 V (400°C) and adjust RG for VOUT equal to +4.000 V. This application circuit will produce a nonlinearized output of about +10 mV/°C for a 0°C to +400°C range. –VISS 19 –VOSS AD210 1000pF 13.7k 10k THERMAL CONTACT 2 18 RG 5k AD590 16 14 +VISS 15 –VISS 30 +VOSS 3 –VOSS 4 ADJUST TO 4mA WITH 0V IN 500Ω –VISS 14 +VISS 15 –VISS 30 +VISS +15V The AD7541 is a current output DAC and, as such, requires an external output amplifier. The uncommitted input amplifier internal to the AD210 may be used for this purpose. For best results, its input offset voltage must be trimmed as shown. +28V CURRENT LOOP The output voltage of the AD210 will go from 0 V to –10 V for digital inputs of 0 and full scale, respectively. However, since the output port is truly isolated, VOUT and OCOM may be freely interchanged to get 0 V to +10 V. +VS GAIN 2N2219 2kΩ 30 AD581 17 16 4 29 SPAN ADJ 12-BIT DIGITAL INPUT 100Ω 1N4149 CURRENT LOOP 4 15 AD7541 –VOSS +VISS +VISS 1kΩ 576Ω –VISS 18 16 1 17 1 VOUT 0 - –10V 2 3 19 AD210 200Ω 2 18 RLOAD 50kΩ +15V HP5082-2811 OR EQUIVALENT Figure 19. Isolated Voltage-to-Current Loop Converter 14 +VISS +VOSS 3 –VOSS 4 100kΩ Isolated Thermocouple Amplifier 15 OFFSET The AD210 application shown in Figure 20 provides amplification, isolation and cold-junction compensation for a standard J type thermocouple. The AD590 temperature sensor accurately REV. A 4 The digital inputs of the AD7541 are TTL or CMOS compatible. Both the AD7541 and AD581 voltage reference are powered by the isolated power supply + VISS. ICOM should be tied to input digital common to provide a digital ground reference for the inputs. 2 3 –VOSS The AD210, when combined with a digital-to-analog converter, can be used to create a fully floating voltage output. Figure 21 shows one possible implementation. –VS +VOSS 3 This circuit provides a precision 0 V–10 V programmable reference with a ± 3500 V common-mode range. AD308 +VISS 29 +VOSS Figure 20. Isolated Thermocouple Amplifier 16 1 2 Precision Floating Programmable Reference 2N2907 15 1k 100k 10k 143Ω 3.0k 14 18 -20k- Isolated V-to-I Converter Illustrated in Figure 19, the AD210 is used to convert a 0 V to +10 V input signal to an isolated 4–20 mA output current. The AD210 isolates the 0 V to +10 V input signal and provides a proportional voltage at the isolator’s output. The output circuit converts the input voltage to a 4–20 mA output current, which in turn is applied to the loop load RLOAD. 18 220pF COLD JUNCTION Figure 18. Self-Powered Isolated Current Source The low leakage diode is used to protect the base-emitter junction against reverse bias voltages. Using –VOSS as a current return allows more than 10 V of compliance. Offset and gain control may be done at the input of the AD210 or by varying the 2 kΩ resistor and summing a small correction current directly into the summing node. A nominal range of 1 mA– 5 mA is recommended since the current output cannot reach zero due to reverse bias and leakage currents. If the AD210 is powered from the input potential, this circuit provides a fully isolated, wide bandwidth current output. This configuration is limited to 5 mA output current. AD210 19 AD210 RG 19 A1 52.3Ω IOUT RETURN +15V VSIG 1 17 "J" 29 17 VOUT AD OP-07 IOUT –VISS 30 29 +15V Figure 21. Precision Floating Programmable Reference –7– AD210 200kΩ 8.25k 10T AD210 16 RG 1kΩ 17 1 19 CHANNEL 1 2 18 25Ω 14 RO 50k 15 10T RF 15.8k 16 RG 5k 17 50k 1kΩ RO 1kΩ AD590 CHANNEL 2 10T +VISS +VOSS 3 –VISS –VOSS 4 30 29 AD210 +V –V COM 1 19 2 18 –VISS C1005–9–9/86 4-20mA 9.31k 50k +VISS AD580 100Ω OFFSET 50k 14 +VISS +VOSS 3 15 –VISS –VOSS 4 10T 30 39k 29 TO A/D AD210 16 +VISS AD OP-07 AD7502 MULTIPLEXER 17 1 19 CHANNEL 3 EIN 1.0µF 0.47µF 50Ω –VISS 50kΩ +VISS 2 18 14 +VISS +VOSS 3 15 –VISS –VOSS 4 30 AD210 16 +10V +VISS 17 A2 19 AD584 20k CHANNEL 4 20k –VISS +VISS 20k 20k 1k –VISS 1 2 18 14 +VISS +VOSS 3 15 –VISS –VOSS 4 1M A1 CHANNEL SELECT 29 30 29 COM +15V A1 A2 = AD547 DC POWER SOURCE Figure 22. Multichannel Data Acquisition Front-End Illustrated in Figure 22 is a four-channel data acquisition frontend used to condition and isolate several common input signals found in various process applications. In this application, each AD210 will provide complete isolation from input to output as well as channel to channel. By using an isolator per channel, maximum protection and rejection of unwanted signals is obtained. The three-port design allows the AD210 to be configured as an input or output isolator. In this application the isolators are configured as input devices with the power port providing additional protection from possible power source faults. Channel 1: The AD210 is used to convert a 4–20 mA current loop input signal into a 0 V–10 V input. The 25 Ω shunt resistor converts the 4-20 mA current into a +100 mV to +500 mV signal. The signal is offset by –100 mV via RO to produce a 0 mV to +400 mV input. This signal is amplified by a gain of 25 to produce the desired 0 V to +10 V output. With an open circuit, the AD210 will show –2.5 V at the output. Channel 2: In this channel, the AD210 is used to condition and isolate a current output temperature transducer, Model AD590. At +25°C, the AD590 produces a nominal current of 298.2 µA. This level of current will change at a rate of 1 µA/°C. At –17.8°C (0°F), the AD590 current will be reduced by 42.8 µA to +255.4 µA. The AD580 reference circuit provides an equal but opposite current, resulting in a zero net current flow, producing a 0 V output from the AD210. At +100°C (+212°F), the AD590 current output will be 373.2 µA minus the 255.4 µA offsetting current from the AD580 circuit to yield a +117.8 µA input current. This current is converted to a voltage via RF and RG to produce an output of +2.12 V. Channel 2 will produce an output of +10 mV/°F over a 0°F to +212°F span. Channel 3: Channel 3 is a low level input channel configured with a high gain amplifier used to condition millivolt signals. With the AD210’s input set to unity and the input amplifier set for a gain of 1000, a ± 10 mV input will produce a ± 10 V at the AD210’s output. Channel 4: Channel 4 illustrates one possible configuration for conditioning a bridge circuit. The AD584 produces a +10 V excitation voltage, while A1 inverts the voltage, producing negative excitation. A2 provides a gain of 1000 V/V to amplify the low level bridge signal. Additional gain can be obtained by reconfiguration of the AD210’s input amplifier. ± VISS provides the complete power for this circuit, eliminating the need for a separate isolated excitation source. Each channel is individually addressed by the multiplexer’s channel select. Additional filtering or signal conditioning should follow the multiplexer, prior to an analog-to-digital conversion stage. –8– REV. A PRINTED IN U.S.A. MULTICHANNEL DATA ACQUISITION FRONT-END