19-4235; Rev 3; 1/10 KIT ATION EVALU E L B AVAILA High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver Features The MAX16834 is a current-mode high-brightness LED (HB LED) driver for boost, boost-buck, SEPIC, and highside buck topologies. In addition to driving an n-channel power MOSFET switch controlled by the switching controller, it also drives an n-channel PWM dimming switch to achieve LED PWM dimming. The MAX16834 integrates all the building blocks necessary to implement a fixed-frequency HB LED driver with wide-range dimming control. The MAX16834 features constant-frequency peak current-mode control with programmable slope compensation to control the duty cycle of the PWM controller. A dimming driver designed to drive an external n-channel MOSFET in series with the LED string provides wide-range dimming control up to 20kHz. In addition to PWM dimming, the MAX16834 provides analog dimming using a DC input at REFI. The programmable switching frequency (100kHz to 1MHz) allows design optimization for efficiency and board space reduction. A single resistor from RT/SYNC to ground sets the switching frequency from 100kHz to 1MHz while an external clock signal at RT/SYNC disables the internal oscillator and allows the MAX16834 to synchronize to an external clock. The MAX16834’s integrated highside current-sense amplifier eliminates the need for a separate high-side LED current-sense amplifier in boost-buck applications. o Wide Input Operating Voltage Range (4.75V to 28V) o Works for Input Voltage > 28V with External Voltage Clamp on VIN for Boost Converter o 3000:1 PWM Dimming/Analog Dimming o Integrated PWM Dimming MOSFET Driver o Integrated High-Side Current-Sense Amplifier for LED Current Sense in Boost-Buck Converter o 100kHz to 1MHz Programmable High-Frequency Operation o External Clock Synchronization Input o Programmable UVLO o Internal 7V Low-Dropout Regulator o Fault Output (FLT) for Overvoltage, Overcurrent, and Thermal Warning Faults o Programmable True Differential Overvoltage Protection o 20-Pin TQFN-EP and TSSOP-EP Packages The MAX16834 operates over a wide supply range of 4.75V to 28V and includes a 3A sink/source gate driver for driving a power MOSFET in high-power LED driver applications. It can also operate at input voltages greater than 28V in boost configuration with an external voltage clamp. The MAX16834 is also suitable for DCDC converter applications such as boost or boostbuck. Additional features include external enable/ disable input, an on-chip oscillator, fault indicator output (FLT) for LED open/short or overtemperature conditions, and an overvoltage protection sense input (OVP+) for true overvoltage protection. MAX16834AUP/V+ -40°C to +125°C 20 TSSOP-EP* +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad. /V denotes an automotive qualified part. The MAX16834 is available in a thermally enhanced 4mm x 4mm, 20-pin TQFN-EP package and in a thermally enhanced 20-pin TSSOP-EP package and is specified over the automotive -40°C to +125°C temperature range. Applications Single-String LED LCD Backlighting Automotive Rear and Front Lighting Projection System RGB LED Light Sources Architectural and Decorative Lighting (MR16, M111) Spot and Ambient Lights DC-DC Boost/Boost-Buck Converters Ordering Information PART TEMP RANGE PIN-PACKAGE MAX16834ATP+ -40°C to +125°C 20 TQFN-EP* MAX16834ATP/V+ -40°C to +125°C 20 TQFN-EP* MAX16834AUP+ -40°C to +125°C 20 TSSOP-EP* Simplified Application Circuit VIN BOOST LED DRIVER LED+ NDRV IN LEDs MAX16834 ON OFF PWMDIM ANALOG DIM CS REFI DIMOUT PGND SENSE+ LED- Pin Configurations appear at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX16834 General Description MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver ABSOLUTE MAXIMUM RATINGS IN, HV, LV to SGND................................................-0.3V to +30V 20-Pin TSSOP (derate 26.5mW/°C above +70°C) ..........2122mW OVP+, SENSE+, DIMOUT, CLV to SGND ..............-0.3V to +30V Junction-to-Ambient Thermal Resistance (θJA) (Note 1) 20-Pin TQFN 4mm x 4mm .................................................39°C/W SENSE+ to LV........................................................-0.3V to +0.3V 20-Pin TSSOP..................................................................37.7°C/W HV, IN to LV ............................................................-0.3V to +30V Junction-to-Case Thermal Resistance (θJC) (Note 1) OVP+, CLV, DIMOUT to LV ......................................-0.3V to +6V 20-Pin TQFN 4mm x 4mm ...............................................6°C/W PGND to SGND .....................................................-0.3V to +0.3V 20-Pin TSSOP..................................................................2°C/W VCC to SGND..........................................................-0.3V to +12V NDRV to PGND...........................................-0.3V to (VCC + 0.3V) Operating Temperature Range .........................-40°C to +125°C All Other Pins to SGND.............................................-0.3V to +6V Junction Temperature ......................................................+150°C NDRV Continuous Current................................................±50mA Storage Temperature Range .............................-65°C to +150°C DIMOUT Continuous Current..............................................±2mA Lead Temperature (soldering, 10s) .................................+300°C VCC Short-Circuit Current to SGND Duration ...........................1s Continuous Power Dissipation (TA = +70°C) *As per JEDEC51 standard (multilayer board). 20-Pin TQFN (4mm x 4mm) (derate 25.6mW/°C* above +70°C) ............................2051mW Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω, RRT = 10kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER Input Voltage Range SYMBOL CONDITIONS VIN Quiescent Supply Current IQ Shutdown Supply Current ISHDN MIN TYP 4.75 MAX UNITS 28 V Excluding ILED 6 10 mA VUVEN = 0 30 60 µA 7 7.7 V 0.65 1 V 300 mA 5.3 V 0.5 V 10 mA 3.775 V INTERNAL LINEAR REGULATOR (VCC) Output Voltage VCC 0 ≤ ICC ≤ 50mA, 9.5V ≤ VIN ≤ 28V Dropout Voltage VDO ICC = 35mA (Note 2) Short-Circuit Current 6.3 VCC = 0, VIN = 12V 80 0 ≤ ICLV ≤ 2mA, 6V ≤ VHV ≤ 28V, 6V ≤ V(HV-LV) ≤ 22V 4.7 LINEAR REGULATOR (CLV) Output Voltage (VCLV - VLV) Dropout Voltage VDO Short-Circuit Current 5 ICLV = 2mA, 0 ≤ VLV ≤ 23.3V (Note 3) VCLV = 12V, VIN = 12V, VHV = 24V 2.2 0 ≤ IREF ≤ 1mA, 4.75V ≤ VIN ≤ 28V 3.625 REFERENCE VOLTAGE (REF) Output Voltage VREF REF Short-Circuit Current VREF = 0 3.70 30 mA UNDERVOLTAGE LOCKOUT/ENABLE INPUT (UVEN) UVEN On Threshold Voltage VUVEN_THUP 1.395 UVEN Threshold Voltage Hysteresis Input Leakage Current ILEAK VUVEN = 0 1.435 1.475 V 200 mV I1I µA PWMDIM PWMDIM On Threshold Voltage PWMDIM Threshold Voltage Hysteresis 2 VPWMDIM 1.395 1.435 200 _______________________________________________________________________________________ 1.475 V mV High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver (VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω, RRT = 10kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER SYMBOL Input Leakage Current CONDITIONS MIN VPWMDIM = 0 TYP MAX I1I UNITS µA OSCILLATOR Oscillator Frequency fOSC RRT/SYNC = 5kΩ 0.9 1 200 RRT/SYNC = 25kΩ 180 Oscillator Frequency Range (Note 4) 100 External Sync Input Clock High Threshold (Note 4) 2 External Sync Input Clock Low Threshold (Note 4) External Sync Input High Pulse Width (Note 4) 1.1 220 kHz 1000 kHz V 0.4 200 Maximum External Sync Period MHz V ns 50 µs SLOPE COMPENSATION (SC) SC Pullup Current ISCPU VSC = 100mV 80 100 120 µA SC Discharge Resistance RSCD VSC = 100mV 8 Ω REFI Input Bias Current VREFI = 1V I1I µA REFI Input Common-Mode Range (Note 4) REFI 0 2 V 250 µA SENSE+ SENSE+ Input Bias Current (VSENSE+ - VLV) = 100mV HIGH-SIDE LED CURRENT-SENSE AMPLIFIER (VSENSE+ - VLV) Input Offset Voltage Voltage Gain AV 3dB Bandwidth VLV > 5V, (VSENSE+ - VLV) = 5mV -2.4 0 +2.4 mV VLV > 5V, (VSENSE+ - VLV) = 0.2V 9.7 9.9 10.1 V/V (VSENSE+ - VLV) = 0.1V, no load 1.8 MHz (VSENSE+ - VLV) = 0.02V, no load 600 kHz LOW-SIDE LED CURRENT-SENSE AMPLIFIER Input Offset Voltage Voltage Gain AV VLV < 1V, (VSENSE+ - VLV) = 0V -2 0 +2 VLV < 1V, (VSENSE+ - VLV) = 0.2V 9.7 9.9 10.1 3dB Bandwidth 600 mV V/V kHz CURRENT ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER) Transconductance gm Open-Loop DC Gain AV VCOMP = 2V, VPWMDIM = 5V -10 VCOMP 500 600 60 Input Offset Voltage COMP Voltage Range 400 (Note 4) 0 0.4 µS dB +10 mV 2.5 V PWM COMPARATOR Input Offset Voltage Propagation Delay 0.6 tPD 50mV overdrive 0.65 40 0.70 V ns _______________________________________________________________________________________ 3 MAX16834 ELECTRICAL CHARACTERISTICS (continued) MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver ELECTRICAL CHARACTERISTICS (continued) (VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω, RRT = 10kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER Minimum On-Time SYMBOL tON(MIN) Duty Cycle CONDITIONS MIN On-time includes blanking time (Note 4) TYP MAX 100 90 UNITS ns 99.5 % CURRENT PEAK LIMIT COMPARATOR Trip Threshold Voltage 0.25 Propagation Delay 0.3 50mV overdrive with respect to NDRV 0.35 40 V ns OVERVOLTAGE PROTECTION INPUT (OVP+) OVP+ On Threshold Voltage VOVP_ON 1.375 OVP+ Hysteresis OVP+ Input Leakage Current 1.435 1.495 200 (VOVP - VLV) = 1.235V -1 Off Threshold VCLV - VLV 4.0 On Threshold VCLV - VLV 4.1 Error Reject Blankout fOSC = 500kHz V mV +1 µA 4.3 4.6 V 4.4 4.7 HIGH-SIDE LED SHORT COMPARATOR 256 V µs LOW-SIDE LED SHORT COMPARATOR Off Threshold 0.27 0.30 Error Reject Blankout 0.33 V 5 µs 8.2 ms 3 A HICCUP TIMER Hiccup Time fOSC = 500kHz GATE-DRIVER OUTPUT (NDRV) NDRV Peak Pullup Current VCC = 7V NDRV Peak Pulldown Current VCC = 7V p-Channel MOSFET RDSON (VCC - VNDRV) = 0.1V 1.2 1.9 Ω n-Channel MOSFET RDSON VNDRV = 0.1V 0.9 1.7 Ω 3 A DIMOUT DIMOUT Peak Pullup Current (VCLV - VLV) = 5V 25 50 mA DIMOUT Peak Pulldown Current (VCLV - VLV) = 5V 25 50 mA p-Channel MOSFET RDSON (VCLV - VDIMOUT) = 0.1V 31 Ω n-Channel MOSFET RDSON (VDIMOUT - VLV) = 0.1V 25 Ω 200 ns PWMDIM to DIMOUT Propagation Delay FAULT FLAG (FLT) FLT Pulldown Current VFLT = 0.2V FLT Leakage Current VFLT = 1.0V Thermal Warning On Threshold 2 5 mA I1I µA +140 °C 20 °C Thermal Warning Threshold Hysteresis Note 2: Dropout voltage is defined as VIN - VCC, when VCC is 100mV below the value of VCC for VIN = 9.5V. Note 3: Dropout is defined as VHV - VCLV, when VCLV is 100mV below the value of VCLV for VHV = 8V. Note 4: Not production tested. Guaranteed by design. 4 10 _______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver VREF vs. SUPPLY VOLTAGE 3.74 3.72 3.75 VIN = 12V 3.7015 3.7010 3.70 3.68 VREF (V) 3.70 VREF (V) VREF (V) VREF vs. IREF 3.7020 MAX16834 toc02 MAX16834 toc01 3.80 MAX16834 toc03 VREF vs. TEMPERATURE 3.65 3.66 3.7005 3.7000 3.6995 3.60 3.64 3.6990 3.55 3.62 3.6985 VIN = 12V 3.50 12 16 20 24 SUPPLY VOLTAGE (V) SUPPLY CURRENT vs. SUPPLY VOLTAGE SUPPLY CURRENT vs. TEMPERATURE 12 10 8 6 MAX16834 toc05 8 6 5 1 20 24 SWITCHING FREQUENCY vs. TEMPERATURE TEMPERATURE (°C) 7.16 7.14 7.12 7.10 7.08 7.06 7.04 7.02 7.00 6.98 6.96 6.94 6.92 6.90 10 9 1000 VCC vs. ICC VIN = 12V 7.2 VIN = 12V TA = +125°C TA = +100°C 7.1 VCC (V) VCC (V) MAX16834 toc07 -40 -25 -10 5 20 35 50 65 80 95 110 125 8 SWITCHING FREQUENCY (kHz) VCC vs. ICC VIN = 12V 7 100 TEMPERATURE (°C) SUPPLY VOLTAGE (V) 605 604 603 602 601 600 599 598 597 596 595 594 593 592 591 590 6 VIN = 12V 1 -40 -25 -10 5 20 35 50 65 80 95 110 125 28 MAX16834 toc08 16 5 10 VIN = 12V PWMDIM = 0 0 12 4 3 2 8 3 4 2 4 2 7 4 0 1 100 RT (kΩ) 14 0 RT vs. SWITCHING FREQUENCY 9 SUPPLY CURRENT (mA) 16 28 IREF (mA) 10 MAX16834 toc04 PWMDIM = 0 18 SWITCHING FREQUENCY (kHz) 8 TEMPERATURE (°C) 20 SUPPLY CURRENT (mA) 3.6980 4 MAX16834 toc06 -40 -25 -10 5 20 35 50 65 80 95 110 125 MAX16834 toc09 3.60 7.0 TA = +25°C TA = -40°C 6.9 6.8 0 10 20 30 40 50 60 70 80 90 100 ICC (mA) 0 10 20 30 40 50 60 70 80 90 100 ICC (mA) _______________________________________________________________________________________ 5 MAX16834 Typical Operating Characteristics (VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω, RRT = 10kΩ, TA = +25°C, unless otherwise noted.) Typical Operating Characteristics (continued) (VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω, RRT = 10kΩ, TA = +25°C, unless otherwise noted.) NDRV RISE/FALL TIME vs. CAPACITANCE VCC vs. VIN TA = +25°C TA = +125°C TA = -40°C 7.14 VCC (V) VIN = 12V 40 NDRV RISE TIME (ns) 7.16 50 7.12 7.10 7.08 7.06 MAX16834 toc11 7.18 MAX16834 toc10 7.20 30 RISE TIME 20 FALL TIME 10 7.04 7.02 0 7.00 10 14 18 22 0 26 1 2 4.50 5 6 7 8 9 5.10 MAX16834 toc12 5.00 4 10 VCLV vs. VHV VCLV vs. ICLV 5.50 3 CAPACITANCE (nF) VIN (V) VIN = 12V 5.09 5.08 4.00 MAX16834 toc13 6 5.07 VCLV (V) 3.50 VCLV (V) MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver 3.00 2.50 5.06 5.05 5.04 2.00 5.03 1.50 5.02 1.00 0.50 5.01 VIN = 12V 5.00 0 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 6 10 14 18 22 26 VHV (V) ICLV (mA) Pin Description PIN 6 NAME FUNCTION TQFN TSSOP 1 3 OVP+ LED-String Overvoltage Protection Input. Connect a resistive voltage-divider between the positive output, OVP+, and LV to set the overvoltage threshold. OVP+ has a 1.435V threshold voltage with a 200mV hysteresis. 2 4 SGND Signal Ground 3 5 COMP Error-Amplifier Output. Connect an RC network from COMP to SGND for stable operation. See the Feedback Compensation section. 4 6 REF 3.7V Reference Output Voltage. Bypass REF to SGND with a 0.1µF to 0.22µF ceramic capacitor. 5 7 REFI Current Reference Input. VREFI provides a reference voltage for the current-sense amplifier to set the LED current. 6 8 SC Current-Mode Slope Compensation Setting. Connect to an appropriate external capacitor from SC to SGND to generate a ramp signal for stable operation. _______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver PIN TQFN TSSOP 7 9 NAME FLT FUNCTION Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (FLT) section. 8 10 RT/SYNC Resistor-Programmable Switching Frequency Setting/Sync Control Input. Connect a resistor from RT/SYNC to SGND to set the switching frequency. Drive RT/SYNC to synchronize the switching frequency with an external clock. 9 11 UVEN Undervoltage-Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable UVLO threshold input with an enable feature. Connect UVEN to VIN through a resistive voltage-divider to program the UVLO threshold. Observe the absolute maximum value for this pin. 10 12 PWMDIM PWM Dimming Input. Connect to an external PWM signal for dimming operation. Current-Sense Amplifier Positive Input. Connect a resistor from CS to PGND to set the inductor peak current limit. 11 13 CS 12 14 PGND Power Ground 13 15 NDRV External n-Channel Gate-Driver Output 14 16 VCC 15 17 IN 16 18 HV 17 19 CLV 18 20 DIMOUT 19 1 LV 20 2 SENSE+ — — EP 7V Low-Dropout Voltage Regulator. Bypass to PGND with at least a 1µF low-ESR ceramic capacitor. VCC provides power to the n-channel gate driver (NDRV). Positive Power-Supply Input. Bypass to PGND with at least a 0.1µF ceramic capacitor. High-Side Positive Supply Input Referred to LV. HV provides power to high-side linear regulator 5V High-Side Regulator Output. CLV provides power to the dimming MOSFET driver. Connect a 0.1µF to 1µF ceramic capacitor from CLV to LV for stable operation. External Dimming MOSFET Gate Driver. DIMOUT is capable of sinking/sourcing 50mA. High-Side Reference Voltage Input. Connect to SGND for boost configuration. Connect to IN for boost-buck configuration. LED Current-Sense Positive Input. Connect a bypass capacitor of at least 0.1µF between SENSE+ and LV close to the IC. Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power dissipation. Do not use as the main IC ground connection. EP must be connected to SGND. Detailed Description The MAX16834 is a current-mode, high-brightness LED (HB LED) driver designed to control a single-string LED current regulator with two external n-channel MOSFETs. The MAX16834 integrates all the building blocks necessary to implement a fixed-frequency HB LED driver with wide-range dimming control. The MAX16834 allows implementation of different converter topologies such as SEPIC, boost, boost-buck, or high-side buck current regulator. The MAX16834 features a constant-frequency, peak-current-mode control with programmable slope compensation to control the duty cycle of the PWM controller. A dimming driver offers a wide-range dimming control for the external n-channel MOSFET in series with the LED string. In addition to PWM dimming, the MAX16834 allows for analog dimming of LED current. The MAX16834 switching frequency (100kHz to 1MHz) is adjustable using a single resistor from RT/SYNC. The MAX16834 disables the internal oscillator and synchronizes if an external clock is applied to RT/SYNC. The switching MOSFET driver sinks and sources up to 3A, making it suitable for high-power MOSFETs driving in HB LED applications, and the dimming control allows for wide PWM dimming at frequencies up to 20kHz. The MAX16834 is suitable for boost and boost-buck LED drivers (Figures 2 and 3). The MAX16834 alone operates over a wide 4.75V to 28V supply range. With a voltage clamp that limits the IN pin voltage to less than 28V, it can operate in boost configuration for input voltages greater than 28V. Additional features include external enable/disable input, an on-chip oscillator, fault indicator output (FLT) for LED open/short or overtemperature conditions, and an overvoltage protection circuit for true differential overvoltage protection (Figure 1). _______________________________________________________________________________________ 7 MAX16834 Pin Description (continued) MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver IN REF TO INTERNAL CIRCUITRY REFERENCE TEMPERATURE SENSE OT SGND VCC 7V LDO UVEN UVLO VBG S Q R RT/SYNC OSC SC 0.6V 5kΩ CS NDRV RAMP GENERATOR PWM COMP OR AND CURRENT-LIMIT COMPARATOR BLANK NDRVB PGND NDRVB 0.3V VREF REFI FLTB LPF FLT ERROR AMPLIFIER SENSE+ AV = 9.9 PWMDIM gm VLV FLTA AND AND OT LED CURRENTSENSE AMPLIFIERS CLV COMP DIMOUT HV HIGH-SIDE 5V REGULATOR VBG LV REFERENCE SWITCH LV VLV REFHI VIN VBG PWMDIM VBG FLTB AND VLV 0.3V VHV OVP+ FLTA VLV 128 TOSC ERROR REJECT DELAY 4.3V SENSE+ VREF REFHI 4096 TOSC HICCUP TIMER 5µs ERROR REJECT DELAY MAX16834 VBG VLV Figure 1. Internal Block Diagram 8 _______________________________________________________________________________________ FLTB High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver Undervoltage Lockout/Enable The MAX16834 features an adjustable UVLO using the enable input (UVEN). Connect UVEN to VIN through a resistive divider to set the UVLO threshold. The MAX16834 is enabled when the VUVEN exceeds the 1.435V (typ) threshold. See the Setting the UVLO Threshold section for more information. UVEN also functions as an enable/disable input to the device. Drive UVEN low to disable the output and high to enable the output. Reference Voltage (REF) The MAX16834 features a 3.7V reference output, REF. REF provides power to most of the internal circuit blocks except for the output drivers and is capable of sourcing 1mA to external circuits. Connect a 0.1µF to 0.22µF ceramic capacitor from REF to SGND. Connect REF to REFI through a resistive divider to set the LED current. Reference Input (REFI) The output current is proportional to the voltage at REFI. Applying an external DC voltage at REFI or using a potentiometer from REF to SGND allows analog dimming of the output current. High-Side Reference Voltage Input (LV) LV is a reference input. Connect LV to SGND for boost and SEPIC topologies. Connect LV to IN for boost-buck and high-side buck topologies. Dimming Driver Regulator Input Voltage (HV) The voltage at HV provides the input voltage for the dimming driver regulator. For boost or SEPIC topology, connect HV either to IN or to VCC. For boost-buck, connect HV to a voltage higher than IN. The voltage at HV must not exceed 28V with respect to PGND. For the high-side buck, connect HV to IN. Dimming MOSFET Driver (DIMOUT) The MAX16834 requires an external n-channel MOSFET for PWM dimming. Connect the gate of the MOSFET to the output of the dimming driver, DIMOUT, for normal operation. The dimming driver is capable of sinking or sourcing up to 50mA of current. n-Channel MOSFET Switch Driver (NDRV) The MAX16834 drives an external n-channel switching MOSFET. NDRV swings between V CC and PGND. NDRV is capable of sinking/sourcing 3A of peak current, allowing the MAX16834 to switch MOSFETs in highpower applications. The average current demanded from the supply to drive the external MOSFET depends on the total gate charge (Q G ) and the operating frequency of the converter, fSW. Use the following equation to calculate the driver supply current I NDRV required for the switching MOSFET: INDRV = QG x fSW Pulse Dimming Inputs (PWMDIM) The MAX16834 offers a dimming input (PWMDIM) for pulse-width modulating the output current. PWM dimming can be achieved by driving PWMDIM with a pulsating voltage source. When the voltage at PWMDIM is greater than 1.435V, the PWM dimming MOSFET turns on and when the voltage on PWMDIM is below 1.235V, the PWM dimming MOSFET turns off. High-Side Linear Regulator (VCLV) The MAX16834’s 5V high-side regulator (CLV) powers up the dimming MOSFET driver. VCLV is measured with respect to LV and sources up to 2mA of current. Bypass CLV to LV with a 0.1µF to 1µF low-ESR ceramic capacitor. The maximum voltage on CLV with respect to PGND must not exceed 28V. This limits the input voltage for boost-buck topology. Low-Side Linear Regulator (VCC) The MAX16834’s 7V low-side linear regulator (VCC) powers up the switching MOSFET driver with sourcing capability of up to 50mA. Use at least a 1µF low-ESR ceramic capacitor from VCC to PGND for stable operation. LED Current-Sense Input (SENSE+) The differential voltage from SENSE+ to LV is fed to an internal current-sense amplifier. This amplified signal is then connected to the negative input of the transconductance error amplifier. The voltage gain factor of this amplifier is 9.9 (typ). Whenever VLV is greater than 5V, the input impedance of the LED current-sense amplifier seen at the SENSE+ pin is 1kΩ ±30%. In that condition, a bias current of 20µA (±30%) is pulled from SENSE+, in addition to the current due to the 1kΩ resistor. When VLV is less than 1V, the amplifier input (SENSE+ pin) is in high impedance and the bias current of 20µA (±30%) is pushed out of that pin. Always have a bypass capacitor of at least 0.1µF value between SENSE+ and LV and close to the IC. _______________________________________________________________________________________ 9 MAX16834 The MAX16834 is also suitable for DC-DC converter applications such as boost or boost-buck (Figures 6 and 7). Other applications include boost LED drivers with automotive load dump protection (Figure 4) and high-side buck LED drivers (Figure 5). MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver Internal Transconductance Error Amplifier The MAX16834 has a built-in transconductance amplifier used to amplify the error signal inside the feedback loop. The amplified current-sense signal is connected to the negative input of the gm amplifier with the current reference connected to REFI. The output of the op amp is controlled by the input at PWMDIM. When the signal at PWMDIM is high, the output of the op amp connects to COMP; when the signal at PWMDIM is low, the output of the op amp disconnects from COMP to preserve the charge on the compensation capacitor. When the voltage at PWMDIM goes high, the voltage on the compensation capacitor forces the converter into a steady state. COMP is connected to the negative input of the PWM comparator with CMOS inputs, which draw very little current from the compensation capacitor at COMP and thus prevent discharge of the compensation capacitor when the PWMDIM input is low. Internal Oscillator The internal oscillator of the MAX16834 is programmable from 100kHz to 1MHz using a single resistor at RT/SYNC. Use the following formula to calculate the switching frequency: fOSC (kHz) = 5000kΩ × (kHz) RT(kΩ) where RT is the resistor from RT/SYNC to SGND. The MAX16834 synchronizes to an external clock signal at RT/SYNC. The application of an external clock disables the internal oscillator allowing the MAX16834 to use the external clock for switching operation. The internal oscillator is enabled if the external clock is absent for more than 50µs. The synchronizing pulse minimum width for proper synchronization is 200ns. Switching MOSFET Current-Sense Input (CS) CS is part of the current-mode control loop. The switching control uses the voltage on CS, set by RCS, to terminate the on pulse width of the switching cycle, thus achieving peak current-mode control. Internal leadingedge blanking is provided to prevent premature turn-off of the switching MOSFET in each switching cycle. Slope Compensation (SC) The MAX16834 uses an internal-ramp generator for slope compensation. The ramp signal also resets at the beginning of each cycle and slews at the rate pro- 10 grammed by the external capacitor connected at SC. The current source charging the capacitor is 100µA. Overvoltage Protection (OVP+) OVP+ sets the overvoltage threshold limit across the LEDs. Use a resistive divider between output OVP+ and LV to set the overvoltage threshold limit. An internal overvoltage protection comparator senses the differential voltage across OVP+ and LV. If the differential voltage is greater than 1.435V, NDRV is disabled and FLT asserts. When the differential voltage drops by 200mV, NDRV is enabled and FLT deasserts. The PWM dimming MOSFET is still controlled by the PWMDIM input. Fault Indicator (FLT) The MAX16834 features an active-low, open-drain fault indicator (FLT). FLT asserts when one of the following occurs: 1) Overvoltage across the LED string 2) Short-circuit condition across the LED string, or 3) Overtemperature condition When the output voltage drops below the overvoltage set point minus the hysteresis, FLT deasserts. Similarly during the short-circuit period, the fault signal deasserts when the dimming MOSFET is on, which happens every hiccup cycle during short circuit. During overtemperature fault, the FLT signal is the inverse of the PWM input. Applications Information Setting the UVLO Threshold The UVLO threshold is set by resistors R1 and R2 (see Figure 2). The MAX16834 turns on when the voltage across R2 exceeds 1.435V, the UVLO threshold. Use the following equation to set the desired UVLO threshold: VUVEN = 1.435V(R1 + R2) R2 In a typical application, use a 10kΩ resistor for R2 and then calculate R1 based on the desired UVLO threshold. Setting the Overvoltage Threshold The overvoltage threshold is set by resistors R4 and R9 (see Figure 2). The overvoltage circuit in the MAX16834 is activated when the voltage on OVP+ with respect to LV exceeds 1.435V. Use the following equation to set the desired overvoltage threshold: VOV = 1.435V(R4 + R9) R9 ______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver MAX16834 VIN C1 L1 R1 LV D1 FLT LED+ C3 IN Q1 NDRV LEDs UVEN HV SC R4 CS C2 ON MAX16834 OFF PWMDIM R3 LED- RT/SYNC C5 Q2 DIMOUT R2 VCC C4 REF SENSE+ OVP+ R6 CLV R5 REFI COMP SGND PGND R8 R9 R10 C8 C7 C6 R7 Figure 2. Boost LED Driver Programming the LED Current The LED current is programmed using the voltage on REFI and the LED current-sense resistor R10 (see Figure 2). The current is given by: ILED = VREF × R5 ( A) R10 × (R6 + R5) × 9.9 where VREF is 3.7V and the resistors R5, R6, and R10 are in ohms. The regulation voltage on the LED currentsense resistor must not exceed 0.3V to prevent activation of the LED short-circuit protection circuit. Boost Configuration In the boost converter (Figure 2), the average inductor current varies with the line voltage. The maximum average current occurs at the lowest line voltage. For the boost converter, the average inductor current is equal to the input current. Calculate maximum duty cycle using the below equation. D MAX = VLED + VD − VINMIN VLED + VD − VFET where VLED is the forward voltage of the LED string in volts, VD is the forward drop of the rectifier diode D1 in volts (approximately 0.6V), VINMIN is the minimum input supply voltage in volts, and VFET is the average drain to source voltage of the MOSFET Q1 in volts when it is on. Use an approximate value of 0.2V initially to calculate DMAX. A more accurate value of the maximum duty cycle can be calculated once the power MOSFET is selected based on the maximum inductor current. Use the following equations to calculate the maximum average inductor current ILAVG, peak-to-peak inductor current ripple ∆IL, and the peak inductor current ILP in amperes: ______________________________________________________________________________________ 11 MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver ILED IL AVG = 1 − D MAX Allowing the peak-to-peak inductor ripple ∆I L to be ±30% of the average inductor current: ∆IL = IL AVG × 0.3 × 2 Allowing the peak-to-peak inductor ripple (∆IL) to be ±30% of the average inductor current: ∆IL = IL AVG × 0.3 × 2 and IL P = IL AVG + ∆IL 2 The inductance value (L) of the inductor L1 in henries (H) is calculated as: L= (VINMIN − VFET ) × D MAX fSW × ∆IL IL P = IL AVG + ∆IL 2 The inductance value (L) of the inductor L1 in henries is calculated as: L= (VINMIN − VFET ) × D MAX fSW × ∆IL where fSW is the switching frequency in hertz, VINMIN and VFET are in volts, and ∆IL is in amperes. Choose an inductor that has a minimum inductance greater than the calculated value. Peak Current-Sense Resistor (R8) where fSW is the switching frequency in hertz, VINMIN and VFET are in volts, and ∆IL is in amperes. Choose an inductor that has a minimum inductance greater than the calculated value. The current rating of the inductor should be higher than ILP at the operating temperature. The value of the switch current-sense resistor R8 for the boost and boost-buck configurations is calculated as follows: Boost-Buck Configuration where 0.25V is the minimum peak current-sense threshold, ILP is the peak inductor current in amperes, and the factor 1.25 provides a 25% margin to account for tolerances. The worst cycle-by-cycle current limiter triggers at 350mV (max). The ISAT of the inductor should be higher than 0.35V/R8. In the boost-buck LED driver (Figure 3), the average inductor current is equal to the input current plus the LED current. Calculate maximum duty cycle using the following equation: D MAX = VLED + VD VLED + VD + VINMIN − VFET where VLED is the forward voltage of the LED string in volts, VD is the forward drop of the rectifier diode D1 (approximately 0.6V) in volts, VINMIN is the minimum input supply voltage in volts, and VFET is the average drain to source voltage of the MOSFET Q1 in volts when it is on. Use an approximate value of 0.2V initially to calculate DMAX. A more accurate value of maximum duty cycle can be calculated once the power MOSFET is selected based on the maximum inductor current. Use the below equations to calculate the maximum average inductor current ILAVG, peak-to-peak inductor current ripple ∆IL, and the peak inductor current ILP in amperes: IL AVG = 12 ILED 1 − D MAX R8 = 0.25 Ω (IL P × 1.25) Output Capacitor The function of the output capacitor is to reduce the output ripple to acceptable levels. The ESR, ESL, and the bulk capacitance of the output capacitor contribute to the output ripple. In most applications, the output ESR and ESL effects can be dramatically reduced by using low-ESR ceramic capacitors. To reduce the ESL and ESR effects, connect multiple ceramic capacitors in parallel to achieve the required bulk capacitance. To minimize audible noise generated by the ceramic capacitors during PWM dimming, it may be necessary to minimize the number of ceramic capacitors on the output. In these cases an additional electrolytic or tantalum capacitor provides most of the bulk capacitance. Boost and boost-buck configurations: The calculation of the output capacitance is the same for both boost and boost-buck configurations. The output ripple is caused by the ESR and the bulk capacitance of the output capacitor if the ESL effect is considered negligible. For simplicity, assume that the contributions from ______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver MAX16834 VIN C1 L1 D1 R1 LV HV IN NDRV UVEN Q1 LEDs CS C2 SC LED+ R3 C3 ON MAX16834 R4 OFF PWMDIM LED- RT/SYNC C5 R2 VCC DIMOUT REF SENSE+ Q2 C4 R6 OVP+ R5 REFI FLT SGND CLV COMP PGND R8 R9 R10 C8 C7 C6 R7 VIN Figure 3. Boost-Buck LED Driver (VLED+ < 28V) ESR and the bulk capacitance are equal, allowing 50% of the ripple for the bulk capacitance. The capacitance is given by: C OUT ≥ ILED × 2 × D MAX ∆VOUTRIPPLE × fSW where ILED is in amperes, COUT is in farads, fSW is in hertz, and ∆VOUTRIPPLE is in volts. The remaining 50% of allowable ripple is for the ESR of the output capacitor. Based on this, the ESR of the output capacitor is given by: ESR COUT < ∆VOUTRIPPLE (Ω) (IL P × 2) where ILP is the peak inductor current in amperes. Use the below equation to calculate the RMS current rating of the output capacitor: ICOUT(RMS) = (IL AVG × (1 - DMAX )) 2 × DMAX +(IL AVG × DMAX ) 2 × (1 - DMAX ) Input Capacitor The input filter capacitor bypasses the ripple current drawn by the converter and reduces the amplitude of high-frequency current conducted to the input supply. The ESR, ESL, and the bulk capacitance of the input capacitor contribute to the input ripple. Use a low-ESR input capacitor that can handle the maximum input RMS ripple current from the converter. For the boost configuration, the input current is the same as the inductor current. For boost-buck ______________________________________________________________________________________ 13 MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver configuration, the input current is the inductor current minus the LED current. But for both configurations, the ripple current that the input filter capacitor has to supply is the same as the inductor ripple current with the condition that the output filter capacitor should be connected to ground for boost-buck configuration. This reduces the size of the input capacitor, as the inductor current is continuous with maximum ±30% ripple. Neglecting the effect of LED current ripple, the calculation of the input capacitor for boost as well as boostbuck configurations is the same. Neglecting the effect of the ESL, the ESR, and the bulk capacitance at the input contributes to the input voltage ripple. For simplicity, assume that the contribution from the ESR and the bulk capacitance is equal. This allows 50% of the ripple for the bulk capacitance. The capacitance is given by: CIN ≥ ∆IL 4 × ∆VIN × fSW where ∆IL is in amperes, CIN is in farads, fSW is in hertz, and ∆VIN is in volts. The remaining 50% of allowable ripple is for the ESR of the output capacitor. Based on this, the ESR of the input capacitor is given by: ∆VIN ESR CIN < ∆IL × 2 where ∆IL is in amperes, ESRCIN is in ohms, and ∆VIN is in volts. Use the below equation to calculate the RMS current rating of the input capacitor: ICIN(RMS) = ∆IL 2 3 Slope Compensation Slope compensation should be added to converters with peak current-mode control operating in continuous conduction mode with more than 50% duty cycle to avoid current loop instability and subharmonic oscillations. The minimum amount of slope added to the peak inductor current to stabilize the current control loop is half of the falling slope of the inductor. In the MAX16834, the slope compensating ramp is added to the current-sense signal before it is fed to the PWM comparator. Connect a capacitor (C2 in the application circuit) from SC to ground for slope compensation. This capacitor is charged with a 100µA current 14 source and discharged at the beginning of each switching cycle to generate the slope compensation ramp. The value of the slope compensation capacitor C2 is calculated as shown below: Boost configuration: C2 = 3 × L × 100 × 10 -6 (VLED - VINMIN ) × R8 × 2 where C2 is in farads, L is the inductance of the inductor L1 in henries, 100µA is the pullup current from SC, VLED and VINMIN are in volts, and R8 is the switch current-sense resistor in ohms. Boost-buck configuration: C2 = 3 × L × 100 × 10 -6 (VLED ) × R8 × 2 where C2 is in farads, L is the inductance of the inductor L1 in henries, 100µA is the pullup current from SC, VLED is in volts, and R8 is the switch current-sense resistor in ohms. Selection of Power Semiconductors Switching MOSFET The switching MOSFET (Q1) should have a voltage rating sufficient to withstand the maximum output voltage together with the diode drop of the rectifier diode D1 and any possible overshoot due to ringing caused by parasitic inductances and capacitances. Use a MOSFET with a drain-to-source voltage rating higher than that calculated by the following equations: Boost configuration: VDS = ( VLED + VD ) × 1.2 where VDS is the drain-to-source voltage in volts and VD is the forward drop of the rectifier diode D1. The factor of 1.2 provides a 20% safety margin. Boost-buck configuration: VDS = ( VLED + VINMAX + VD ) × 1.2 where VDS is the drain-to-source voltage in volts and VD is the forward drop of the rectifier diode D1. The factor of 1.2 provides a 20% safety margin. The continuous drain current rating of the selected MOSFET, when the case temperature is at +70°C, should be greater than the value calculated by the fol- ______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver ⎛ ID RMS = ⎜ ⎝ (IL AVG ) 2 × DMAX ⎞⎟⎠ × 1.3 Rectifier Diode Use a Schottky diode as the rectifier (D1) for fast switching and to reduce power dissipation. The selected Schottky diode must have a voltage rating 20% above the maximum converter output voltage. The maximum converter output voltage is VLED in boost configuration and VLED + VINMAX in boost-buck configuration. The current rating of the diode should be greater than ID in the following equation: where IDRMS is the MOSFET Q1’s drain RMS current in amperes. ID = IL AVG × (1 - DMAX ) × 1.5 The MOSFET Q1 will dissipate power due to both switching losses as well as conduction losses. The conduction losses in the MOSFET is calculated as follows: Dimming MOSFET Select a dimming MOSFET (Q2) with continuous current rating at +70°C, higher than the LED current by 30%. The drain-to-source voltage rating of the dimming MOSFET must be higher than VLED by 20%. PCOND = (IL AVG ) × D MAX × R DSON 2 where RDSON is the on-resistance of Q1 in ohms with an assumed junction temperature of +100°C, PCOND is in watts, and ILAVG is in amperes. Use the following equations to calculate the switching losses in the MOSFET: Boost configuration: ⎛ IL × VLED 2 × C GD × fSW ⎞ PSW = ⎜ AVG ⎟ 2 ⎝ ⎠ ⎛ 1 1 ⎞ ×⎜ + ⎝ IGON IGOFF ⎟⎠ Boost-buck configuration: ⎛ IL × (VLED + VINMAX ) 2 × C GD × fSW ⎞ PSW = ⎜ AVG ⎟ 2 ⎝ ⎠ ⎛ 1 1 ⎞ + ×⎜ ⎝ IGON IGOFF ⎟⎠ where IGON and IGOFF are the gate currents of the MOSFET Q1 in amperes when it is turned on and turned off, respectively, VLED and VINMAX are in volts, ILAVG is in amperes, fSW is in hertz, and CGD is the gate-to-drain MOSFET capacitance in farads. Choose a MOSFET that has a higher power rating than that calculated by the following equation when the MOSFET case temperature is at +70°C: PTOT (W) = PCOND (W) + PSW (W) Feedback Compensation The LED current control loop comprising of the switching converter, the LED current amplifier, and the error amplifier should be compensated for stable control of the LED current. The switching converter small-signal transfer function has a right half-plane (RHP) zero for both boost and boost-buck configurations as the inductor current is in continuous conduction mode. The RHP zero adds a 20dB/decade gain together with a 90° phase lag, which is difficult to compensate. The easiest way to avoid this zero is to roll off the loop gain to 0dB at a frequency less than one-fifth of the RHP zero frequency with a -20dB/decade slope. The worst-case RHP zero frequency (fZRHP) is calculated as follows: Boost configuration: fZRHP = VLED × (1 - DMAX ) 2 2π × L × ILED Boost-buck configuration: fZRHP = VLED × (1 - DMAX ) 2 2π × L × ILED × DMAX where fZRHP is in hertz, VLED is in volts, L is the inductance value of L1 in henries (H), and ILED is in amperes. The switching converter small-signal transfer function also has an output pole for both boost and boost-buck configurations. The effective output impedance that determines the output pole frequency together with the output filter capacitance is calculated as: ______________________________________________________________________________________ 15 MAX16834 lowing equation. The MOSFET must be mounted on a board as per manufacturer specifications to dissipate the heat. The RMS current rating of the switching MOSFET Q1 is calculated as follows for boost and boost-buck configurations: MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver Boost configuration: (R LED + R10) × VLED R OUT = (R LED + R10) × ILED + VLED Boost-buck configuration: R OUT = (R LED + R10) × VLED (R LED + R10) × ILED × D MAX + VLED where RLED is the dynamic impedance (rate of change of voltage with current) of the LED string at the operating current, R10 is the LED current-sense resistor in ohms, VLED is in volts, and ILED is in amperes. The output pole frequency for both boost and boostbuck configurations is calculated as follows: fP2 = 1 2π × C OUT × R OUT where fP2 is in hertz, COUT is the output filter capacitance in farads, ROUT is the effective output impedance in ohms calculated above. Compensation components R7 and C7 perform two functions. C7 introduces a low-frequency pole that introduces a -20dB/decade slope into the loop gain. R7 flattens the gain of the error amplifier for frequencies above the zero formed by R7 and C7. For compensation, this zero is placed at the output pole frequency fP2 such that it provides a -20dB/decade slope for frequencies above fP2 for the complete loop gain. The value of R7 needed to fix the total loop gain at fP2 such that the total loop gain crosses 0dB at -20dB/decade at one-fifth of the RHP zero can be calculated as follows: R7 = fZRHP × R8 5 × fP2 × (1 − D MAX ) × R10 × 9.9 × GM COMP where R7 is the compensation resistor in ohms, fZRHP and fP2 are in hertz, R8 is the switch current-sense resistor in ohms, R10 is the LED current-sense resistor in ohms, factor 9.9 is the gain of the LED current amplifier, and GMCOMP is the transconductance of the error amplifier in Siemens. The value of C7 can be calculated as: 16 C7 = 1 2π × R7 × fP2 where C7 is in farads, fP2 is in hertz, and R7 is in ohms. To minimize switching frequency noise, an additional capacitor can be added in parallel with the series combination of R7 and C7. The pole from this capacitor and R7 must be a decade higher than the loop crossover frequency. Short-Circuit Protection Boost Configuration In the boost configuration (Figure 2), if the LED string is shorted then the excess current flowing in the LED current-sense resistor will cause NDRV to stop switching. The input voltage will appear on the output capacitor, and this causes very high peak currents to flow in the LED current-sense resistor R10 because the dimming MOSFET (Q2) is on. Once the voltage across the LED current-sense resistor exceeds 300mV for more than 5µs, then the dimming MOSFET Q2 turns off and stays off for 4096 switching clock cycles. At the same time, NDRV is also off. The MAX16834 goes into the hiccup mode and recovers from hiccup once the short has been removed. The power dissipation in the dimming MOSFET (Q2) is minimized during a short across the LED string. During the same period, FLT only goes high when the dimming MOSFET is on. Boost-Buck Configuration In the case of the boost-buck configuration (Figure 3), once an LED string short occurs then the behavior is different. A short across the LED string causes a high current spike due to the external capacitors at the output. The regulation loop will cause NDRV to stop switching. This causes the voltage on HV to drop if its voltage is derived from LED+. The voltage on CLV will drop, and this drop is detected after 128 clock cycles. The dimming MOSFET and the switching MOSFET will stop switching. It stays off for 4096 clock cycles, and the cycle repeats itself. The short across the LED string will cause the MAX16834 to go into a hiccup mode. At the same time the FLT signal asserts itself for 4096 clock cycles every hiccup cycle. In the case where the HV voltage is derived from a source different than LED+, then the LED current will stay in regulation even during a short across the LED string. In this case, FLT does not assert itself during the short. ______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver MAX16834 VIN C1 L1 Q3 R1 C8 LV FLT IN NDRV D1 LED+ D2 24V C3 Q1 LEDs UVEN HV SC R4 CS C2 ON MAX16834 OFF PWMDIM R3 LED- RT/SYNC C5 Q2 DIMOUT R2 VCC C4 REF SENSE+ OVP+ R6 CLV R5 REFI COMP SGND PGND R8 R9 R10 C9 C7 C6 R7 Figure 4. Boost LED Driver with Automotive Load Dump Protection ______________________________________________________________________________________ 17 MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver VIN LED+ C1 C3 D1 L1 R1 LV HV IN NDRV Q1 VLV LEDs UVEN C2 SC R3 RT/SYNC CS ON MAX16834 OFF PWMDIM R4 LED- C5 R2 VCC C4 Q2 DIMOUT REF SENSE+ R6 OVP+ R5 REFI CLV FLT COMP SGND PGND R8 C7 R9 C8 C6 R7 VLV VLV Figure 5. High-Side Buck LED Driver 18 R10 ______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver MAX16834 VIN C1 L1 VOUT R1 LV FLT IN NDRV D1 Q1 UVEN C3 HV C2 SC R3 RT/SYNC C5 R4 CS MAX16834 VREF PWMDIM DIMOUT R2 VCC C4 REF SENSE+ OVP+ CLV R6 R5 REFI COMP SGND PGND OPTIONAL C6 R9 R10 C7 R8 R7 Figure 6. Boost DC-DC Converter ______________________________________________________________________________________ 19 MAX16834 High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver VIN C1 L1 R1 D1 LV HV IN NDRV VOUT Q1 UVEN C2 C3 SC MAX16834 R3 R4 R11 R9 R10 CS VREF RT/SYNC PWMDIM C5 R2 VCC C4 REF DIMOUT SENSE+ OVP+ R6 R5 CLV N.C. REFI COMP FLT SGND PGND C6 R7 Figure 7. Boost-Buck DC-DC Converter 20 C7 R8 ______________________________________________________________________________________ VIN High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver 1) Use a large contiguous copper plane under the MAX16834 package. Ensure that all heat-dissipating components have adequate cooling. 2) Isolate the power components and high-current path from the sensitive analog circuitry. 3) Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation. Keep switching loops short such that: a) The anode of D1 must be connected very close to the drain of the MOSFET Q1. b) The cathode of D1 must be connected very close to COUT. c) COUT and the current-sense resistor R8 must be connected directly to the ground plane. 4) Connect PGND and SGND to a star-point configuration. 5) Keep the power traces and load connections short. This practice is essential for high efficiency. Use thick copper PCBs (2oz vs.1oz) to enhance full-load efficiency. 6) Route high-speed switching nodes away from the sensitive analog areas. Use an internal PCB layer for the PGND and SGND plane as an EMI shield to keep radiated noise away from the device, feedback dividers, and analog bypass capacitors. 7) To prevent discharge of the compensation capacitors during the off-time of the dimming cycle, ensure that the PCB area close to these components has extremely low leakage. Discharge of these capacitors due to leakage results in reduced performance of the dimming circuitry. ______________________________________________________________________________________ 21 MAX16834 Layout Recommendations Typically, there are two sources of noise emission in a switching power supply: high di/dt loops and high dv/dt surfaces. For example, traces that carry the drain current often form high di/dt loops. Similarly, the heatsink of the MOSFET connected to the device drain presents a dv/dt source; therefore, minimize the surface area of the heatsink as much as is compatible with the MOSFET power dissipation or shield it. Keep all PCB traces carrying switching currents as short as possible to minimize current loops. Use ground planes for best results. Careful PCB layout is critical to achieve low switching losses and clean, stable operation. Use a multilayer board whenever possible for better noise immunity and power dissipation. Follow these guidelines for good PCB layout: High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver MAX16834 Pin Configurations VCC NDRV PGND CS TOP VIEW IN TOP VIEW 15 14 13 12 11 + LV 1 20 DIMOUT SENSE+ 2 PWMDIM OVP+ 3 9 UVEN SGND 4 8 RT/SYNC 7 FLT 6 SC 10 HV 16 CLV 17 19 CLV 18 HV MAX16834 COMP 5 DIMOUT 18 MAX16834 LV 19 3 4 5 REF REFI 2 COMP 1 SGND + OVP+ SENSE+ 20 *EP 17 IN 16 VCC REF 6 15 NDRV REFI 7 14 PGND SC 8 13 CS FLT 9 12 PWMDIM RT/SYNC 10 11 UVEN TSSOP TQFN *EP = EXPOSED PAD. Package Information Chip Information PROCESS: BiCMOS–DMOS 22 For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 20-TQFN-EP T2044-3 21-0139 20-TSSOP-EP U20E+1 21-0108 ______________________________________________________________________________________ High-Power LED Driver with Integrated High-Side LED Current Sense and PWM Dimming MOSFET Driver REVISION NUMBER REVISION DATE 0 8/08 Initial release 1 2/09 Added TSSOP package and automotive version. Also updated Electrical Characteristics, Pin Description, Detailed Description, and LED CurrentSense Input (SENSE+) section, Pin Configuration and Package Information 2 5/09 Added automotive version of TQFN package 3 1/10 Added requirement for a capacitor on the SENSE+ pin DESCRIPTION PAGES CHANGED — 1, 2, 6, 7, 8, 9, 22 1 1, 2, 7, 9, 11, 13, 17–20 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 23 © 2010 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc. MAX16834 Revision History