MAXIM MAX16818ETI+

19-0666; Rev 0; 10/06
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
The MAX16818 pulse-width modulation (PWM) LED driver controller provides high-output-current capability in
a compact package with a minimum number of external
components. The MAX16818 is suitable for use in synchronous and nonsynchronous step-down (buck)
topologies, as well as in boost, buck-boost, SEPIC, and
Cuk LED drivers. The MAX16818 is the first LED driver
controller that enables Maxim’s patent-pending technology for fast LED current transients of up to 20A/µs and
30kHz dimming frequency.
This device utilizes average-current-mode control that
enables optimal use of MOSFETs with optimal charge
and on-resistance characteristics. This results in the
minimized need for external heatsinking even when
delivering up to 30A of LED current. True differential
sensing enables accurate control of the LED current. A
wide dimming range is easily implemented to accommodate an external PWM signal. An internal regulator
enables operation over a wide input voltage range:
4.75V to 5.5V or 7V to 28V and above with a simple
external biasing device. The wide switching frequency
range, up to 1.5MHz, allows for the use of small inductors and capacitors.
The MAX16818 features a clock output with 180° phase
delay to control a second out-of-phase LED driver to
reduce input and output filter capacitors size or to minimize ripple currents. The MAX16818 offers programmable hiccup, overvoltage, and overtemperature protection.
The MAX16818ETI+ is rated for the extended temperature range (-40°C to +85°C) and the MAX16818ATI+ is
rated for the automotive temperature range (-40°C to
+125°C). This LED driver controller is available in a
lead-free, 0.8mm high, 5mm x 5mm 28-pin TQFN package with exposed paddle.
Features
o High-Current LED Driver Controller IC, Up to 30A
Output Current
o Average-Current-Mode Control
o True-Differential Remote-Sense Input
o 4.75V to 5.5V or 7V to 28V Input Voltage Range
o Programmable Switching Frequency or External
Synchronization from 125kHz to 1.5MHz
o Clock Output for 180° Out-of-Phase Operation
o Integrated 4A Gate Drivers
o Output Overvoltage and Hiccup Mode
Overcurrent Protection
o Thermal Shutdown
o Thermally Enhanced 28-Pin Thin QFN Package
o -40°C to +125°C Operating Temperature Range
Ordering Information
PART
TEMP RANGE
PINPACKAGE
PKG
CODE
MAX16818ATI+
-40°C to +125°C
28 TQFN-EP*
T2855-3
MAX16818ETI+
-40°C to +85°C
28 TQFN-EP*
T2855-3
+Denotes lead-free package.
*EP = Exposed paddle.
Simplified Diagram
7V TO 28V
C1
Applications
Front Projectors/Rear Projection TVs
IN
EN
DH
Q1
ILIM
VLED
L1
Portable and Pocket Projectors
MAX16818
Automotive, Bus/Truck Exterior Lighting
DL
LCD TVs and Display Backlight
Q2
C2
Q3
Automotive Emergency Lighting and Signage
OVI
CSP
CLP
PGND
R1
Typical Operating Circuit and Pin Configuration located at
end of data sheet.
.
HIGH-FREQUENCY
PULSE TRAIN
NOTE: MAXIM PATENT-PENDING TOPOLOGY
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX16818
General Description
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
ABSOLUTE MAXIMUM RATINGS
IN to SGND.............................................................-0.3V to +30V
BST to SGND..........................................................-0.3V to +35V
BST to LX..................................................................-0.3V to +6V
DH to LX .......................................-0.3V to [(VBST - VLX_) + 0.3V]
DL to PGND................................................-0.3V to (VDD + 0.3V)
VCC to SGND............................................................-0.3V to +6V
VCC, VDD to PGND ...................................................-0.3V to +6V
SGND to PGND .....................................................-0.3V to +0.3V
All Other Pins to SGND...............................-0.3V to (VCC + 0.3V)
Continuous Power Dissipation (TA = +70°C)
28-Pin TQFN (derate 34.5mW/°C above +70°C) .......2758mW
Operating Temperature Range
MAX16818ATI+..............................................-40°C to +125°C
MAX16818ETI+................................................-40°C to +85°C
Maximum Junction Temperature .....................................+150°C
Storage Temperature Range .............................-60°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VCC = 5V, VDD = VCC, TA = TJ = TMIN to TMAX, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
SYSTEM SPECIFICATIONS
Input Voltage Range
VIN
Short IN and VCC together for 5V input
operation
Quiescent Supply Current
IQ
EN = VCC or SGND, not switching
7
28
4.75
5.50
2.7
5.5
V
mA
LED CURRENT REGULATOR
SENSE+ to SENSE- Accuracy
(Note 2)
Soft-Start Time
No load, VIN = 4.75V to 5.5V, fSW = 500kHz
0.594
0.6
0.606
No load, VIN = 7V to 28V, fSW = 500kHz
0.594
0.6
0.606
tSS
V
Clock
Cycles
1024
STARTUP/INTERNAL REGULATOR
VCC Undervoltage Lockout
UVLO
VCC rising
4.1
VCC Undervoltage Hysteresis
4.3
4.5
200
VCC Output Voltage
VIN = 7V to 28V, ISOURCE = 0 to 60mA
4.85
V
mV
5.1
5.30
V
1.1
3.0
Ω
MOSFET DRIVERS
Output Driver Impedance
Output Driver Source/Sink Current
Nonoverlap Time
RON
Low or high output, ISOURCE/SINK = 20mA
IDH,IDL
tNO
CDH/DL = 5nF
4
A
35
ns
OSCILLATOR
Switching Frequency Range
125
Switching Frequency
Switching Frequency
Switching Frequency
Switching Frequency Accuracy
2
fSW
1500
RT = 500kΩ
121
125
129
RT = 120kΩ
495
521
547
RT = 39.9kΩ
1515
1620
1725
120kΩ ≤ RT ≤ 500kΩ
-5
+5
40kΩ ≤ RT ≤ 120kΩ
-8
+8
_______________________________________________________________________________________
kHz
kHz
%
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
(VCC = 5V, VDD = VCC, TA = TJ = TMIN to TMAX, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
φ_CLKOUT
With respect to DH, fSW = 125kHz
CLKOUT Output Low Level
VCLKOUTL
ISINK = 2mA
CLKOUT Output High Level
VCLKOUTH
ISOURCE = 2mA
CLKOUT Phase Shift
SYNC Input-High Pulse Width
MIN
0.4
ns
200
2.0
SYNC Input Clock Low Threshold
VSYNCL
ISYNC_OUT
V
V
tSYNC
VSYNC_OFF
UNITS
Degrees
4.5
VSYNCH
SYNC Power-Off Level
MAX
180
SYNC Input Clock High Threshold
SYNC Pullup Current
TYP
VRT/SYNC = 0V
V
250
0.4
V
750
µA
0.4
V
INDUCTOR CURRENT LIMIT
Average Current-Limit Threshold
VCL
CSP to CSN
24.0
26.9
28.2
mV
Reverse Current-Limit Threshold
VCLR
CSP to CSN
-3.2
-2.3
-0.1
mV
Cycle-by-Cycle Current Limit
Cycle-by-Cycle Overload
CSP to CSN
VCSP to VCSN = 75mV
Hiccup Divider Ratio
LIM to VCM, no switching
60
260
0.547
Hiccup Reset Delay
LIM Input Impedance
mV
LIM to SGND
0.558
ns
0.565
V/V
200
ms
55.9
kΩ
4
kΩ
CURRENT-SENSE AMPLIFIER
CSP or CSN Input Resistance
Common-Mode Range
Input Offset Voltage
RCS
VCMR(CS)
VIN = 7V to 28V
0
5.5
V
VOS(CS)
0.1
mV
Amplifier Gain
AV(CS)
34.5
V/V
3dB Bandwidth
f3dB
4
MHz
550
µS
50
dB
CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance
gm
Open-Loop Gain
AVOL(CE)
No load
DIFFERENTIAL VOLTAGE AMPLIFIER FOR LED CURRENT (DIFF)
Common-Mode Voltage Range
VCMR(DIFF)
0
DIFF Output Voltage
VCM
Input Offset Voltage
VOS(DIFF)
-1
AV(DIFF)
0.994
Amplifier Gain
3dB Bandwidth
f3dB
Minimum Output-Current Drive
SENSE+ to SENSE- Input
VSENSE+ = VSENSE- = 0V
RVS
1
3
4
VSENSE- = 0V
V
+1
mV
1.006
V/V
0.6
CDIFF = 20pF
IOUT(DIFF)
+1.0
50
V
MHz
mA
100
kΩ
4
MHz
1
MHz
V_IOUT AMPLIFIER
Gain-Bandwidth Product
VV_IOUT = 2.0V
3dB Bandwidth
VV_IOUT = 2.0V
Output Sink Current
30
µA
Output Source Current
90
µA
_______________________________________________________________________________________
3
MAX16818
ELECTRICAL CHARACTERISTICS (continued)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
ELECTRICAL CHARACTERISTICS (continued)
(VCC = 5V, VDD = VCC, TA = TJ = TMIN to TMAX, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
Maximum Load Capacitance
TYP
MAX
50
V_IOUT Output to IOUT Transfer
Function
RSENSE = 1mΩ, 100mV ≤ V_IOUT ≤ 5.5V
132.3
Offset Voltage
135
UNITS
pF
137.7
1
mV/A
mV
VOLTAGE-ERROR AMPLIFIER (EAOUT)
Open-Loop Gain
AVOLEA
70
dB
Unity-Gain Bandwidth
fGBW
3
MHz
EAN Input Bias Current
IB(EA)
Error Amplifier Output Clamping
Voltage
VEAN = 2.0V
VCLAMP(EA) With respect to VCM
-0.2
+0.03
+0.2
µA
883
930
976
mV
87.5
90
92.5
%VOUT
POWER-GOOD AND OVERVOLTAGE PROTECTION
PGOOD Trip Level
PGOOD Output Low Level
PGOOD Output Leakage Current
OVI Trip Threshold
OVI Input Bias Current
VUV
VPGLO
IPG
OVPTH
PGOOD goes low when VOUT is below this
threshold
ISINK = 4mA
PGOOD = VCC
With respect to SGND
1.244
IOVI
1.276
0.4
V
1
µA
1.308
0.2
V
µA
ENABLE INPUT
EN Input High Voltage
VEN
EN rising
2.437
EN Input Hysteresis
EN Pullup Current
2.5
2.562
V
16.5
µA
0.28
IEN
13.5
15
V
THERMAL SHUTDOWN
Thermal Shutdown
Thermal Shutdown Hysteresis
Temperature rising
150
°C
30
°C
Note 1: Specifications at TA = +25° are 100% tested. Specifications over the temperature range are guaranteed by design.
Note 2: Does not include an error due to finite error amplifier gain. See the Voltage-Error Amplifier (EAOUT) section.
4
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
VIN = 12V
20
VIN = 5V
300
500
700
900
28.0
27.5
27.0
26.5
VIN = 12V
fSW = 250kHz
26.0
-40
1100 1300 1500
MAX16818 toc03
MAX16818 toc02
VIN = 12V
fSW = 250kHz
CDL/CDH = 22nF
-15
10
35
85
60
0
1
2
3
4
TEMPERATURE (°C)
VOUT (V)
HICCUP CURRENT LIMIT vs. REXT
VCC LOAD REGULATION
vs. INPUT VOLTAGE
DRIVER RISE TIME
vs. DRIVER LOAD CAPACITANCE
5.25
MAX16818 toc04
26.0
25.5
100
5.15
5
MAX16818 toc06
FREQUENCY (kHz)
MAX16818 toc05
100
VIN = 12V
fSW = 250kHz
80
VIN = 24V
24.5
5.05
VIN = 12V
tR (ns)
25.0
VCC (V)
CURRENT LIMIT (A)
64
60
0
60
DL
40
4.95
DH
VIN = 5V
24.0
VIN = 12V
fSW = 250kHz
R1 = 1mΩ
VOUT = 1.5V
23.5
20
4.85
23.0
0
4.75
0
4
8
12
16
20
0
25
50
75
100
125
1
150
6
11
16
21
REXT (MΩ)
VCC LOAD CURRENT (mA)
CAPACITANCE (nF)
DRIVER FALL TIME
vs. DRIVER LOAD CAPACITANCE
HIGH-SIDE DRIVER (DH) SINK
AND SOURCE CURRENT
LOW-SIDE DRIVER (DL) SINK
AND SOURCE CURRENT
MAX16818 toc09
MAX16818 toc07
MAX16818 toc08
100
VIN = 12V
fSW = 250kHz
80
tF (ns)
28.5
66
62
10
29.0
(VCSP - VCSN) (mV)
VIN = 24V
40
30
68
SUPPLY CURRENT (mA)
EXTERNAL CLOCK
NO DRIVER LOAD
50
SUPPLY CURRENT (mA)
70
MAX16818 toc01
60
CURRENT-SENSE THRESHOLD
vs. OUTPUT VOLTAGE
SUPPLY CURRENT vs. TEMPERATURE
SUPPLY CURRENT (IQ) vs. FREQUENCY
CLOAD = 22nF
VIN = 12V
CLOAD = 22nF
VIN = 12V
60
3A/div
2A/div
DL
40
DH
20
0
1
6
11
16
21
100ns/div
100ns/div
CAPACITANCE (nF)
_______________________________________________________________________________________
5
MAX16818
Typical Operating Characteristics
(TA = +25°C, using Figure 5, unless otherwise noted.)
Typical Operating Characteristics (continued)
(TA = +25°C, using Figure 5, unless otherwise noted.)
HIGH-SIDE DRIVER (DH) FALL TIME
HIGH-SIDE DRIVER (DH) RISE TIME
LOW-SIDE DRIVER (DL) RISE TIME
MAX16818 toc11
MAX16818 toc10
MAX16818 toc12
CLOAD = 22nF
VIN = 12V
CLOAD = 22nF
VIN = 12V
CLOAD = 22nF
VIN = 12V
2V/div
2V/div
2V/div
40ns/div
40ns/div
40ns/div
MAX16818 toc13
10,000
2V/div
fSW (kHz)
CLOAD = 22nF
VIN = 12V
MAX16818 toc14
FREQUENCY vs. RT
LOW-SIDE DRIVER (DL) FALL TIME
VIN = 12V
1000
100
110
30
40ns/div
70
190
150
270
230
350
310
430
390
510
470
RT (kΩ)
FREQUENCY vs. TEMPERATURE
VIN = 12V
258
256
SYNC, CLKOUT, AND LX WAVEFORM
MAX16818 toc16
MAX16818 toc15
260
SYNC
5V/div
254
fSW (kHz)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
252
CLKOUT
5V/div
250
VIN = 12V
fSW = 250kHz
248
246
LX
10V/div
244
242
240
-40
-15
10
35
60
85
1μs/div
TEMPERATURE (°C)
6
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
PIN
NAME
1
PGND
FUNCTION
2, 7
N.C.
3
DL
Low-Side Gate Driver Output
4
BST
Boost Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver
supply. Connect a ceramic capacitor between BST and LX.
Power-Supply Ground
No Connection. Not internally connected.
5
LX
Source connection for the high-side MOSFET.
6
DH
High-Side Gate Driver Output. Drives the gate of the high-side MOSFET.
8, 22, 25
SGND
Signal Ground. Ground connection for the internal control circuitry. Connect SGND and PGND
together at one point near the IC.
9
CLKOUT
Oscillator Output. Rising edge of CLKOUT is phase-shifted from the rising edge of DH by 180°.
10
PGOOD
Power-Good Output
11
EN
12
RT/SYNC
Switching Frequency Programming and Chip-Enable Input. Connect a resistor from RT/SYNC to
SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency
with external clock.
13
V_IOUT
Voltage Source Output Proportional to the Inductor Current. The voltage at V_IOUT = 135 x ILED x RS.
14
LIM
Current-Limit Setting Input. Connect a resistor from LIM to SGND to set the hiccup current-limit
threshold. Connect a capacitor from LIM to SGND to ignore short output overcurrent pulses.
15
OVI
Overvoltage Protection. Connect OVI to DIFF. When OVI exceeds 12.7% above the programmed
output voltage, DH is latched low and DL is latched high. Toggle EN or recycle the input power to
reset the latch.
16
CLP
Current-Error Amplifier Output. Compensate the current loop by connecting an RC network to ground.
17
EAOUT
18
EAN
Voltage-Error Amplifier Inverting Input
19
DIFF
Differential Remote-Sense Amplifier Output. DIFF is the output of a precision unity-gain amplifier
whose inputs are SENSE+ and SENSE-.
20
CSN
Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
Output Enable. Drive high or leave unconnected for normal operation. Drive low to shut down the
power drivers. EN has an internal 15µA pullup current. Connect a capacitor from EN to SGND to
program the hiccup-mode duty cycle.
Voltage-Error Amplifier Output. Connect to the external compensation network.
_______________________________________________________________________________________
7
MAX16818
Pin Description
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX16818
Pin Description (continued)
8
PIN
NAME
FUNCTION
21
CSP
Current-Sense Differential Amplifier Positive Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
23
SENSE-
Differential LED Current-Sensing Negative Input. SENSE- is used to sense the LED current. Connect
SENSE- to the negative side of the LED current-sense resistor.
24
SENSE+
Differential LED Current-Sensing Positive Input. SENSE+ is used to sense the LED current. Connect
SENSE+ to the positive side of the LED current-sense resistor.
26
IN
27
VCC
Internal +5V Regulator Output. VCC is derived from the IN voltage. Bypass VCC to SGND with 4.7µF
and 0.1µF ceramic capacitors.
28
VDD
Supply Voltage for Low-Side and High-Side Drivers. Connect a parallel combination of 0.1µF and 1µF
ceramic capacitors to PGND and a 1Ω resistor to VCC to filter out the high peak currents of the driver
from internal circuitry.
—
EP
Supply Voltage Connection. Connect IN to VCC for a +5V system.
Exposed Paddle. Connect the exposed paddle to a copper pad (SGND) to improve power
dissipation.
_______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
ON/OFF
R6
C3
R3
VLED
R4
VIN
7V TO 28V
VCC
R5
C2
14
LIM
C10
C9
13
12
V_IOUT RT/SYNC
9
10
PGOOD CLKOUT
11
EN
L1
8
SGND
15 OVI
N.C. 7
16 CLP
DH 6
17 EAOUT
LX 5
VLED
D1
Q1
R12
R11
LED
STRING
C1
C8
R7
C7
18 EAN
BST 4
MAX16818
R10
19 DIFF
DL 3
20 CSN
N.C. 2
R1
PGND 1
21 CSP
SGND
22
R2
SENSE23
SENSE+
24
SGND
25
IN
26
VDD
28
VCC
27
VCC
VIN
C6
R8
C5
C4
Figure 1. Typical Application Circuit for a Boost LED Driver (Nonsynchronous)
_______________________________________________________________________________________
9
MAX16818
Typical Application Circuits
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX16818
Typical Application Circuits (continued)
ON/OFF
R6
C3
R3
VLED
R4
VIN
7V TO 28V
VCC
R5
LED
STRING
1 TO 6
LEDS
R2
C2
14
LIM
C10
C9
13
12
V_IOUT RT/SYNC
9
10
11
EN
L1
8
SGND
PGOOD CLKOUT
15 OVI
N.C. 7
16 CLP
DH 6
17 EAOUT
LX 5
D1
VLED
Q1
R12
VCC
R11
RS+ VCC
C8
R7
C7
18 EAN
MAX4073T
RS-
BST 4
MAX16818
OUT
C1
R10
19 DIFF
DL 3
20 CSN
N.C. 2
PGND 1
21 CSP
SGND
22
R1
SENSE23
SENSE+
24
SGND
25
IN
26
VCC
27
VDD
28
VCC
VIN
C6
R8
C5
C4
Figure 2. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (Input: 7V to 28V, Output: 1 to 6 LEDs in Series)
10
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
ON/OFF
R6
C4
R3
VLED
R4
VIN
7V TO 28V
VCC
R5
C3
14
LIM
C10
13
12
9
10
11
EN
8
15 OVI
SGND
N.C. 7
16 CLP
DH 6
17 EAOUT
LX 5
C11
V_IOUT RT/SYNC
L1
PGOOD CLKOUT
C1
VLED
D1
Q1
R12
R11
L2
C9
C8
18 EAN
C2
LED
STRING
BST 4
MAX16818
R10
19 DIFF
DL 3
20 CSN
N.C. 2
R1
PGND 1
21 CSP
SGND
22
R2
R7
SENSE23
SENSE+
24
SGND
25
IN
26
VDD
28
VCC
27
VCC
VIN
C7
R8
C6
C5
Figure 3. Typical Application Circuit for a SEPIC LED Driver
______________________________________________________________________________________
11
MAX16818
Typical Application Circuits (continued)
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX16818
Typical Application Circuits (continued)
ON/OFF
C3
R3
VLED
R4
VCC
R5
14
13
12
9
10
15 OVI
16 CLP
DH 6
V_IOUT RT/SYNC
11
EN
VIN
7V TO 18V
8
SGND
N.C. 7
LIM
C11
C10
R6
PGOOD CLKOUT
C2
Q1
R12
R11
LX 5
C9
C4
R7
C8
VLED
L1
17 EAOUT
18 EAN
Q3
BST 4
MAX16818
Q2
R10
19 DIFF
DL 3
20 CSN
N.C. 2
C1
LED
STRING
D2
R2
R1
PGND 1
21 CSP
SGND
22
SENSE23
SENSE+
24
SGND
25
IN
26
VDD
28
VCC
27
VCC
VIN
C7
R8
C6
C5
Figure 4. Application Circuit for a Ground-Referred Buck-Boost LED Driver
12
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
VCC
R4
14
LIM
C11
C10
R3
C3
13
12
VIN
7V TO 28V
ON/OFF
V_IOUT RT/SYNC
9
10
PGOOD CLKOUT
11
EN
8
SGND
15 OVI
N.C. 7
16 CLP
DH 6
17 EAOUT
LX 5
C2
R10
R9
L1
C9
C4
R5
C8
18 EAN
BST 4
MAX16818
Q1
D1
R8
19 DIFF
DL 3
20 CSN
N.C. 2
C1
LED
STRING
R2
R1
PGND 1
21 CSP
SGND
22
SENSE23
SENSE+
24
SGND
25
IN
26
VDD
28
VCC
27
VCC
VIN
C7
R6
C6
C5
Figure 5. Application Circuit for a Buck LED Driver
______________________________________________________________________________________
13
MAX16818
Typical Application Circuits (continued)
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
MAX16818
Functional Diagram
VCC
IS
EN
0.5V x VCC
5V
LDO
REGULATOR
IN
VCC
UVLO
POR
TEMP SENSOR
TO INTERNAL
CIRCUITS
LIM
HICCUP MODE
CURRENT LIMIT
126.7kΩ
MAX16818
VCM
100kΩ
0.5 x VCLAMP
CLP
CA
CSN
Q
R
Q
RT
Ct
AV = 34.5
CSP
S
VCM
gm = 500μS
VDD
PWM
COMPARATOR
CEA
AV = 4
BST
V_IOUT
VCLAMP
HIGH
VCLAMP
LOW
SGND
RAMP
CPWM
S
Q
DH
LX
2 x fS (V/s)
CLK
RT/SYNC
OSCILLATOR
R
Q
DL
CLKOUT
RAMP
GENERATOR
DIFF
+0.6V
SENSESENSE+
PGND
PGOOD
N
DIFF
AMP
0.1 x VREF
EAOUT
ERROR AMP
EAN
0.12 x VREF
OVP LATCH
VEA
LATCH
SOFTSTART
OVP COMP
VREF = 0.6V
VCM (0.6V)
CLEAR ON UVLO RESET OR
ENABLE LOW
OVI
Figure 6. MAX16818 Functional Diagram
14
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
The MAX16818 is a high-performance average-currentmode PWM controller for high-power, high-brightness
LEDs (HBLEDs). Average current-mode control is the
ideal method for driving HBLEDs. This technique offers
inherently stable operation, reduces component derating and size by accurately controlling the inductor current. The device achieves high efficiency at high
current (up to 30A) with a minimum number of external
components. The high- and low-side drivers source
and sink up to 4A for lower switching losses while driving high-gate-charge MOSFETs. The MAX16818’s
CLKOUT output is 180° out-of-phase with respect to the
high-side driver. CLKOUT drives a second MAX16818
LED driver out of phase, reducing the input-capacitor
ripple current.
The MAX16818 consists of an inner average current loop
representing inductor current and an outer voltage loop
voltage-error amplifier (VEA) that directly controls LED
current. The combined action of the two loops results in
a tightly regulated LED current. The inductor current is
sensed across a current-sense resistor. The differential
amplifier senses LED current through a sense resistor in
series with the LEDs and the resulting sensed voltage is
compared against an internal 0.6V reference at the erroramplifier input. The MAX16818 will adjust the LED current to within 1% accuracy to maintain emitted spectrum
of the light in HBLEDs.
IN, VCC, and VDD
The MAX16818 accepts either a 4.75V to 5.5V or 7V to
28V input voltage range. All internal control circuitry
operates from an internally regulated nominal voltage of
5V (VCC). For input voltages of 7V or greater, the internal VCC regulator steps the voltage down to 5V. The
VCC output voltage is a regulated 5V output capable of
sourcing up to 60mA. Bypass the VCC to SGND with
4.7µF and 0.1µF low-ESR ceramic capacitors for highfrequency noise rejection and stable operation.
The MAX16818 uses VDD to power the low-side and
high-side drivers. Isolate VDD from VCC with a 1Ω resistor and put a 1µF capacitor in parallel with a 0.1µF
capacitor to ground to prevent high-current noise spikes
created by the driver from disrupting internal circuitry.
The TQFN is a thermally enhanced package and can
dissipate up to 2.7W. The high-power packages allow
the high-frequency, high-current converter to operate
from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
total VCC regulator output current (ICC). ICC includes quiescent current (IQ) and gate-drive current (IDD):
PD = VIN x ICC
ICC = IQ + [fSW x (QG1 + QG2)]
where QG1 and QG2 are the total gate charge of the
low-side and high-side external MOSFETs at VGATE =
5V, IQ is 3.5mA (typ), and fSW is the switching frequency of the converter.
Undervoltage Lockout (UVLO)
The MAX16818 includes an undervoltage lockout with
hysteresis and a power-on-reset circuit for converter
turn-on. The UVLO rising threshold is internally set at
4.35V with a 200mV hysteresis. Hysteresis at UVLO
eliminates chattering during startup.
Most of the internal circuitry, including the oscillator,
turns on when the input voltage reaches 4V. The
MAX16818 draws up to 3.5mA of current before the
input voltage reaches the UVLO threshold.
Soft-Start
The MAX16818 has an internal digital soft-start for a
monotonic, glitch-free rise of the output current. Softstart is achieved by the controlled rise of the error
amplifier dominant input in steps using a 5-bit counter
and a 5-bit DAC. The soft-start DAC generates a linear
ramp from 0 to 0.7V. This voltage is applied to the error
amplifier at a third (noninverting) input. As long as the
soft-start voltage is lower than the reference voltage,
the system converges to that lower reference value.
Once the soft-start DAC output reaches 0.6V, the reference takes over and the DAC output continues to climb
to 0.7V, assuring that it does not interfere with the reference voltage.
Internal Oscillator
The internal oscillator generates a clock with the frequency proportional to the inverse of RT. The oscillator
frequency is adjustable from 125kHz to 1.5MHz with
better than 8% accuracy using a single resistor connected from RT/SYNC to SGND. The frequency accuracy avoids the over-design, size, and cost of passive
filter components like inductors and capacitors. Use
the following equation to calculate the oscillator frequency:
For 120kΩ ≤ RT ≤ 500kΩ:
6.25 x 1010
fSW
RT =
For 40kΩ ≤ RT ≤ 120kΩ:
RT =
6.40 x 1010
fSW
______________________________________________________________________________________
15
MAX16818
Detailed Description
The oscillator also generates a 2VP-P voltage-ramp signal for the PWM comparator and a 180° out-of-phase
clock signal for CLKOUT to drive a second LED regulator out-of-phase.
Synchronization
The MAX16818 can be easily synchronized by connecting an external clock to RT/SYNC. If an external
clock is present, then the internal oscillator is disabled
and the external clock is used to run the device. If the
external clock is removed, the absence of clock for
32µs is detected and the circuit starts switching from
the internal oscillator. Pulling RT/SYNC to ground for at
least 50µs disables the converter. Use an open-collector transistor to synchronize the MAX16818 with the
external system clock.
Control Loop
The MAX16818 uses an average-current-mode control
scheme to regulate the output current (Figure 7). The
main control loop consists of an inner current loop for
controlling the inductor current and an outer current
loop for regulating the LED current. The inner current
loop absorbs the inductor pole reducing the order of the
outer current loop to that of a single-pole system. The
current loop consists of a current-sense resistor (RS), a
current-sense amplifier (CA), a current-error amplifier
(CEA), an oscillator providing the carrier ramp, and a
PWM comparator (CPWM) (Figure 7). The precision CA
amplifies the sense voltage across RS by a factor of
34.5. The inverting input to the CEA senses the CA output. The CEA output is the difference between the voltage-error amplifier output (EAOUT) and the amplified
voltage from the CA. The RC compensation network
connected to CLP provides external frequency compensation for the CEA. The start of every clock cycle
enables the high-side drivers and initiates a PWM oncycle. Comparator CPWM compares the output voltage
from the CEA with a 0V to 2V ramp from the oscillator.
The PWM on-cycle terminates when the ramp voltage
exceeds the error voltage. Compensation for the outer
LED current loop varies based upon the topology.
The MAX16818 outer LED current control loop consists
of the differential amplifier (DIFF AMP), reference voltage, and VEA. The unity-gain differential amplifier provides true differential remote sensing of the voltage
across the LED current set resistor, RLS. The differential
amplifier output connects to the inverting input (EAN) of
the VEA. The DIFF AMP is bypassed and the inverting
input is available to the pin for direct feedback. The
noninverting input of the VEA is internally connected to
an internal precision reference voltage, set to 0.6V. The
VEA controls the inner current loop (Figure 6). A feedback network compensates the outer loop using the
EAOUT and EAIN pins.
CCF
RCF
CCFF
CSN
CSP
CA
EAOUT
SENSE+
Z COMP
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
CLP
VIN
MAX16818
IL
600mV
DIFF
AMP
SENSEEAN
CEA
VEA
VREF + VCM = 1.2V
CPWM
LED
STRING
DRIVE
COUT
RLS
DIFF
RS
Figure 7. MAX16818 Control Loop
16
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Inductor Peak-Current Comparator
The peak-current comparator provides a path for fast
cycle-by-cycle current limit during extreme fault conditions, such as an inductor malfunction (Figure 8). Note
the average current-limit threshold of 26.9mV still limits
the output current during short-circuit conditions. To
prevent inductor saturation, select an inductor with a
saturation current specification greater than the average
current limit. Proper inductor selection ensures that only
the extreme conditions trip the peak-current comparator, such as an inductor with a shorted turn. The 60mV
threshold for triggering the peak-current limit is twice the
full-scale average current-limit voltage threshold. The
peak-current comparator has only a 260ns delay.
Current-Error Amplifier
(For Inductor Currents)
The MAX16818 has a transconductance current-error
amplifier (CEA) with a typical gm of 550µS and 320µA
output sink- and source-current capability. The currenterror amplifier output CLP serves as the inverting input
to the PWM comparator. CLP is externally accessible to
provide frequency compensation for the inner current
loops (Figure 7). Compensate (CEA) so the inductor
current negative slope, which becomes the positive
slope to the inverting input of the PWM comparator, is
less than the slope of the internally generated voltage
ramp (see the Compensation section).
PWM Comparator and R-S Flip-Flop
The PWM comparator (CPWM) sets the duty cycle for
each cycle by comparing the output of the current-error
amplifier to a 2VP-P ramp. At the start of each clock
cycle, an R-S flip-flop resets and the high-side driver
(DH) goes high. The comparator sets the flip-flop as
soon as the ramp voltage exceeds the CLP voltage,
thus terminating the on-cycle (Figure 8).
VDD
PEAK-CURRENT
COMPARATOR
60mV
CLP
AV = 34.5
CSP
CA
CSN
MAX16818
gm = 550μS
BST
CEA
SET
EAN
Q
S
VEA
DH
CPWM
RAMP
EAOUT
2 x fS (V/s)
LX
CLK
R
CLR
Q
DL
SHDN
PGND
Figure 8. MAX16818 Phase Circuit
______________________________________________________________________________________
17
MAX16818
Inductor Current-Sense Amplifier
The differential current-sense amplifier (CA) provides a
DC gain of 34.5. The maximum input offset voltage of
the current-sense amplifier is 1mV and the commonmode voltage range is 0 to 5.5V (IN = 7V to 28V). The
current-sense amplifier senses the voltage across a
current-sense resistor. The maximum common-mode
voltage is 3.6V when VIN = 5V.
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Differential Amplifier
BST
The DIFF AMP facilitates remote sensing at the load
(Figure 7). It provides true differential LED current
(through the RLS sense resistor) sensing while rejecting
the common-mode voltage errors due to high-current
ground paths. The VEA provides the difference
between the differential amplifier output (DIFF) and the
desired LED current-sense voltage. The differential
amplifier has a bandwidth of 3MHz. The difference
between SENSE+ and SENSE- is regulated to 0.6V.
Connect SENSE+ to the positive side of the LED currentsense resistor and SENSE- to the negative side of the
LED current-sense resistor (which is often PGND).
The MAX16818 uses VDD to power the low- and highside MOSFET drivers. The high-side driver derives its
power through a bootstrap capacitor and VDD supplies
power internally to the low-side driver. Connect a
0.47µF low-ESR ceramic capacitor between BST and
LX. Connect a Schottky rectifier from BST to VDD. Keep
the loop formed by the boost capacitor, rectifier, and IC
small on the PCB.
MOSFET Gate Drivers (DH, DL)
The high-side (DH) and low-side (DL) drivers drive the
gates of external n-channel MOSFETs (Figures 1–5).
The drivers’ 4A peak sink- and source-current capability provides ample drive for the fast rise and fall times of
the switching MOSFETs. Faster rise and fall times result
in reduced cross-conduction losses. Due to physical
realities, extremely low gate charges and R DS(ON)
resistance of MOSFETs are typically exclusive of each
other. MOSFETs with very low RDS(ON) will have a higher gate charge and vice versa. Choosing the high-side
MOSFET (Q1) becomes a trade-off between these two
attributes. Applications where the input voltage is much
higher than the output voltage result in a low duty cycle
where conduction losses are less important than
switching losses. In this case, choose a MOSFET with
very low gate charge and a moderate R DS(ON).
Conversely, for applications where the output voltage is
near the input voltage resulting in duty cycles much
greater than 50%, the RDS(ON) losses become at least
equal, or even more important than the switching losses.
In this case, choose a MOSFET with very low RDS(ON)
and moderate gate charge. Finally, for the applications
where the duty cycle is near 50%, the two loss components are nearly equal, and a balanced MOSFET with
moderate gate charge and RDS(ON) work best.
In a buck topology, the low-side MOSFET (Q2) typically
operates in a zero voltage switching mode, thus it does
not have switching losses. Choose a MOSFET with very
low RDS(ON) and moderate gate charge.
Size both the high-side and low-side MOSFETs to handle the peak and RMS currents during overload conditions. The driver block also includes a logic circuit that
provides an adaptive nonoverlap time to prevent shootthrough currents during transition. The typical nonoverlap time between the high-side and low-side MOSFETs
is 35ns.
18
Protection
The MAX16818 includes output overvoltage protection
(OVP). During fault conditions when the load goes to
high impedance (opens), the controller attempts to
maintain LED current. The OVP protection disables the
MAX16818 whenever the voltage exceeds the threshold, protecting the external circuits from undesirable
voltages.
Current Limit
The VEA output is clamped to 930mV with respect to
the common-mode voltage (V CM). Average-currentmode control has the ability to limit the average current
sourced by the converter during a fault condition. When
a fault condition occurs, the VEA output clamps to
930mV with respect to the common-mode voltage
(0.6V) to limit the maximum current sourced by the converter to ILIMIT = 26.9mV / RS. The hiccup current limit
overrides the average current limit. The MAX16818
includes hiccup current-limit protection to reduce the
power dissipation during a fault condition. The hiccup
current-limit circuit derives inductor current information
from the output of the current amplifier. This signal is
compared against one half of V CLAMP(EA) . With no
resistor connected from the LIM pin to ground, the hiccup current limit is set at 90% of the full-load average
current limit. Use REXT to increase the hiccup current
limit from 90% to 100% of the full load average limit.
The hiccup current limit can be disabled by connecting
LIM to SGND. In this case, the circuit follows the average current-limit action during overload conditions.
Overvoltage Protection
The OVP comparator compares the OVI input to the
overvoltage threshold. A detected overvoltage event
latches the comparator output forcing the power stage
into the OVP state. In the OVP state, the high-side
MOSFET turns off and the low-side MOSFET latches on.
Connect OVI to the center tap of a resistor-divider from
VLED to SGND. In this case, the center tap is compared
against 1.276V. Add an RC delay to reduce the sensitivity
of the overvoltage circuit and avoid nuisance tripping of
the converter. Disable the overvoltage function by connecting OVI to SGND.
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Application Circuit Descriptions
This section provides some detail regarding the application circuits in the Simplified Diagram and Figures
1–5. The discussion includes some description of the
topology as well as basic attributes.
High-Frequency LED Current Pulser
The Simplified Diagram shows the MAX16818 providing
high-frequency, high-current pulses to the LEDs. The
basic topology must be a buck, since the inductor
always connects to the load in that configuration (in all
other topologies, the inductor disconnects from the
load at one time or another). The design minimizes the
current ripple by oversizing the inductor, which allows
for a very small (0.01µF) output capacitor. When MOSFET Q3 turns on, it diverts the current around the LEDs
at a very fast rate. Q3 also discharges the output
capacitor, but since the capacitor is so small, it does
not stress the MOSFET. Resistor R1 senses the LED/Q3
current and there is no reaction to the short that Q3
places across the LEDs. This design is superior in that
it does not attempt to actually change the inductor current at high frequencies and yet the current in the LEDs
varies from zero to full in very small periods of time. The
efficiency of this technique is very high. Q3 must be
able to dissipate the LED current applied to its RDS(ON)
at some maximum duty cycle. If the circuit needs to
control extremely high currents, use paralleled
MOSFETs. PGOOD is low during LED pulsed-current
operation.
Boost LED Driver
In Figure 1, the external components configure the
MAX16818 as a boost converter. The circuit applies the
input voltage to the inductor during the on-time, and
then during the off-time the inductor, which is in series
with the input capacitor, charges the output capacitor.
Because of the series connection between the input
voltage and the inductor, the output voltage can never
go lower than the input voltage. The design is nonsynchronous, and since the current-sense resistor connects to ground, the power supply can go to any output
voltage (above the input) as long as the components are
rated appropriately. R2 again provides the sense voltage
the MAX16818 uses to regulate the LED current.
Input-Referenced LED Driver
The circuit in Figure 2 shows a step-up/step-down regulator. It is similar to the boost converter in Figure 1 in
that the inductor is connected to the input and the
MOSFET is essentially connected to ground. However,
rather than going from the output to ground, the LEDs
span from the output to the input. This effectively
removes the boost-only restriction of the regulator in
Figure 1, allowing the voltage across the LEDs to be
greater than or less than the input voltage. LED current
sensing is not ground-referenced, so a high-side current-sense amplifier is used to measure current.
SEPIC LED Driver
Figure 3 shows the MAX16818 configured as a SEPIC
LED driver. While buck topologies require the output to
be lesser than the input, and boost topologies require
the output to be greater than the input, a SEPIC topology allows the output voltage to be greater than, equal
to, or less than the input. In a SEPIC topology, the voltage across C1 is the same as the input voltage, and L1
and L2 are the same inductance. Therefore, when Q1
conducts (on-time), both inductors ramp up current at
the same rate. The output capacitor supports the output voltage during this time. During the off-time, L1 current recharges C1 and combines with L2 to provide
current to recharge C2 and supply the load current.
Since the voltage waveform across L1 and L2 are
exactly the same, it is possible to wind both inductors
on the same core (a coupled inductor). Although voltages on L1 and L2 are the same, RMS currents can be
quite different so the windings may have a different
gauge wire. Because of the dual inductors and segmented energy transfer, the efficiency of a SEPIC converter is somewhat lower than standard bucks or
boosts. As in the boost driver, the current-sense resistor connects to ground, allowing the output voltage of
the LED driver to exceed the rated maximum voltage of
the MAX16818.
Ground-Referenced Buck/Boost LED Driver
Figure 4 depicts a buck/boost topology. During the ontime with this circuit, the current flows from the input
capacitor, through Q1, L1, and Q3 and back to the
input capacitor. During the off-time, current flows up
through Q2, L1, D1, and to the output capacitor C1.
This topology resembles a boost in that the inductor
sits between the input and ground during the on-time.
However, during the off-time the inductor resides
between ground and the output capacitor (instead of
between the input and output capacitors in boost
topologies), so the output voltage can be any voltage
less than, equal to, or greater than the input voltage. As
compared to the SEPIC topology, the buck/boost does
not require two inductors or a series capacitor, but it
does require two additional MOSFETs.
Buck Driver with Synchronous Rectification
In Figure 5, the input voltage can go from 7V to 28V and,
because of the ground-based current-sense resistor, the
______________________________________________________________________________________
19
MAX16818
Applications Information
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
output voltage can be as high as the input. The synchronous MOSFET keeps the power dissipation to a minimum,
especially when the input voltage is large when compared to the voltage on the LED string. It is important to
keep the current-sense resistor, R1, inside the LC loop,
so that ripple current is available. To regulate the LED
current, R2 creates a voltage that the differential amplifier
compares to 0.6V. If power dissipation is a problem in R2,
add a noninverting amplifier and reduce the value of the
sense resistor accordingly.
surface-mount inductor series available from various
manufacturers.
For example, for a buck regulator and 2 LEDs in series,
calculate the minimum inductance at VIN(MAX) = 13.2V,
VLED = 7.8V, ΔIL = 400mA, and fSW = 330kHz:
Buck regulators:
Inductor Selection
For a boost regulator with four LEDs in series, calculate
the minimum inductance at VIN(MAX) = 13.2V, VLED =
15.6V, ΔIL =400mA, and fSW = 330kHz:
Boost regulators:
The switching frequencies, peak inductor current, and
allowable ripple at the output determine the value and
size of the inductor. Selecting higher switching frequencies reduces the inductance requirement, but at the
cost of lower efficiency. The charge/discharge cycle of
the gate and drain capacitances in the switching
MOSFETs create switching losses. The situation worsens at higher input voltages, since switching losses are
proportional to the square of the input voltage. The
MAX16818 can operate up to 1.5MHz, however for
VIN > +12V, use lower switching frequencies to limit the
switching losses.
The following discussion is for buck or continuous
boost-mode topologies. Discontinuous boost, buckboost, and SEPIC topologies are quite different in
regards to component selection.
Use the following equations to determine the minimum
inductance value:
Buck regulators:
LMIN =
(VINMAX − VLED) x VLED
VINMAX x fSW x ΔIL
Boost regulators:
LMIN =
(VLED − VINMAX) x VINMAX
VLED x fSW x ΔIL
where VLED is the total voltage across the LED string.
As a first approximation choose the ripple current, ΔIL,
equal to approximately 40% of the output current.
Higher ripple current allows for smaller inductors, but it
also increases the output capacitance for a given voltage ripple requirement. Conversely, lower ripple current increases the inductance value, but allows the
output capacitor to reduce in size. This trade-off can be
altered once standard inductance and capacitance values are chosen. Choose inductors from the standard
20
LMIN =
LMIN =
(13.2 − 7.8) x 7.8
= 24.2μH
13.2 x 330k x 0.4
(15.6 − 13.2) x 13.2
= 15.3μH
15.6 x 330k x 0.4
The average-current-mode control feature of the
MAX16818 limits the maximum peak inductor current
and prevents the inductor from saturating. Choose an
inductor with a saturating current greater than the
worst-case peak inductor current. Use the following
equation to determine the worst-case inductor current:
LLPEAK =
VCL
ΔICL
+
RS
2
where R S is the inductor sense resistor and V CL =
0.0282V.
Switching MOSFETs
When choosing a MOSFET for voltage regulators, consider the total gate charge, RDS(ON), power dissipation,
and package thermal impedance. The product of the
MOSFET gate charge and on-resistance is a figure of
merit, with a lower number signifying better performance. Choose MOSFETs optimized for high-frequency switching applications.
The average current from the MAX16818 gate-drive
output is proportional to the total capacitance it drives
at DH and DL. The power dissipated in the MAX16818
is proportional to the input voltage and the average
drive current. See the IN, V CC, and V DD section to
determine the maximum total gate charge allowed from
the combined driver outputs. The gate-charge and
drain-capacitance (CV 2) loss, the cross-conduction
loss in the upper MOSFET due to finite rise/fall times,
and the I2R loss due to RMS current in the MOSFET
RDS(ON) account for the total losses in the MOSFET.
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Boost Regulator
Estimate the power loss (PDMOS_) caused by the MOSFET using the following equations:
PDMOS − HI = (QG x VDD x fSW ) +
PDFET = (QG x VDD x fSW ) +
⎛ VIN x IOUT x (tR + tF) x fSW ⎞
⎜
⎟ + (RDS(ON) x IRMS − HI2)
⎝
⎠
2
⎛ VIN x IOUT x (tR + tF) x fSW ⎞
⎜
⎟
⎝
⎠
2
2
+ (RDS(ON) x IRMS − HI )
IRMS − HI =
where QG, RDS(ON), tR, and tF are the upper-switching
MOSFET’s total gate charge, on-resistance at maximum
operating temperature, rise time, and fall time, respectively.
IRMS − HI =
(IVALLEY2 + IPK 2 + IVALLEY x IPK ) x
D
3
For the buck regulator, D = V LEDs / V IN, I VALLEY =
(IOUT - ΔIL / 2) and IPK = (IOUT + ΔIL / 2).
PDMOS − LO = (QG x VDD x fSW ) +
(RDS(ON) x IRMS − LO2)
IRMS − LO =
(IVALLEY2 + IPK2 + IVALLEY x IPK) x
(1− D)
3
For example, from the typical specifications in the
Applications Information section with VOUT = 7.8V, the
high-side and low-side MOSFET RMS currents are
0.77A and 0.63A, respectively, for a 1A buck regulator.
Ensure that the thermal impedance of the MOSFET
package keeps the junction temperature at least +25°C
below the absolute maximum rating. Use the following
equation to calculate the maximum junction temperature: TJ = (PDMOS x θJA) + TA, where θJA and TA are
the junction-to-ambient thermal impedance and ambient temperature, respectively.
To guarantee that there is no shoot-through from VIN to
PGND, the MAX16818 produces a nonoverlap time of
35ns. During this time, neither high- nor low-side MOSFET is conducting, and since the output inductor must
maintain current flow, the intrinsic body diode of the
low-side MOSFET becomes the conduction path. Since
this diode has a fairly large forward voltage, a Schottky
diode (in parallel to the low-side MOSFET) diverts current
flow from the MOSFET body diode because of its lower
forward voltage, which, in turn, increases efficiency.
(IVALLEY2 + IPK 2 + IVALLEY x IPK ) x
D
3
For a boost regulator in continuous mode, D = VLEDs /
(VIN + VLEDs), IVALLEY = (IOUT - ΔL / 2) and IPK = (IOUT
+ ΔIL / 2).
The voltage across the MOSFET:
VMOSFET = VLED + VF
where VF is the maximum forward voltage of the diode.
The output diode on a boost regulator must be rated to
handle the LED series voltage, VLED. It should also
have fast reverse-recovery characteristics and should
handle the average forward current that is equal to the
LED current.
Input Capacitors
For buck regulator designs, the discontinuous input
current waveform of the buck converter causes large
ripple currents in the input capacitor. The switching frequency, peak inductor current, and the allowable peakto-peak voltage ripple reflected back to the source
dictate the capacitance requirement. Increasing
switching frequency or paralleling out-of-phase converters lowers the peak-to-average current ratio, yielding a lower input capacitance requirement for the same
LED current. The input ripple is comprised of ΔV Q
(caused by the capacitor discharge) and ΔV ESR
(caused by the ESR of the capacitor). Use low-ESR
ceramic capacitors with high-ripple-current capability at
the input. Assume the contributions from the ESR and
capacitor discharge are equal to 30% and 70%, respectively. Calculate the input capacitance and ESR required
for a specified ripple using the following equation:
ESRIN =
ΔVESR
ΔIL ⎞
⎛
⎜IOUT +
⎟
⎝
2 ⎠
______________________________________________________________________________________
21
MAX16818
Buck Regulator
Estimate the power loss (PDMOS_) caused by the high-side
and low-side MOSFETs using the following equations:
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Current Limit
Buck:
I
x D(1− D)
CIN = OUT
ΔVQ x fSW
where IOUT is the output current of the converter. For
example, at VIN = 13.2V, VLED = 7.8V, IOUT = 1A, ΔIL =
0.4A, and fSW = 330kHz, the ESR and input capacitance are calculated for the input peak-to-peak ripple
of 100mV or less yielding an ESR and capacitance
value of 25mΩ and 10µF.
For boost regulator designs, the input-capacitor current
waveform is dominated by the inductor, a triangle wave
a magnitude of ΔIL. For simplicity’s sake, the current
waveform can be approximated by a square wave with
a magnitude that is half that of the triangle wave.
Calculate the input capacitance and ESR required for a
specified ripple using the following equation:
ESRIN =
ΔVESR
ΔIL
Boost:
ΔIL
x D
2
CIN =
ΔVQ x fSW
Duty cycle, D, for a boost regulator is equal to (VOUT VIN) / VOUT. As an example, at VIN = 13.2V, VLED =
15.6V, IOUT = 1A, ΔIL = 0.4A, and fSW = 330kHz, the
ESR and input capacitance are calculated for the input
peak-to-peak ripple of 100mV or less yielding an ESR
and capacitance value of 250mΩ and 1µF, respectively.
Output Capacitor
For buck converters, the inductor always connects to
the load, so the inductance controls the ripple current.
The output capacitance shunts a fraction of this ripple
current and the LED string absorbs the rest. The
capacitor reactance (which includes the capacitance
and ESR) and the dynamic impedance of the LED
diode string form a conductance divider that splits the
ripple current between the LEDs and the capacitor. In
many cases, the capacitor is very large as compared to
the ESR, and this divider reduces to the ESR and the
LED resistance.
In addition to the average current limit, the MAX16818
also has hiccup current limit. The hiccup current limit is
set to 10% below the average current limit to ensure that
the circuit goes in hiccup mode during continuous output short circuit. Connecting a resistor from LIM to
ground increases the hiccup current limit, while shorting
LIM to ground disables the hiccup current-limit circuit.
Average Current Limit
The average-current-mode control technique of the
MAX16818 accurately limits the maximum output current.
The MAX16818 senses the voltage across the sense
resistor and limit the peak inductor current (I L-PK )
accordingly. The on-cycle terminates when the currentsense voltage reaches 25.5mV (min). Use the following
equation to calculate the maximum current-sense resistor value:
RS =
PDR =
0.0255
IOUT
0.75 x 10 − 3
RS
where PDR is the dissipation in the series resistors.
Select a 5% lower value of RS to compensate for any
parasitics associated with the PCB. Also, select a noninductive resistor with the appropriate power rating.
Hiccup Current Limit
The hiccup current-limit value is always 10% lower than
the average current-limit threshold, when LIM is left
unconnected. Connect a resistor from LIM to SGND to
increase the hiccup current-limit value from 90% to
100% of the average current-limit value. The average
current-limit architecture accurately limits the average
output current to its current-limit threshold. If the hiccup
current limit is programmed to be equal or above the
average current-limit value, the output current does not
reach the point where the hiccup current limit can trigger. Program the hiccup current limit at least 5% below
the average current limit to ensure that the hiccup current-limit circuit triggers during overload. See the
Hiccup Current Limit vs. R EXT graph in the Typical
Operating Characteristics.
Boost converters place a harsher requirement on the
output capacitors as they must sustain the full load during the on-time of the MOSFET and are replenished
during the off-time. The ripple current in this case is the
full load current, and the holdup time is equal to the
duty cycle times the switching period.
22
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
VRAMP × fSW × L
A V × RS × VOUT × gm
fSW x L
RCF ≤ 105
RS x VOUT
RCF ≤
where VRAMP = 2V, gm = 550µs, AV = 34.5.
Boost:
VRAMP × fSW × L
A V × RS × (VOUT − VIN ) × gm
fSW x L
RCF ≤ 105
RS x (VOUT − VIN)
RCF ≤
Solving for the gain of the CEA amplifier,
Buck:
gm × RCF =
V
x fSW x L
ΔVCEA
= RAMP
VOUT x RS x AV
ΔVCA
ΔVCA
R x VIN x AV
= S
ΔVCEA
VRAMP x sL
where AV is the gain of the current amplifier (34.5) and
V RAMP is voltage peak (2V) of the internal ramp.
Multiplying the external loop gain with the CEA amplifier
gain gives the total loop equation and solves for the frequency that yields a gain of 1 results in:
Total Loop Buck:
fCMAX =
ΔVCEA
VRAMP x fSW x L
=
ΔVCA
(VOUT − VIN) x RS x AV
VIN x fSW
2πVOUT
To be stable, the gain of the CEA amplifier must have a
zero placed before fCMAX. CCF creates a pole at the
origin and the combination of RCF and CCF creates the
zero. Lower frequency zeros result in less bandwidth,
but greater phase margin. The pole created by CCFF
(in conjunction with RCF) is for noise reduction and can
be placed well past the crossover frequency.
The following equation describes the external loop gain
for a boost regulator:
External Loop Boost:
ΔVCA
R x VOUT x AV
= S
ΔVCEA
VRAMP x sL
To get the total loop gain for a boost regulator, multiply
the external loop gain with the gain of the CEA amplifier
to arrive at the following:
Total Loop Boost:
fCMAX =
Boost:
gm × RCF =
In order to choose CCF, the external loop gain must be
considered. The following equation describes the overall loop gain for a buck regulator, which is the ratio of a
small-signal change in the output of amplifier CA to the
output of amplifier CEA:
External Loop Buck:
fSW x VOUT
2π (VOUT − VIN)
As in the buck regulator, the zero created by RCF and
CCF sits at a frequency lower than fCMAX to maintain
stable operation.
______________________________________________________________________________________
23
MAX16818
Compensation
The main control loop consists of an inner current loop
(inductor current) and an outer LED current loop. The
MAX16818 uses an average current-mode control
scheme to regulate the LED current (Figure 7). The VEA
output provides the controlling voltage for the current
source. The inner current loop absorbs the inductor
pole reducing the order of the LED current loop to that
of a single-pole system.
The major consideration when designing the current
control loop is making certain that the inductor downslope (which becomes an upslope at the output of the
CEA) does not exceed the internal ramp slope. This is a
necessary condition to avoid subharmonic oscillations
similar to those in peak current mode with insufficient
slope compensation. This requires that the gain at the
output of the CEA be limited based on the following
equation (Figure 6):
Buck:
Power Dissipation
The TQFN is a thermally enhanced package and can dissipate about 2.7W. The high-power package makes the
high-frequency, high-current LED driver possible to operate from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
total VCC regulator output current (ICC). ICC includes quiescent current (IQ) and gate drive current (IDD):
7) Avoid long traces between the VDD bypass capacitors, the driver output of the MAX16818, the MOSFET gates, and PGND. Minimize the loop formed by
the VCC bypass capacitors, bootstrap diode, bootstrap capacitor, the MAX16818, and the upper
MOSFET gate.
8) Distribute the power components evenly across the
board for proper heat dissipation.
PD = VIN x ICC
ICC = IQ + [fSW x (QG1 + QG2)]
9) Provide enough copper area at and around the
switching MOSFETs, inductor, and sense resistors
to aid in thermal dissipation.
where QG1 and QG2 are the total gate charge of the lowside and high-side external MOSFETs at VGATE = 5V, IQ
is estimated from the Supply Current (IQ) vs. Frequency
graph in the Typical Operating Characteristics, and fSW
is the switching frequency of the LED driver. For boost
drivers, only consider one gate charge, QG1.
Use the following equation to calculate the maximum
power dissipation (PDMAX) in the chip at a given ambient temperature (TA):
PDMAX = 34.5 x (150 - TA) mW.
10) Use wide copper traces (2oz) to keep trace inductance and resistance low to maximize efficiency.
Wide traces also cool heat-generating components.
6) Run the current-sense lines CSP and CSN very
close to each other to minimize the loop area.
Similarly, run the remote voltage-sense lines
SENSE+ and SENSE- close to each other. Do not
cross these critical signal lines through power circuitry. Sense the current right at the pads of the
current-sense resistors.
24
DIFF
EAN
EAOUT
CLP
OVI
PCB Layout Guidelines
21
20
19
18
17
16
15
SGND
22
14
LIM
SENSE-
23
13
V_IOUT
SENSE+
24
12
RT/SYNC
SGND
25
IN
26
VCC
27
VDD
MAX16818
11
EN
10
PGOOD
9
CLKOUT
8
SGND
* EXPOSED PAD
28
5
6
7
N.C.
4
DH
3
LX
2
BST
1
DL
+
N.C.
Use the following guidelines to layout the switching
voltage regulator:
1) Place the IN, V CC , and V DD bypass capacitors
close to the MAX16818.
2) Minimize the area and length of the high current
loops from the input capacitor, upper switching
MOSFET, inductor, and output capacitor back to
the input capacitor negative terminal.
3) Keep short the current loop formed by the lower
switching MOSFET, inductor, and output capacitor.
4) Place the Schottky diodes close to the lower
MOSFETs and on the same side of the PCB.
5) Keep the SGND and PGND isolated and connect
them at one single point.
CSN
TOP VIEW
CSP
Pin Configuration
PGND
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
TQFN
Chip Information
TRANSISTOR COUNT: 5654
PROCESS: BiCMOS
______________________________________________________________________________________
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
QFN THIN.EPS
______________________________________________________________________________________
25
MAX16818
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
MAX16818
1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2006 Maxim Integrated Products
Heaney
is a registered trademark of Maxim Integrated Products, Inc.